WO2014077281A1

WO2014077281A1 - Power conversion apparatus

Info

Publication number: WO2014077281A1
Application number: PCT/JP2013/080694
Authority: WO
Inventors: 浩志田村; 久保　謙二; 充休波
Original assignee: 日立オートモティブシステムズ株式会社
Priority date: 2012-11-15
Filing date: 2013-11-13
Publication date: 2014-05-22
Also published as: JP2014100022A; JP5933418B2

Abstract

The present invention suppresses bias magnetism without deteriorating control responsiveness or stability of output voltages. This power conversion apparatus is provided with: a switching circuit, which applies voltages in one direction and the reverse direction to a primary winding wire of a transformer by sequentially switching switching elements, each of which is configured from an upper arm and a lower arm; a rectifying circuit, which rectifies an alternating current output generated from a secondary winding wire of the transformer; and a control apparatus that on/off controls the switching elements. The control apparatus is provided with a detector that detects bias magnetism in the transformer, and corresponding to an output of the detector, the control apparatus increases, by a predetermined quantity, an application time of a voltage to be applied in the one direction, and reduces, by the predetermined quantity, an application time of a voltage to be applied in the reverse direction, thereby reducing the bias magnetism.

Description

Power converter

The present invention relates to a power conversion device, and more particularly to a power conversion device capable of suppressing the bias magnetism of a transformer used.

Patent Document 1 is known as a technique for suppressing the bias magnetism of a transformer used in a power converter. In this document, a switching signal (ON) is detected which detects the amount of biased magnetism of a transformer and supplies the switching elements that constitute a pair of upper and lower arms among the switching elements that configure the inverter unit according to the detected amount of biased magnetism. It is described that the amount of biased magnetization is reduced by changing the duty ratio of the off signal).

JP 2003-37973 A

However, according to the technology described in Patent Document 1, only the duty ratio of the on / off signal supplied to the switching elements constituting the pair of upper and lower arms is changed according to the amount of biased magnetization.

Therefore, the voltage output by the power conversion device does not accurately follow the voltage command value. That is, in such a biased magnetization suppression method, the control response and stability of the output voltage of the power conversion device may be degraded. The present invention has been made in view of such problems, and Abstract: A power converter capable of suppressing biased magnetization without deteriorating control response and stability.

The present invention adopts the following means in order to solve the above problems.

A switching circuit that sequentially switches switching elements including an upper arm and a lower arm to apply a voltage in one direction and a reverse direction to the primary winding of the transformer, and rectifies an AC output generated in the secondary winding of the transformer And a controller for controlling on / off of the plurality of switching elements, the controller including a detector for detecting a bias magnetism of the transformer, and applying in one direction according to an output of the detector The application time of the voltage to be applied is increased by a predetermined amount, and the application time of the voltage applied in the reverse direction is decreased by the predetermined amount to reduce the biased magnetization.

Since the present invention has the above configuration, it is possible to suppress the bias magnetism of the power conversion device without deteriorating the control response or the stability of the output voltage.

It is a figure explaining the power converter concerning a 1st embodiment. It is a figure explaining the control device of a power converter. It is a figure explaining the outline of phase shift PWM control. It is the schematic which shows the relationship between the amount V21 of positive magnetic deviation, and the amount Dcomp of duty corrections. FIG. 7 is a schematic view showing a relationship between a negative amount of magnetic bias V21 and a duty correction amount Dcomp. It is a figure which shows the relationship between switching instruction | command H1-H4 in case the duty correction amount Dcomp is a negative value, and switching instruction | command correction value H1'-H4 '. FIG. 7 is a diagram showing the relationship between switching commands H1 to H4 and switching command correction values H1 'to H4' when duty correction amount Dcomp is a positive value. FIG. 17 is a diagram showing a relationship between gate voltages V30 to V60 supplied when the potential difference V20 is a positive value and voltages applied to the primary winding 40 of the transformer 50. FIG. 17 is a diagram showing a relationship between gate voltages V30 to V60 supplied when the potential difference V20 is a negative value and a voltage applied to the primary winding 40 of the transformer 50. It is a figure explaining the DC-DC converter using a center tap type transformer and an active clamp circuit. It is a figure explaining a current doubler type DC-DC converter. It is a figure explaining the power converter concerning a 2nd embodiment. It is a figure explaining the power converter concerning 3rd Embodiment.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.

(Embodiment 1)
FIG. 1 is a diagram for explaining a power converter (DC-DC converter) according to a first embodiment of the present invention.

(Description of primary side circuit)
The primary side (high voltage side) of the power conversion device is a middle point where one end of the smoothing capacitor 20, the drain of the MOSFET 210 of the upper arm of the first phase, the cathode of the diode 250 and one end of the snubber capacitor 140, The drain of the two-phase upper arm MOSFET 270, the cathode of the diode 270, and one end of the snubber capacitor 160 are connected to a middle point connected.

The middle point where the source of the first phase upper arm MOSFET 210 and the anode of the diode 250 and one end of the snubber capacitor 140 are connected is the drain of the first phase lower arm MOSFET 220 and the cathode of the diode 260 and one end of the snubber capacitor 150 Are connected to one end of the resonant inductor 30.

The other end of the resonant inductor 30 is connected to one end of the primary winding 40 of the transformer 50, and the other end of the primary winding 40 of the transformer 50 is the source of the MOSFET 230 of the upper arm of the second phase , And the middle point where the anode of diode 270 and one end of snubber capacitor 160 are connected, and the middle point where the drain side of MOSFET 240 of the lower arm of the second phase, the cathode side of diode 280 and one end of snubber capacitor 170 are connected Be done.

Here, the resonant inductor 30 may be replaced by the leakage inductance or wiring inductance of the transformer 50.

The source of the first phase lower arm MOSFET 220, the anode of the diode 260 and one end of the snubber capacitor 150 are connected to each other at the source of the second phase lower arm MOSFET 240, the anode of the diode 280 and one end of the snubber capacitor 170 Are connected to one end of the smoothing capacitor 20 and the low potential side of the DC power supply 10.

(Description of secondary circuit)
The high potential side of the DC power supply 100 on the secondary side (low voltage side) of the power conversion device is connected to one end of the load 110, one end of the smoothing capacitor 90, and one end of the smoothing inductor 80.

The other end of the smoothing inductor 80 is connected to a midpoint between the secondary winding 60 and the secondary winding 70 of the transformer 50 via the current sensor 200.

Here, for the current sensor 200, a hole current sensor or a shunt resistor is used.

The other end of the secondary winding 60 of the transformer 50 is connected to one end of the resistor 120 of the low pass filter 135 and the cathode side of the rectifying diode 300.

The other end of the secondary side winding 70 of the transformer 50 is connected to one end of the capacitor 130 of the low pass filter 135 and the cathode side of the rectifying diode 290.

The anode of the rectification diode 290 on the secondary side is connected to the anode of the rectification diode 300, one end of the smoothing capacitor 90, the low potential side of the DC power supply 100, and one end of the load 110.

The other end of the resistor 120 of the low pass filter 135 is connected to the other end of the capacitor 130 of the low pass filter 135.

Here, the cutoff frequency of the low pass filter 135 is set to a value that can sufficiently attenuate the switching frequency components of the MOSFETs 210 to 240 on the primary side.

Specifically, it is desirable to set the cutoff frequency of the low pass filter 135 to 1/10 or less of the switching frequency of the MOSFETs 210 to 240 on the primary side.

The voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected potential difference to the control device 310.

Here, the potential difference between both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 is the amount of biased magnetization of the transformer 50.

The bias magnetism of the transformer 50 is a variation in on resistance of the MOSFETs 210 to 240 on the primary side, or a variation in rise time or fall time at the time of switching of the MOSFETs 210 to 240, a variation in impedance of the main circuit wiring, or the primary side. This is caused by the voltage fluctuation of the DC power supply 10 or the like.

The voltage sensor 190 is connected to both ends of the smoothing capacitor 90 on the secondary side, detects a potential difference between the both ends of the smoothing capacitor 90, and inputs the detected potential difference to the control device 310.

Here, as the

voltage sensors

180 and 190, a non-inverted amplification circuit configured by an operational amplifier or the like is used.

The current sensor 200 is attached to a wire connecting the middle point of the secondary side winding 60 and the secondary side winding 70 of the transformer 50 and the smoothing inductor 80, and detects the current flowing in the smoothing inductor 80 for control. Input to the device 310.

The position where the current sensor 200 is attached may be a wiring portion connecting the anode of the secondary side rectifier diode 300 and one end of the smoothing capacitor 90. Further, although the rectifying

diodes

290 and 300 are used as the rectifying elements on the secondary side, there is no problem even if they are changed to MOSFETs.

(Description of control device)
FIG. 2 is a diagram for explaining a control device of the power conversion device according to the first embodiment of the present invention.

The control device 310 includes an A / D converter 320, a duty command generation unit 330, a switching command generation unit 340, a correction amount calculation unit 350 that calculates the duty correction amount Dcomp, a switching command correction unit 360, and a gate drive circuit 370.

The duty command generation unit 330 detects the output voltage command Vref (voltage command to the smoothing capacitor 90 on the secondary side) and the output of the A / D converter 320 that converts an analog value to a digital value, that is, the voltage sensor 190 The duty command Dref is generated using a digital value V11 representing the potential difference V10 across the smoothing capacitor 90 and a digital value I11 representing the current I10 flowing through the smoothing inductor 200 detected by the current sensor 200.

Further, switching command generation unit 340 generates switching commands H1 to H4 based on duty command Dref.

The correction amount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the digital value V21 of the potential difference V20 of the both ends of the capacitor 180 which constitutes the low pass filter 135 detected by the voltage sensor 180.

Further, switching command correction unit 360 corrects the duty of switching commands H1 to H4 based on duty correction amount Dcomp. The gate drive circuit 370 converts the corrected switching commands H1 'to H4' into gate voltages V30 to V60 of the MOSFETs 210 to 240 on the primary side.

(Description of each block constituting the control device)
(Description of A / D converter)
The A / D converter 320 converts the potential difference V10 across the smoothing capacitor 90 detected by the voltage sensor 190 into a digital value V11, and converts the converted digital value V11 (hereinafter referred to as the output voltage V11) into a duty command generating unit Input to 330.

The A / D converter 320 converts the current I10 flowing through the smoothing inductor 80 detected by the current sensor 200 into a digital value I11, and converts the converted digital value I11 (hereinafter referred to as the smoothing inductor current I11) into a duty command Input to the generation unit 330.

Further, the A / D converter 320 converts the potential difference V20 at both ends of the capacitor 130 of the low pass filter 135 detected by the voltage sensor 180 into a digital value V21 and converts it into a digital value V21 (hereinafter referred to as a biased magnetization amount V21). ) Is input to the correction amount calculation unit 350.

(Description of duty command generator)
Duty command generation unit 330 compares output voltage command Vref with output voltage V11 to calculate a voltage deviation, converts the calculated voltage deviation into a current command of smoothing inductor 200 by proportional integral control, etc. A current deviation is calculated by comparing the smoothed inductor current I11, the calculated current deviation is converted into a duty command Dref by proportional integral control or the like, and the converted duty command Dref is input to the switching command generation unit 340.

Duty command generation unit 330 may compare output voltage command Vref with output voltage V11 to calculate a voltage deviation, and may directly convert the calculated voltage deviation into duty command Dref by proportional integral control or the like (switching command). Description of generation unit)
Switching command generation unit 340 generates switching commands H1 to H4 based on input duty command Dref. The switching commands H1 to H4 are switching commands for turning on and off the primary side MOSFETs 210 to 240, respectively.

As a method of generating the switching commands H1 to H4 from the duty command Dref, there is, for example, phase shift PWM control.

FIG. 3 is a diagram for explaining an outline of phase shift PWM control in the first embodiment.

The phase shift PWM control is a method of fixing the ratio of the on time to the off time to 50% and changing the phase difference between the switching signals H1 to H4, and the on overlap period of H1 and H4 and the on of H2 and H3. The overlap period is adjusted, and a voltage corresponding to the duty command Dref is output.

Here, a method of generating switching commands H1 to H4 based on switching command H4 of MOSFET 240 on the primary side will be described as an example.

First, the switching command H4 is generated as a pulse signal in which the ratio of the on time to the off time is fixed to 50%. For example, when the switching frequency is 100 kHz, the on time and the off time are each 5 μs.

The switching command H3 is generated by inverting the on / off signal of the switching command H4. As a result, the switching command H3 is turned off during the on period of the switching command H4, and is turned on during the off period of the switching command H4.

The switching command H2 is turned on at a timing when the on overlap period of the switching command H3 and the switching command H2 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.

The switching command H1 is turned on at a timing when the on overlap period of the switching command H4 and the switching command H1 coincides with the duty command, and is turned off when 50% of one switching cycle has elapsed from the moment of turning on.

As described above, by generating the switching commands H1 to H4, the power conversion device can output a voltage corresponding to the duty command.

In order to prevent short circuiting of the MOSFETs of the upper and lower arms of each phase, it is desirable to provide dead times for the switching commands H1 to H4, respectively, but here the dead times are omitted and described.

(Description of correction amount calculation unit)
The correction amount calculation unit 350 calculates the duty correction amount Dcomp of the switching command H1 to H4 based on the input bias amount V21 for each switching cycle, and the calculated duty correction amount Dcomp is used as the switching command correction unit 360. Enter in

FIG. 4 is a schematic diagram showing the relationship between the amount of positive magnetic deviation V21 and the amount of duty correction Dcomp in the first embodiment.

FIG. 5 is a schematic view showing the relationship between the negative amount of biased magnetization V21 and the duty correction amount Dcomp in the first embodiment.

When the switching command H3 falls, the correction amount calculation unit 350 compares the input bias amount V21 with the command value zero to calculate the deviation, and converts the calculated deviation into the duty correction amount Dcomp by proportional integral control or the like. Do.

When the amount of biased magnetization V21 is a positive value, the duty correction amount Dcomp is converted to a negative value, and when the amount of biased magnetization V21 is a negative value, the duty correction amount Dcomp is positive. Convert to a value.

Then, the correction amount calculation unit 350 inputs the converted duty correction amount Dcomp into the switching command correction unit 360.

Here, although the duty correction amount Dcomp is set to be calculated when the switching command H3 falls, it may be calculated when any of the switching commands H1 to H4 rises or falls. .

(Description of switching command correction unit)
Switching command correction unit 360 corrects the duty of switching commands H1 to H4 based on the input duty correction amount Dcomp.

FIG. 6 is a diagram showing the relationship between the switching commands H1 to H4 and the switching command correction values H1 'to H4' when the duty correction amount Dcomp is a negative value in the first embodiment.

When the input duty correction amount Dcomp is a negative value, the switching command correction unit 360 delays the ON timing of the switching command H1 of the first-phase upper arm MOSFET 210 by the duty correction amount Dcomp. A switching command correction value H1 'is generated.

Further, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H2 of the MOSFET 220 of the lower arm of the first phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H2 'is generated.

In addition, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H3 of the MOSFET 230 of the upper arm of the second phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H3 'is generated.

In addition, when the duty correction amount Dcomp input is a negative value, the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned on by the duty correction amount Dcomp. Thus, the switching command correction value H4 'is generated.

FIG. 7 shows the relationship between switching commands H1 to H4 and switching command correction values H1 'to H4' when the duty correction amount Dcomp is a positive value in the first embodiment.

When the input duty correction amount Dcomp is a positive value, the switching command correction unit 360 delays the timing at which the switching command H1 of the MOSFET 210 of the upper arm of the first phase is turned off by the duty correction amount Dcomp. A switching command correction value H1 'is generated.

In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the ON timing of the switching command H2 of the MOSFET 220 of the lower arm of the first phase by the duty correction amount Dcomp. Thus, the switching command correction value H2 'is generated.

In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the ON timing of the switching command H3 of the MOSFET 230 of the upper arm of the second phase by the duty correction amount Dcomp. Thus, the switching command correction value H3 'is generated.

In addition, when the duty correction amount Dcomp input is a positive value, the switching command correction unit 360 delays the timing at which the switching command H4 of the MOSFET 240 of the lower arm of the second phase is turned off by the duty correction amount Dcomp. Thus, the switching command correction value H4 'is generated.

(Description of gate drive circuit)
The gate drive circuit 370 converts the inputted switching command correction values H1 ′ to H4 ′ into gate voltages V30 to V60, and inputs the converted gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240.

Thus, the MOSFETs 210 to 240 are driven in accordance with the on / off signals of the gate voltages V30 to V60.

(Description of biased magnetization suppression method)
The bias magnetism of the transformer 50 can be suppressed by controlling the switching of the MOSFETs 210 to 240 of the power converter as described above.

FIG. 8 shows the gate voltages V30 to V60 supplied when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a positive value and the primary of the transformer 50 in the first embodiment. FIG. 7 is a diagram showing a relationship of voltages applied to a side winding 40.

By performing on / off control of the switching of the MOSFETs 210 to 240 of the power converter as described above, (1) when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a positive value The on time of signal V30 is less than 50% of one switching cycle, the off time of gate signal V30 is more than 50% of one switching cycle, and the on time of gate signal V40 is more than 50% of one switching cycle, The off time of the gate signal V40 is less than 50% of one switching cycle, the on time of the gate signal V50 is 50% or more of one switching cycle, and the off time of the gate signal V50 is less than 50% of one switching cycle. , The on time of the gate signal V60 is one cycle of switching It becomes less than 50%, off-time of the gate signal V60 becomes a switching cycle of 50% or more.

(2) For this reason, the on overlap period of the gate voltage V30 and the gate voltage V60 is shorter than the on overlap period of the gate voltage V40 and the gate voltage V50, the voltage is applied to the primary side winding 40 of the transformer 50. Voltage is longer than the period during which a negative voltage is applied to a positive voltage.

(3) The voltage applied to the primary side winding 40 of the transformer 50 is transformed and output to the

secondary side windings

60 and 70 of the transformer 50 according to the turns ratio of the transformer 50, The potential difference across the low pass filter 130 increases in the negative direction.

That is, since the potential difference V20 between both ends of the capacitor 130 of the low pass filter detected by the voltage sensor 180 shown in FIG. 8 approaches zero, the bias magnetism of the transformer 50 is suppressed.

Since the on overlap period of the gate voltage V30 and the gate voltage V60 is shorter than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is smaller than the voltage command value. However, since the on overlap period of the gate voltage V40 and the gate voltage V50 becomes longer than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device becomes a larger value than the voltage command value.

That is, since the on overlap period of gate voltage V40 and gate voltage V50 becomes longer by the amount that the on overlap period of gate voltage V30 and gate voltage V60 becomes shorter, the voltage output by the power conversion device in the switching cycle becomes a voltage command Match the value.

Thereby, it is possible to suppress the biased magnetization without deteriorating the control response or the stability of the output voltage of the power conversion device. Vhigh described in FIG. 8 is a voltage value of the DC power supply 10 on the primary side.

FIG. 9 shows the gate voltage V30 to V60 and the primary side winding of the transformer 50 when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a negative value in the first embodiment. FIG. 7 is a diagram showing the relationship of voltages applied to a line 40.

By performing on / off control of the switching of the MOSFETs 210 to 240 of the power conversion device as described above, (1) when the potential difference V20 across the capacitor 130 of the low pass filter detected by the voltage sensor 180 is a negative value The on time of the signal V30 is 50% or more of one switching cycle, the off time of the gate signal V30 is less than 50% of one switching cycle, and the on time of the gate signal V40 is less than 50% of one switching cycle, The off time of the gate signal V40 is 50% or more of one switching cycle, the on time of the gate signal V50 is less than 50% of one switching cycle, and the off time of the gate signal V50 is 50% or more of one switching cycle , The on time of the gate signal V60 is one cycle of switching It is 50% or more of the off time of the gate signal V60 becomes less than 50% of the switching cycle.

(2) For this reason, the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the on overlap period of the gate voltage V30 and the gate voltage V60, and thus applied to the primary side winding 40 of the transformer 50. Voltage decreases the period during which a negative voltage is applied to a positive voltage.

secondary side windings

60 and 70 of the transformer 50 according to the turns ratio of the transformer 50, The potential difference between both ends of the low pass filter 130 increases in the positive direction.

That is, since the potential difference V20 between both ends of the capacitor 130 of the low-pass filter detected by the voltage sensor 180 shown in FIG. 9 approaches zero, the bias magnetism of the transformer 50 is suppressed.

Since the on overlap period of the gate voltage V30 and the gate voltage V60 is longer than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is larger than the voltage command value. However, since the on overlap period of the gate voltage V40 and the gate voltage V50 is shorter than the duty command value Dref by the duty correction amount Dcomp, the voltage output by the power conversion device is smaller than the voltage command value.

That is, since the on overlap period of gate voltage V40 and gate voltage V50 is shortened by the lengthened on overlap period of gate voltage V30 and gate voltage V60, the voltage output by the power conversion device is a voltage command in one switching cycle. Match the value.

Thereby, it is possible to suppress the biased magnetization without deteriorating the control response or the stability of the output voltage of the power conversion device.

Note that the above-described method for suppressing the biased magnetization is not limited to the DC-DC converter using the center tap transformer shown in FIG. 1, but a DC-DC converter using the center tap transformer and the active clamp circuit shown in FIG. The present invention can be applied to the current doubler type DC-DC converter shown in FIG.

Second Embodiment
FIG. 12 is a diagram for explaining the power conversion device according to the second embodiment.

In the first embodiment described above, the voltage generated by the

secondary side windings

60 and 70 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed. In the present embodiment, the voltage applied to the primary side winding 40 of the transformer 50 is detected by the voltage sensor 180 through the low pass filter 135, and the bias magnetism of the transformer 50 is detected by the bias control method described above. Suppress.

In addition, since the circuit system of a power converter and the control apparatus 310 are the same as Embodiment 1, description is abbreviate | omitted.

One end of the resistor 120 of the low pass filter 135 is connected to the middle point where one end of the resonant inductor 30 and one end of the primary side winding 40 of the transformer 50 are connected, and the other end of the resistor 120 of the low pass filter 135 is One end of the capacitor 130 of the low pass filter 135 is connected, and the other end of the capacitor 130 of the low pass filter 135 is connected to the other end of the primary winding 40 of the transformer 50.

The voltage sensor 180 is connected to both ends of the capacitor 130 of the low pass filter 135, detects a potential difference between both ends of the capacitor 130 of the low pass filter 135, and inputs the detected voltage V20 to the control device 310.

The controller 310 generates the gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment, inputs the generated gate voltages V30 to V60 to the gates of the MOSFETs 210 to 240, and turns on the MOSFETs 210 to 240, Turn off. Thereby, the biased magnetism of the transformer 50 is suppressed.

(Embodiment 3)
FIG. 13 is a diagram for explaining the power conversion device according to the third embodiment.

In the first embodiment described above, the voltage generated by the

secondary side windings

60 and 70 of the transformer 50 is detected by the voltage sensor 180 via the low pass filter 135, and the bias magnetism of the transformer 50 is suppressed. In the present embodiment, the current flowing through the resonance inductor 30 is detected by the current sensor 600, the detected current value is converted to a voltage value through the low pass filter 135, and the converted voltage value is detected by the voltage sensor 180 and detected. The input voltage is input to the controller 310. Control device 310 suppresses the biased magnetism of transformer 50 in the same manner as in the first embodiment.

The first terminal of the current sensor 600 is connected to one end of the resonant inductor 30, and the second terminal of the current sensor 600 is connected to one end of the primary winding 40 of the transformer 50. Is connected to one end of the resistor 120 of the low pass filter 135.

The other end of the resistor 120 constituting the low pass filter 135 is connected to one end of the capacitor 130 of the low pass filter 135, and the other end of the capacitor 130 of the low pass filter 135 is connected to the ground.

The controller 310 generates gate voltages V30 to V60 of the MOSFETs 210 to 240 in the same manner as in the first embodiment. The generated gate voltages V30 to V60 are input to the gates of the MOSFETs 210 to 240, respectively, and the MOSFETs 210 to 240 are turned on and off, whereby the bias magnetism of the transformer 50 is suppressed.

As described above, according to the embodiment of the present invention, the ratio of the on time to the off time is fixed to 50%, and each switching command (H1 (gate voltage V30), H2 (gate voltage V40), H3 (h). In a power conversion device provided with a phase shift PWM control device that changes the phase difference between gate voltages V50) and H4 (gate voltage V60), when potential difference V20 across capacitor 130 is a positive value, gate voltage V30 and The on overlap period of the gate voltage V60 is corrected to be shorter than the on overlap period of the gate voltage V40 and the gate voltage V50. For this reason, the voltage applied to the primary side winding 40 of the transformer 50 has a long period during which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the negative direction.

On the other hand, when the potential difference V20 between both ends of the capacitor 130 is a negative value, the on overlap period of the gate voltage V30 and the gate voltage V60 is corrected to be longer than the on overlap period of the gate voltage V40 and the gate voltage V50. For this reason, the voltage applied to the primary side winding 40 of the transformer 50 has a short period in which a negative voltage is applied to a positive voltage. Therefore, the potential difference between both ends of the low pass filter 130 increases in the positive direction. Thus, the potential difference V20 across the capacitor 130 of the low-pass filter detected by the voltage sensor 180 approaches zero and the bias magnetism of the transformer 50 is suppressed. The present invention is limited to the above embodiment. Rather, various modifications are included. For example, the above-described embodiment is described in detail to explain the present invention in an easy-to-understand manner, and is not necessarily limited to one having all the described configurations.

Further, part of the configuration of one embodiment can be replaced with the configuration of another embodiment, and the configuration of another embodiment can be added to the configuration of one embodiment. Moreover, it is possible to add, delete, and replace other configurations for part of the configurations of the respective embodiments. Further, each of the configurations, functions, processing units, processing means, etc. described above may be realized by hardware, for example, by designing part or all of them with an integrated circuit. Further, each configuration, function, etc. described above may be realized by software by the processor interpreting and executing a program that realizes each function. Information such as programs and files for realizing each function can be placed in a memory or a recording apparatus such as a hard disk, SSD (Solid State Drive), or an IC card, an SD card, a DVD, etc. . Further, control lines or information lines indicate what is considered to be necessary for the description, and not all control lines or information lines in a product are shown. In practice, almost all configurations may be considered to be mutually connected.

10 ... DC power supply on primary side, 20, 90 ... smoothing capacitor, 30 ... inductor for resonance
40 ... Primary winding of transformer, 50 ... Transformer, 60, 70 ... Secondary winding of transformer,
80, 490, 500 ... smoothing inductor, 100 ... secondary side DC power supply, 110 ... load,
120 ... resistance of low pass filter, 130 ... capacitor of low pass filter,
135 ... low pass filter, 140, 150, 160, 170 ... snubber capacitor,
180, 190 ... voltage sensor, 200, 600 ... current sensor,
210, 220, 230, 240, 400, 410, 420, 430 ... MOSFET,
250, 260, 270, 280, 290, 300, 440, 450, 460, 470 ... diode, 310 ... controller, 320 ... A / D converter, 330 ... duty command generator,
340: switching command generation unit, 350: correction amount calculation unit,
360 ... switching command correction unit, 370 ... gate drive circuit,
480 ... clamp capacitor

Claims

A switching circuit that sequentially switches switching elements including an upper arm and a lower arm to apply a voltage in one direction and a reverse direction to the primary winding of the transformer, and rectifies an AC output generated in the secondary winding of the transformer A control circuit for on / off controlling the plurality of switching elements,
The control device includes a detector for detecting the bias magnetism of the transformer, and the application time of the voltage applied in one direction is increased by a predetermined amount according to the output of the detector, and the application of the voltage applied in the reverse direction A power converter characterized by reducing time by the predetermined amount to reduce the bias.
Bridge type switching comprising a first phase switching element consisting of an upper arm and a lower arm and a second phase switching element consisting of an upper arm and a lower arm and switching a DC input to supply a primary winding of a transformer Circuit,
A rectifier circuit that rectifies an AC output generated in a secondary winding of the transformer;
A control device for driving the switching elements forming the upper arm and the lower arm of the same phase in opposite phases and driving the switching elements forming the first phase and the second phase with a phase difference;
The control device includes a detector for detecting a bias magnetism of the transformer, and the switching device generates an upper arm of the first phase and a lower arm of the second phase connected in a bridge according to the detector output. One of the duty ratio of the switching output to be switched and the duty ratio of the switching output generated by the switching element forming the upper arm and the lower arm of the first phase of the bridge-connected second phase are increased. A power converter characterized by reducing.
In the power converter according to claim 2,
The control device delays the timing of turning on the upper arm of the first phase and the timing of turning on the lower arm of the second phase by a predetermined amount,
A power converter characterized in that the timing to turn off the upper arm of the second phase and the timing to turn off the lower arm of the first phase are delayed by the predetermined amount.
In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a DC component included in an AC voltage on the primary side of the transformer.
In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a direct current component included in an alternating voltage on the secondary side of the transformer.
In the power converter according to claim 1 or 2,
The power conversion device according to claim 1, wherein the amount of biased magnetization of the transformer is calculated based on a direct current component of a current flowing through a resonant inductor inserted in the primary side of the transformer.
In the power converter according to claim 2,
The ON time of the switching element forming the upper arm of the first phase and the switching element forming the lower arm of the second phase is the switching element forming the lower arm of the first phase and the switching forming the upper arm of the second phase A power converter characterized in that it is reduced by an increase in the on time of the element and increased by a decrease in the on time of the switching element constituting the upper arm of the lower arm of the first phase and the upper arm of the second phase. .