WO2013120363A1 - 逆变器电路及逆变器电路的控制方法 - Google Patents

逆变器电路及逆变器电路的控制方法 Download PDF

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Publication number
WO2013120363A1
WO2013120363A1 PCT/CN2012/083905 CN2012083905W WO2013120363A1 WO 2013120363 A1 WO2013120363 A1 WO 2013120363A1 CN 2012083905 W CN2012083905 W CN 2012083905W WO 2013120363 A1 WO2013120363 A1 WO 2013120363A1
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Prior art keywords
inductor
switch tube
current
tube
diode unit
Prior art date
Application number
PCT/CN2012/083905
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English (en)
French (fr)
Inventor
张彦忠
Original Assignee
华为技术有限公司
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Publication date
Application filed by 华为技术有限公司 filed Critical 华为技术有限公司
Priority to AU2012343352A priority Critical patent/AU2012343352B2/en
Priority to EP12840852.3A priority patent/EP2731252B1/en
Publication of WO2013120363A1 publication Critical patent/WO2013120363A1/zh

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Classifications

    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00Details of apparatus for conversion
    • H02M1/0048Circuits or arrangements for reducing losses
    • H02M1/0054Transistor switching losses
    • H02M1/0058Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • HELECTRICITY
    • H02GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02MAPPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815Resonant converters
    • YGENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02BCLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present invention relates to the field of power electronics, and in particular, to an inverter circuit and a method for controlling an inverter circuit.
  • An inverter circuit is a circuit for converting direct current into alternating current, and is also a circuit for converting direct current energy into alternating current energy, which includes an output alternating voltage form, such as an uninterruptible power supply; and an output alternating current following an external alternating current voltage.
  • the form of current such as solar grid-connected inverters, wind-connected grid generators, etc.
  • high-frequency switching of the switching elements in the inverter circuit is required. Due to the different switching losses generated by the switching elements during the switching between turn-on and turn-off, the switches can be divided into soft switches and hard switches.
  • the voltage across the switch drops and the current flowing through the switch rises.
  • the power dissipation caused by the overlap in time is the turn-on loss; or during the switch-off process, the current flowing through the switch drops and the voltage rise across the switch overlaps in time to generate a power-off loss.
  • a switch that cannot avoid switching loss is called a hard switch, and a switch that can avoid switching loss is called a soft switch.
  • Embodiments of the present invention provide a method for controlling an inverter circuit and an inverter circuit for implementing soft switching in an inverter circuit.
  • An embodiment of the present invention provides an inverter circuit including a DC source, an AC circuit, and one or more inverter bridges, where the inverter bridge includes a first bridge arm and a second bridge arm, and the first bridge Arm Connecting a positive pole and a negative pole of the direct current source respectively, the second bridge arm is respectively connected to a positive pole and a negative pole of the direct current source, and the first bridge arm and the second bridge arm are arranged in parallel; the first bridge arm The first switching element group and the second switching element group are connected in series, the first switching element group includes a first switching tube and a first diode unit connected in parallel, a negative pole of the first diode unit and the a positive electrode connection of the DC source, the second switching element group includes a second switching tube and a second diode unit connected in parallel, and a cathode
  • the second bridge arm includes a third switching element group and a fourth switching element group connected in series, the third switching element group includes a third switching tube and a third diode unit connected in parallel, the third diode a cathode of the unit is connected to the anode of the DC source, the fourth group of switching elements includes a fourth switch tube and a fourth diode unit connected in parallel, a cathode of the fourth diode unit and the DC source
  • the AC circuit includes a first AC source and a first capacitor connected in parallel, or includes a first AC load and a first capacitor connected in parallel, and one end of the AC circuit is connected to the first bridge arm through a first inductor Between the first switching element group and the second switching element group, the other end of the alternating current circuit is connected between the third switching element group and the fourth switching element group of the second bridge arm by a second inductance.
  • An embodiment of the present invention provides a control method for the above inverter circuit.
  • control is performed according to the following method: closing the first switch tube and the fourth switch a tube, the second switch tube and the third switch tube are disconnected, so that an inductor current flowing through the first inductor and the second inductor is gradually increased, and the inductor current is divided into a first AC source or a first AC load. a current, and a current that charges the first capacitor; after the inductor current flowing through the first inductor and the second inductor reaches a peak for a first period of time that is continuously set, the first switch transistor is turned off;
  • the second diode unit is in an on state, and the second switching tube is closed to achieve low voltage conduction of the second switching tube.
  • An embodiment of the present invention provides another control method for the above inverter circuit. For any inverter bridge in the inverter circuit, control is performed according to the following method: closing the first switch tube and the fourth Switching tube, disconnecting the second switching tube and the third switching tube, Increasing the inductor current flowing through the first inductor and the second inductor, the inductor current being divided into a current flowing through the first alternating current source or the first alternating current load, and a current charging the first capacitor; After the first period of time causes the inductor current flowing through the first inductor and the second inductor to reach a peak value, the first switch tube and the fourth switch tube are disconnected;
  • the second diode unit and the third diode unit are in an on state, and the second switching tube and the third switching tube are closed to achieve low voltage conduction of the second switching tube and the third switching tube.
  • the embodiment of the present invention provides another control method for the inverter circuit. When the AC circuit includes the first AC source, any inverter bridge in the inverter circuit is performed according to the following method. Control:
  • the first diode unit is in an on state, and the first switching tube is closed to achieve low voltage conduction of the first switching tube.
  • the inverter circuit and the control method of the inverter circuit provided by the embodiments of the present invention provide a parallel switch tube and a diode unit in a switching element group of each inverter bridge, and a parallel voltage source is arranged in the AC circuit. And the first capacitor, or the parallel capacitor and the AC load, and through appropriate control methods, the switch tube can be turned off in a low voltage state, and turned on in a low current state, so that the switch tube has a soft switch Function, which can effectively reduce the energy loss during the opening and closing of the light.
  • FIG. 1 is a schematic structural view 1 of an inverter circuit according to an embodiment of the present invention.
  • FIG. 2 is a second schematic structural diagram of an inverter circuit according to an embodiment of the present invention.
  • FIG. 3 is a schematic diagram 1 of an operation principle of an inverter circuit according to an embodiment of the present invention.
  • FIG. 4 is a second schematic diagram of the working principle of the inverter circuit according to an embodiment of the present invention.
  • FIG. 5 is a schematic flow chart of a method for controlling unipolar output of an inverter circuit according to an embodiment of the present invention
  • FIG. 6A and FIG. 6B are schematic diagrams showing output of unipolar modulation of an inverter circuit according to an embodiment of the present invention
  • FIG. 3 is a schematic diagram of the working principle of the inverter circuit in the embodiment of the invention.
  • FIG. 8 is a schematic diagram of a working principle of an inverter circuit according to an embodiment of the present invention
  • FIG. 9 is a schematic flowchart of a method for controlling bipolar output of an inverter circuit according to an embodiment of the present invention
  • Figure 11 is a schematic diagram of the output of the inverter circuit of Figure 10.
  • FIG. 13 is a first schematic diagram of the working principle of reactive power compensation according to an embodiment of the present invention.
  • FIG. 15 is a third schematic diagram of the working principle of reactive power compensation according to an embodiment of the present invention.
  • 16 is a schematic diagram 4 of the working principle of reactive power compensation according to an embodiment of the present invention.
  • 17 is a schematic diagram 5 of a working principle of reactive power compensation according to an embodiment of the present invention.
  • FIG. 18 is a schematic diagram 6 of a working principle of reactive power compensation according to an embodiment of the present invention.
  • FIG. 19 is a schematic diagram 7 of the working principle of reactive power compensation according to an embodiment of the present invention.
  • FIG. 20 is a schematic diagram 8 of the working principle of reactive power compensation according to an embodiment of the present invention.
  • the present invention provides a method for controlling an inverter circuit and an inverter circuit, which can realize soft switching by controlling the inverter circuit, thereby reducing the inverter process. Energy loss in the energy, improve energy conversion efficiency.
  • 1 is a schematic structural diagram 1 of an inverter circuit according to an embodiment of the present invention. As shown in FIG.
  • the inverter circuit includes a DC source DC, an AC circuit, and one or more inverter bridges, and the inverter bridge
  • the first bridge arm and the second bridge arm are included, wherein the output voltage of the DC source DC is U dc , and the first bridge arm and the second bridge arm are disposed in parallel, and are respectively connected between the positive pole and the negative pole of the DC source, specifically
  • the first bridge arm includes a first switching element group 11 and a second switching element group 12 connected in series, wherein the first switching element group 11 includes a first switching tube Q1 and a first diode unit D1, the first two The cathode of the pole unit D1 is connected to the anode of the DC source DC, and the second group of switching elements 12 includes the second switching transistor Q2 and the second diode unit D2, and the cathode of the second diode unit D2 and the DC
  • the second bridge arm also includes a third switching element group 13 and a fourth switching element group 14 connected in series, and the third switching element
  • the inverter bridge includes one or more inverter bridges, and each of the inverter bridges includes a first bridge arm and a second bridge arm disposed in parallel, and further includes an AC circuit, and one end of the AC circuit is connected to the first Between the first switching element group and the second switching element group of one bridge arm, the other end of the alternating current circuit is connected between the third switching element group and the fourth switching element group of the second bridge arm, and the alternating current circuit includes parallel connection The first capacitor and the first AC source; or the first AC load and the first capacitor connected in parallel, and the inverter circuit described above can realize the function of the soft switch through the control of the inverter bridge.
  • the first switch tube and the fourth switch tube may be closed first, and the second switch tube and the third switch tube may be disconnected, so that The inductor current flowing through the first inductor and the second inductor is gradually increased, and the inductor current is divided into a current flowing through the first alternating current source or the first alternating current load, and a current charging the first capacitor;
  • the first switch transistor is turned off; as described above, after the first switch transistor is turned off, the second diode unit First turned on and in a conducting state, and the second switching transistor is connected in parallel with the second diode unit. At this time, the voltage across the second diode unit is equal to the forward voltage drop of the diode.
  • the value is Only
  • the voltage across the second switch can be reduced to 4 ⁇ , and the second switch can be closed to achieve low voltage conduction of the second switch. Further, after the second switch tube is closed, the inductor current flowing through the first inductor and the second inductor is gradually reduced to zero, and the charged first capacitor discharge is generated to flow through the first inductor and the second inductor. The reverse inductor current is gradually increased. In the process, if it is detected that the absolute value of the inductor current or the reverse inductor current flowing through the first inductor and the second inductor is less than the first threshold, the second is disconnected.
  • the first threshold value of the switch tube can be set according to actual needs.
  • the value can be set to 1A, so that the low current turn-off of the second switch tube can be realized.
  • the first diode unit is turned on first, causing the voltage of the two ends of the first diode unit to be close to zero.
  • the first switch tube is closed, that is, the first switch can be realized.
  • the low voltage of the tube is turned on.
  • the low voltage conduction and the low current turn-off of the respective switching tubes can be realized by appropriate control of the inverter circuit, thereby achieving the effect of soft switching.
  • the time for controlling the peak current of the inductor current flowing through the first inductor and the second inductor on the different inverter bridges to be peaked is shifted.
  • the closed state of the fourth switch tube is maintained, and the state of the high frequency switch of the first switch tube and the second switch tube is different from that of the soft switch, and the closed state of the second switch tube is maintained, and the high frequency conversion is performed.
  • the states of the three switch tubes and the fourth switch tube are different from each other.
  • the second switch tube and the third switch tube are closed, so that the first switch tube and the fourth switch tube are disconnected, so as to flow through the first inductor and
  • the inductor current of the second inductor is gradually increased, and the inductor current is divided into a current flowing through the first alternating current source or the first alternating current load, and a current for charging the first capacitor, and the first time period is continuously set.
  • the inductor current flowing through the first inductor and the second inductor is gradually reduced to zero, and the charged first capacitor discharge generates a reverse flow through the first inductor and the second inductor.
  • the inductor current is gradually increased.
  • the inductor current and the reverse inductor current flowing through the first inductor and the second inductor are detected, and if the inductance flowing through the first inductor and the second inductor is detected When the absolute value of the current or reverse inductor current is less than the first threshold, the fourth switching transistor is turned off, and the low current of the fourth switching transistor can be turned off.
  • the third diode unit is in an on state, and the voltage across the third diode unit is small, and the third switch tube is closed at this time, which can be realized.
  • the low voltage of the third switching transistor is turned on.
  • the resistor R may be further disposed in series with the first capacitor C, the resistor R can function to suppress resonance, and the third inductor L3 and the fourth inductor L4 are disposed, and the AC circuit includes In the first AC source, the first capacitor C is connected in parallel with the circuit composed of the third inductor L3, the first alternating current source and the fourth inductor L4; when the alternating current circuit includes the first alternating current load, The first capacitor C is connected in series with a circuit composed of the third inductor L3, the first alternating current load, and the fourth inductor L4.
  • the unipolar modulation output of the inverter circuit can be realized by controlling the state of the switching tube in the inverter bridge of the inverter circuit.
  • Step 101 Close the first switch tube Q1 and The fourth switching transistor Q4 turns on Q1 and Q4, and simultaneously turns off the second switching transistor Q2 and the third switching transistor Q3.
  • Step 101 Close the first switch tube Q1 and The fourth switching transistor Q4 turns on Q1 and Q4, and simultaneously turns off the second switching transistor Q2 and the third switching transistor Q3.
  • the current is output from the positive pole of the direct current source, and passes through Ql, Ll, C, L2, and Q4 are returned to the negative pole of the DC source, and the resistor R can be further set in series with the first capacitor C.
  • the current of the loop charges the first capacitor C, and the set resistor R can suppress the resonance.
  • the inductor current flowing through L1 and L2 gradually increases, and after the current flowing through L1 and L2 reaches a peak for the first time period, Q1 is turned off, the first The time period can be preset according to the output requirement, and the peak value of the inductor current is related to the length of the first time period;
  • Step 102 Close Q2.
  • the inductors L1 and L2 will play a freewheeling action.
  • the D2 will be turned on and turned on.
  • the voltages across D2 and Q2 are small. Close to zero, at this time closing Q2, can achieve the low voltage conduction of Q2, in the process, the inductor continues to flow, the inductor current flowing through L1 and L2 gradually becomes smaller to zero;
  • Step 103 In step 101 and step 102, C is charged. After the inductor current flowing through L1 and L2 gradually becomes smaller to zero, the capacitor is also discharged, and a reverse inductor current flowing through L1 and L2 is generated. During the process, the inductor current and the reverse inductor current flowing through the L 1 and L2 can be detected, and when the inductor current flowing through L1 and L2 is less than the first threshold, Q2 is turned off to realize zero current shutdown of Q2.
  • the first threshold can be set to 1A; Step 104, closing Q1, since the discharge current of the first capacitor C flows through D1 after Q2 is turned off, D1 is in an on state, and both ends of D1 and Q1 are close to zero.
  • the zero voltage conduction of Q1 can be realized, and the process returns to the stage where Q1 and Q4 are simultaneously closed in the above step 101; in the output cycle of the first half of the inverter circuit, the above steps 101 to 104 are repeatedly performed, By controlling the time during which Q1 is closed to off, that is, the first period of time, the peak value of the inductor current flowing through L1 and L2 is controlled, and in the first quarter period, the first period of time can be gradually increased. And in the second During the 1/4 cycle, the first time period is gradually reduced.
  • the inductor current flowing through L1 and L2 can be as shown in FIG. 6A, wherein the sawtooth wave represents the actual inductor current flowing through L1 and L2.
  • the first capacitor C is used, and the first capacitor C can function to smooth the inductor current.
  • the voltage U ab between the two points B and B is switched at a high frequency between U dc and 0.
  • the U ab voltage is within a half cycle of the opening of Q4. It is equivalent to the sinusoidal half-wave, and the high-frequency voltage pulse of U ab is filtered by L1 and L2, and is in phase with the sinusoidal positive half of the AC source to achieve voltage follow-up.
  • the first half of the above cycle is realized by controlling Q1, Q2 and Q4.
  • Q2 is kept closed. status. Specifically, the following steps are included:
  • Step 105 closing the second switch tube Q2 and the third switch tube Q3, turning on the second switch tube Q2 and the third switch tube Q3, and disconnecting the first switch tube Q1 and the fourth switch tube Q4.
  • the current is output from the positive pole of the DC source, and is returned to the cathode of the DC source through Q3, L2, C, L1, and Q2, and the resistor R can be further set in series with the first capacitor C.
  • the current of the loop will be The first capacitor C is charged, and the resistor R is set to suppress the resonance; the circuit further includes a loop, that is, the current is output from the positive pole of the DC source, and returns to the DC source through Q3, L2, AC, L1, and Q2.
  • the negative electrode and the inductors L3 and L4 can be further connected in series at both ends of the AC, and can also suppress the resonance.
  • the inductor current flowing through the first inductor and the second inductor gradually increases, and the inductor current flowing through the first inductor and the second inductor reaches a peak for the first period of time.
  • the second time period can be preset according to the output requirement, and the peak value of the inductor current is related to the length of the first time period; Step 106, closing Q4, at this time, after turning off Q3 in step 105, Since the inductors L2 and L1 will act as a freewheeling, D4 will be turned on after being turned off, and will be in the on state. After the turn-on, the voltage across D4 and Q4 is small, nearly zero, and then turned on. Q4, the zero voltage conduction of Q4 can be realized.
  • step 107 C is charged in the above steps 105 and 106, and is to be flowed through L2 and After the inductor current of L1 gradually becomes smaller to zero, the first capacitor also discharges, generating a reverse inductor current flowing through L1 and L2, in which the inductor current and the reverse inductor current flowing through the L1 and L2 can be performed.
  • Step 108 Q3 is closed, after Q4 is turned off, the discharge current of the first capacitor C flows through D3, and D3 is in conduction state, D3 and Q3 The end is close to zero. At this time, Q3 is closed, and zero voltage conduction of Q3 can be realized.
  • Step 105 to 108 can control the peak value of the inductor current flowing through L1 and L2 by controlling the time from Q3 to OFF.
  • Q3 can be gradually increased from closed to closed. Broken time, and at During the fourth quarter period, the time from the closing to the closing of Q3 is gradually reduced.
  • the specific inductor current flowing through L1 and L2 can be as shown in Fig. 6B.
  • it is realized by separately controlling the closing or opening of one switching tube each time, and also by simultaneously controlling the closing or opening of the two switching tubes. , to achieve bipolar modulation output, but also to achieve the effect of soft switching. Specifically, first, the first switch tube and the fourth switch tube are closed, and the second switch tube and the third switch tube are disconnected, so that the inductor current flowing through the first inductor and the second inductor is gradually increased.
  • the second diode unit and the third diode unit are in an on state, and the second switching tube and the third switching tube are closed to achieve low voltage conduction of the second switching tube and the third switching tube. Further, after the second switch tube and the third switch tube are closed, the inductor current flowing through the first inductor and the second inductor is gradually reduced to zero, the charged first capacitor is discharged, and the DC source is powered.
  • the first diode unit and the fourth diode unit are in an on state, and voltages at both ends thereof are low.
  • the first switch tube and the fourth switch tube are closed, and low voltage conduction of the first switch tube and the fourth switch tube can be realized.
  • the low voltage conduction and the low current shutdown of the switch tube are realized, and the effect of the soft switch can be realized.
  • different inverses are controlled. The time at which the inductor current flowing through the first inductor and the second inductor reaches a peak on the transformer bridge is staggered.
  • the second switch tube and the third switch tube may be closed first, so that the first switch tube and the fourth switch tube are disconnected, so as to flow through the first
  • the inductor current of one inductor and the second inductor gradually increases, and the inductor current is divided into the first alternating current a current of the source or the first alternating current load, and a current for charging the first capacitor, turning off the second switch after the inductor current flowing through the first inductor and the second inductor reaches a peak for a first period of time that is continuously set a tube and the third switch tube;
  • the first diode unit and the fourth diode unit are in an on state, and the first switching tube and the fourth switching tube are closed to achieve low voltage conduction of the first switching tube and the fourth switching tube. Further, after the first switch tube and the fourth switch tube are closed, the inductor current flowing through the first inductor and the second inductor is gradually reduced to zero, the charged first capacitor is discharged, and the DC source is powered.
  • a reverse inductor current flowing through the first inductor and the second inductor is gradually increased, and in the process, an inductor current and a reverse inductor current flowing through the first inductor and the second inductor are detected, to be detected Disabling the first switch tube and the fourth switch tube when an absolute value of an inductor current or a reverse inductor current flowing through the first inductor and the second inductor is less than a first threshold, implementing the first switch The low current of the tube and the fourth switching tube is turned off.
  • the second diode unit and the third diode unit are in an on state, and the second switch tube and the door are closed.
  • the third switch tube realizes low voltage conduction of the second switch tube and the third switch tube.
  • the low voltage conduction and the low current shutdown of the switch tube are realized, and the effect of the soft switch can be realized.
  • different inverses are controlled.
  • the time at which the inductor current flowing through the first inductor and the second inductor reaches a peak on the transformer bridge is staggered.
  • the bipolar modulation output mode is different from the unipolar modulation in that the bipolar modulation causes Ql and Q4 to be simultaneously closed or disconnected, and Q3 and Q2 are also disconnected at the same time.
  • the modulation step can refer to Figure 9.
  • the bipolar modulation output may include the following steps: Step 201: Close the first switch tube Q1 and the fourth switch tube Q4, and make the second switch tube Q2 and the third switch tube Q3 Disconnected, in this case, in a loop, the current is output from the positive terminal of the DC source, and will return to the negative pole of the DC source through Ql, L l, C, R, L2, and Q4.
  • the current of the loop will charge the first capacitor C.
  • the set resistor R can suppress the resonance; it also includes a loop, that is, the current is output from the positive pole of the DC source, and will return to the cathode of the DC source through Ql, Ll, L3, AC, L4, L2, and Q4.
  • the third inductor L3 and the fourth inductor L4 can also function to suppress resonance.
  • the electricity flowing through the inductors L1 and L2 at the beginning of the closing of Q1 and Q4 The sense current will gradually increase, and after the current flowing through L1 and L2 reaches a peak for the first time period, Q1 and Q4 are turned off, and the first time period can be preset according to the output demand, and the inductor current thereof
  • the peak value is related to the length of the first time period;
  • Step 202 closing the second switch tube Q2 and the third switch tube Q3, at this time, the inductors L1 and L2 will play a freewheeling action, after turning off Q1, D2 and D3 will It is turned on and is in the on state.
  • Step 203 C is charged in the above steps 201 and 202, and flows through L1 and L2. After the current gradually decreases to zero, the first capacitor will discharge, and the DC source DC supply will generate a reverse inductor current flowing through the inductors L1 and L2. In this process, the inductor current flowing through L1 and L2 can be controlled to be small. Therefore, the inductor current flowing through L1 and L2 can be measured and to be detected. L1 and L2 flowing through the current is less than a first threshold Q2 turns off and Q3, Q2 and Q3 low current shutdown;
  • Step 204 closing Q1 and Q4, since after Q2 and Q3 are turned off, the discharge current of the first capacitor C flows through D1 and D4, and D1 and D4 are turned on at this time, and both ends of Q1 and Q4 are close to zero, so The low voltage conduction of Q1 and Q4 can be realized.
  • the phases of Q1 and Q4 being simultaneously turned on are continued; in the output cycle of the first half of the alternating current, the above steps 201 to 204 are repeatedly performed, and Q1 can be controlled.
  • the first time period controls the peak value of the inductor current flowing through L1 and L2, and in the first quarter period, the first time period can be gradually increased, that is, Increasing the time from Q1 and Q4 from closed to open, and gradually decreasing the first time period described above during the second 1/4 cycle, ie reducing the time from Q1 and Q4 from closed to open.
  • Step 205 Close the second switch tube Q2 and the third switch tube Q3, and disconnect the first switch tube Q1 and the fourth switch tube Q4, and then in a loop
  • the current is output from the positive output of the DC source and will return to the negative pole of the DC source through Q3, L2, R, C, L1, and Q2.
  • the current of the loop charges the first capacitor C, and the set resistor R can Suppresses the effect of resonance; it also includes a loop, that is, the current is output from the positive pole of the DC source, passing through Q3, L2, L4, AC, L3, Ll, Q2 returns to the negative pole of the DC source, and the third inductor L3 and the fourth inductor L4 can also suppress the resonance.
  • the inductor current flowing through the inductors L1 and L2 is gradually increased, and Q2 and Q3 are turned off after the first period of time continues.
  • the first period of time can be preset according to actual needs, and the peak value of the inductor current Related to the length of the first time period;
  • Step 206 closing Q1 and Q4.
  • the inductor continues to flow through L2 and The current of L1 gradually becomes smaller to zero;
  • Step 207 Charging C in the above steps 205 and 206, after the current flowing through L2 and L1 gradually becomes smaller to zero, the first capacitor is discharged, and the DC source DC is supplied with power, so that the flow flows through the inductor L2. And the inductor current in the L1 direction, in this process, the inductor current flowing through L1 and L2 can be controlled to be small, so the inductor current flowing through L1 and L2 can be measured, and the inductor current flowing through L1 and L2 is detected to be smaller than the Q1 and Q4 are disconnected for one threshold to achieve low current shutdown of Q1 and Q4;
  • Step 208 Close Q2 and Q3. After Q1 and Q4 are turned off, the discharge current of C flows through D2 and D3, and D2 and D3 are turned on. At this time, both ends of Q2 and Q3 are close to zero, and Q2 and Q3 are closed at this time. The low voltage conduction of Q2 and Q3 can be realized, and the process returns to the same stage in step 201 where Q2 and Q3 are simultaneously turned on;
  • Repeating the above steps 205 to 208 can control the peak value of the inductor current flowing through L1 and L2 by controlling the first time period, that is, the time from the closing to the opening of Q2 and Q3, in the first quarter cycle
  • the time from Q2 and Q3 from closed to open can be gradually increased, and in the second 1/4 cycle, the time from Q2 and Q3 from closed to open is gradually reduced.
  • the inverter circuit provided in the above embodiments of the present invention can realize the effect of the switch tube as a soft switch by the DC source power supply.
  • the AC circuit includes an AC source
  • the AC source can also be powered.
  • Soft switching is also achieved by controlling any inverter bridge Effect. Specifically, referring to FIG.
  • the second switch tube Q2 and the fourth switch tube Q4 are first closed, and the first switch tube Q1 and the third switch tube Q3 are disconnected, and the first AC source AC is powered to pass through the first
  • the inductor current of the inductor 1 and the second inductor L2 gradually increases, and after the inductor current flowing through the first inductor L1 and the second inductor L2 reaches a peak for the first time period that is continuously set, the second switch transistor is turned off.
  • Q2 the magnitude of the peak value of the inductor current flowing through the first inductor L1 and the second inductor L2 is related to the length of the set first time period;
  • the first diode unit D1 After the second switch tube Q2 is turned off, the first diode unit D1 is in an on state, and the voltage at both ends thereof is low. At this time, the first switch tube Q1 is closed, and the low voltage conduction of the first switch tube Q1 can be realized. .
  • the inductor current flowing through the first inductor L1 and the second inductor L2 gradually decreases to zero, and the DC source DC power supply flows through the first inductor L1 and the second inductor L2.
  • the first switching transistor Q1 implements a low current shutdown of the first switching transistor Q1.
  • the second diode unit D2 is in an on state, and the voltages at both ends are low, and the second switching transistor Q2 is closed to realize the low of the second switching transistor Q2. The voltage is turned on.
  • the soft switching performance of the switch tube can be realized, and further, the above steps can be repeatedly performed in the first half output period of the inverter circuit, and for the 1/4 cycle, the first step is sequentially increased.
  • the length of the time period, for the second quarter cycle decreases the length of the first time period in turn, so that the peak value of the current flowing through L1 and L2 first increases and then decreases.
  • the following method can be used for control. Referring to FIG. 16, first, the second switch tube Q2 and the fourth switch tube Q4 are closed, and the first switch tube Q1 and the third unit are closed. The switch tube Q3 is disconnected, the first AC source AC is powered, and the inductor current flowing through the first inductor L1 and the second inductor L2 is gradually increased, and the first period of time is continuously set to flow through the first inductor L1 and the second inductor.
  • the fourth switch tube Q4 After the inductance circuit of L2 reaches a peak, the fourth switch tube Q4 is turned off; After the fourth switch tube Q4 is disconnected, the third diode unit D3 is in an on state, and at this time, the third switch tube Q3 is closed to achieve low voltage conduction of the third switch tube Q3. Further, in the foregoing embodiment, after the third switch transistor Q3 is closed, the inductor current flowing through the first inductor L1 and the second inductor L2 is gradually reduced to zero, and the DC source power is generated to flow through the first inductor.
  • the third switch transistor Q3 is turned off, and the low current turn-off of the third switch transistor Q3 is realized.
  • the fourth diode unit D4 is in an on state, and the voltage at both ends is low, and the fourth switching transistor Q4 is closed at this time, so that the low voltage of the fourth switching transistor Q4 can be realized. Turn on.
  • an inverter bridge is taken as an example, and how to implement the soft switch is described.
  • more than two inverter bridges may be included in the inverter circuit, as shown in FIG.
  • the inverter circuit further includes another inverter bridge, that is, a third bridge arm and a fourth bridge arm, the third bridge, based on the first inverter bridge.
  • the arm is respectively connected to the positive pole and the negative pole of the direct current source, the third bridge arm and the first bridge arm are arranged in parallel; the fourth bridge arm is respectively connected to the positive pole and the negative pole of the direct current source, and the fourth bridge arm and The second bridge arms are arranged in parallel.
  • the third bridge arm includes a fifth switching element group 15 and a sixth switching element group 16 connected in series, and the fifth switching element group 15 includes a fifth switching tube Q12 and a fifth diode unit D12 connected in parallel, the The cathode of the five diode unit D12 is connected to the anode of the DC source, and the sixth switch
  • the component group 16 includes a sixth switching transistor and a Q22 sixth transistor unit D22, the negative pole of the sixth pole unit D22 is connected to the anode of the DC source; and the fourth bridge arm includes a seventh switch connected in series.
  • the seventh switching element group 21 includes a seventh switching transistor Q32 and a seventh diode unit D32 connected in parallel, a negative pole of the seventh diode unit D32 and the The anode of the DC source is connected
  • the eighth switching element group 22 includes an eighth switching transistor Q42 and an eighth diode unit D42 connected in parallel
  • a cathode of the eighth diode unit D42 is connected to the anode of the DC source.
  • one end is connected between the fifth switching element group 15 and the sixth switching element group 16 of the third bridge arm through the inductor L5, and the other end is connected to the seventh switching element group of the fourth bridge arm through the inductor L6.
  • the above-described inductances L5 and L6 have the same action and connection relationship as the first inductance L1 and the second inductance L2 in the first inverter bridge.
  • Each of the inverter bridges of the inverter circuit shown in FIG. 10 is controlled by the control method provided in the above embodiment, and the inductor current flowing through the first inductor and the second inductor on different inverter bridges is controlled to reach a peak value.
  • the time is staggered, that is, for the technical solution of the two inverter bridges in Fig. 10, the current flowing through L1 and L2 is staggered with the time at which the current flowing through L5 and L6 reaches a peak.
  • the upper half can be regarded as the current flowing through L1 and L2
  • the lower half can be regarded as the current flowing through L5 and L6, the currents of the two are superimposed, and the time to reach the peak is staggered.
  • the switching tube and the diode unit in each of the switching element groups may be independently arranged electronic components, that is, the first switching transistor Q1 and the first diode unit D1 connected in parallel are independent.
  • the set electronic component, the second switch tube Q2 and the second diode unit D2 are independently arranged electronic components, and the third switch transistor Q3 and the third diode unit D3 are independently arranged electronic components, and
  • the fourth switching transistor Q4 and the fourth diode unit D4 are independently arranged electronic components.
  • the first diode unit D1, the second diode unit D2, the third diode unit D3 or the fourth diode unit D4 may be It can be configured in any of the following ways: a diode set separately or a diode group consisting of two or more diodes connected in series.
  • the first switch tube Q1, the second switch tube Q2, the third switch tube Q3, and the fourth switch tube Q4 may be any of the following switches: MOSFET, triode, IGBT Tube or thyristor.
  • a diode is parasitic in the MOSFET
  • the MOSFET itself includes the functions of a switching transistor and a diode, so the parasitic diode can be used as a diode unit, and the first switching element group and the second
  • the MOSFET can be directly used as the switching element group, the third switching element group, and the fourth switching element group.
  • a first switching transistor Q1 and the first diode unit D1, the second switching transistor Q2 and the second diode unit D2, the third switching transistor Q3 and the third diode unit D3, and the fourth switching transistor Q4 and the fourth diode unit D4 are all components in the corresponding MOSFET.
  • the technical solution provided by the above embodiments of the present invention can also implement the power compensation function.
  • the specific reactive power compensation can be as shown in FIG. 12, in the time periods T1 and T3, wherein the output voltage and current of the voltage source AC are in phase; Like the pure active output, only the current setting will be slightly different, which can be achieved by controlling the conduction time. In the T2 and T3 time periods, the output voltage and current of the voltage source AC can be controlled to be inverted. Power compensation.
  • the circuit can be as shown in FIGS. 13, 14, 15, and 16, respectively corresponding to the time periods T1, ⁇ 2, ⁇ 3, and ⁇ 4, and the working principle thereof will be described below: a), corresponding to the time period T1 of FIG. 12, as shown in FIG.
  • the current direction can be controlled from L1, through the first voltage source AC to L2, and the positive pole of the first voltage source AC is controlled on the L1 side, the first voltage The negative pole of the source is on the L2 side.
  • Q4 is kept in the on state, Q3 is disconnected, Q1 is used as the main control switch, and the high frequency of Q1 and Q2 is alternately operated.
  • the first voltage source in the T1 period The output voltage is in phase with the current direction, which can achieve normal pure active output;
  • the anode of the first voltage source AC is on the L1 side
  • the cathode of the first voltage source is on the L2 side
  • the first voltage source is powered by AC
  • the current is from L2, through voltage source AC to L1
  • one of the current loops is L3, Ll, Q2, Q4 (D4), L2, L4 and AC source
  • the other loop is C, Ll, Q2, Q4
  • the positive pole of the voltage source is on the L2 side
  • the negative pole of the voltage source is on the L1 side
  • the current is from L1
  • the voltage source is connected to L2, and Q2 and Q4 can be closed first, one of which is a current loop.
  • L4, L2, Q4, Q2 (D2), L2, L3 and AC source the other loop is composed of C, R, L2, Q4 ( D4 ), Ll, C.
  • T4 the output voltage of the current source And the current is reversed, the utility can realize the reactive power to the inverter circuit, during the period Q2—straight In the conducting state, Q3 and Q4 operate alternately high frequency, Q4 as a main Controls.
  • reactive power compensation can also be implemented, as shown in FIG. 17, FIG. 18, FIG. 19 and FIG.
  • Q1 and Q4 are first closed, and Q3 and Q2 are turned off.
  • the current is from L1, through the first voltage source AC to L2, and the anode of the first voltage source is located on the L1 side.
  • the negative pole of a voltage source is located on the L2 side.
  • Q1 and Q4 are simultaneously turned off, and Q3 and Q3 are simultaneously closed, so that the output voltage and current are in phase during the T1 period, and pure active output can be realized. ;
  • Q1 and Q4 are first closed, and Q3 and Q2 are turned off.
  • the current is from L1, through the first voltage source AC to L2, and the anode of the first voltage source is located on the L1 side.
  • the negative pole of a voltage source is located on the L2-side.

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Abstract

一种逆变器电路及逆变器电路的控制方法。逆变器电路包括直流源(DC)、交流电路和一个以上的逆变电桥。逆变电桥包括第一桥臂和第二桥臂。第一桥臂分别连接直流源的正极和负极,第二桥臂分别连接直流源的正极和负极。第一桥臂和第二桥臂并联。第一桥臂包括串联的第一开关元件组(11)和第二开关元件组(12)。第二桥臂包括串联的第三开关元件组(13)和第四开关元件组(14)。交流电路包括并联的第一交流源(AC)和第一电容(C),或者包括并联的第一交流负载(AC)和第一电容(C)。该逆变器电路及其控制方法能够实现软开关。

Description

逆变器电路及逆变器电路的控制方法
技术领域 本发明涉及电力电子技术领域, 尤其涉及一种逆变器电路及逆变器电 路的控制方法。
背景技术 逆变器电路是一种将直流电转换成交流电的电路,也是一种将直流能量 转换为交流能量的电路, 其包括输出交流电压形式, 如不间断电源; 以及 跟随外部交流电压而输出交流电流的形式, 如太阳能并网逆变器, 风力并 网发电机等。 现有的逆变器电路, 在使用过程中为实现将直流能量转换为交流能量, 需要对逆变器电路中的开关元件进行高频切换。 因开关元件在开通和关断 的切换过程中产生的开关损耗不同, 可以将开关分为软开关和硬开关, 具 体的, 在开关开通过程中, 开关两端电压下降与流过开关的电流上升在时 间上出现交叠而产生的功耗为开通损耗; 或在开关关断过程中, 流过开关 的电流下降与开关两端电压上升在时间上出现交叠而产生的功耗为关断损 耗。 无法避免开关损耗的开关叫做硬开关, 可以避免开关损耗的开关叫做 软开关。
换为交流电的过程中功耗过大, 能量转换效率低。
发明内容 本发明实施例提供一种逆变器电路及逆变器电路的控制方法, 用以在 逆变器电路中实现软开关。 本发明实施例提供了一种逆变器电路, 包括直流源、 交流电路和一个 以上的逆变电桥, 所述逆变电桥包括第一桥臂和第二桥臂, 所述第一桥臂 分别连接所述直流源的正极和负极, 所述第二桥臂分别连接所述直流源的 正极和负极, 所述第一桥臂和所述第二桥臂并联设置; 所述第一桥臂包括串联的第一开关元件组和第二开关元件组, 所述第 一开关元件组包括并联的第一开关管和第一二极管单元, 所述第一二极管 单元的负极与所述直流源的正极连接, 所述第二开关元件组包括并联的第 二开关管和第二二极管单元, 所述第二二极管单元的负极与所述直流源的 正极连接;
所述第二桥臂包括串联的第三开关元件组和第四开关元件组, 所述第 三开关元件组包括并联的第三开关管和第三二极管单元, 所述第三二极管 单元的负极与所述直流源的正极连接, 所述第四开关元件组包括并联的第 四开关管和第四二极管单元, 所述第四二极管单元的负极与所述直流源的 正极连接; 所述交流电路包括并联的第一交流源和第一电容,或包括并联的第一交流负 载和第一电容, 所述交流电路的一端通过第一电感连接在所述第一桥臂的第一开 关元件组和第二开关元件组之间, 所述交流电路的另一端通过第二电感连接在所 述第二桥臂的第三开关元件组和第四开关元件组之间。 本发明实施例提供了一种针对上述逆变器电路的控制方法, 对于所述 逆变器电路中的任一逆变电桥, 根据如下的方法进行控制: 闭合第一开关管和第四开关管, 令第二开关管和第三开关管断开, 以 使流经第一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经 第一交流源或第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时间段使所述流经第一电感和第二电感的电感电流 达到峰值后, 断开所述第一开关管;
第二二极管单元处于导通状态, 闭合第二开关管, 实现第二开关管的 低电压导通。 本发明实施例提供了另一种针对上述逆变器电路的控制方法, 对于所 述逆变器电路中的任一逆变电桥, 根据如下的方法进行控制: 闭合第一开关管和第四开关管, 令第二开关管和第三开关管断开, 以 使流经第一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经 第一交流源或第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时间段使流经第一电感和第二电感的电感电流达到 峰值后, 断开所述第一开关管和所述第四开关管;
第二二极管单元和第三二极管单元处于导通状态, 闭合第二开关管和 第三开关管, 以实现第二开关管和第三开关管的低电压导通。 本发明实施例提供了再一种针对上述逆变器电路的控制方法, 在交流电路包括第一交流源时,对于所述逆变器电路中的任一逆变电桥, 根据如下的方法进行控制:
闭合第二开关管和第四开关管, 令第一开关管和第三开关管断开, 第 一交流源供电, 以使流经第一电感和第二电感的电感电流逐渐增大; 在持续设定的第一时间段使流经第一电感和第二电感的电感电流达到 峰值后, 断开所述第二开关管;
第一二极管单元处于导通状态, 闭合第一开关管, 实现第一开关管的 低电压导通。
本发明实施例提供的逆变器电路及逆变器电路的控制方法, 通过在各 个逆变电桥的开关元件组中设置并联的开关管和二极管单元, 以及在交流 电路中设置并联的电压源和第一电容, 或者是并联的电容和交流负载, 并 通过适当的控制方法, 使得开关管可以在低电压状态下关断, 并在低电流 状态下导通, 从而使得开关管具有软开关的功能, 能够有效减少在开光开 断过程中的能量损耗。 附图说明 为了更清楚地说明本发明实施例或现有技术中的技术方案, 下面将对 实施例或现有技术描述中所需要使用的附图作一简单地介绍, 显而易见地, 下面描述中的附图是本发明的一些实施例, 对于本领域普通技术人员来讲, 在不付出创造性劳动性的前提下, 还可以根据这些附图获得其他的附图。 图 1为本发明实施例中逆变器电路的结构示意图一;
图 2为本发明实施例中逆变器电路的结构示意图二;
图 3为本发明实施例中逆变器电路的工作原理示意图一;
图 4为本发明实施例中逆变器电路的工作原理示意图二;
图 5为本发明实施例中逆变器电路单极性输出的控制方法流程示意图; 图 6 A和图 6B为本发明实施例中逆变器电路单极性调制的输出示意图; 图 7为本发明实施例中逆变器电路的工作原理示意图三;
图 8为本发明实施例中逆变器电路的工作原理示意图四; 图 9为本发明实施例中逆变器电路双极性输出的控制方法流程示意图; 图 10为本发明实施例中逆变器电路的结构示意图三;
图 11为图 10所述的逆变器电路的输出示意图;
图 12为本发明实施例中无功补偿的电压、 电流对比图;
图 13为本发明实施例中无功补偿的工作原理示意图一;
图 14为本发明实施例中无功补偿的工作原理示意图二;
图 15为本发明实施例中无功补偿的工作原理示意图三;
图 16为本发明实施例中无功补偿的工作原理示意图四;
图 17为本发明实施例中无功补偿的工作原理示意图五;
图 18为本发明实施例中无功补偿的工作原理示意图六;
图 19为本发明实施例中无功补偿的工作原理示意图七;
图 20为本发明实施例中无功补偿的工作原理示意图八。
具体实施方式 为使本发明实施例的目的、 技术方案和优点更加清楚, 下面将结合本 发明实施例中的附图, 对本发明实施例中的技术方案进行清楚、 完整地描 述, 显然, 所描述的实施例是本发明一部分实施例, 而不是全部的实施例。 基于本发明中的实施例, 本领域普通技术人员在没有作出创造性劳动前提 下所获得的所有其他实施例, 都属于本发明保护的范围。
电转换为交流电的过程中损耗过大的问题, 本发明实施例提供了一种逆变 电路及逆变电路的控制方法, 其能够通过对逆变电路的控制实现软开关, 进而降低逆变过程中的能量损耗, 提高能量转换效率。 图 1为本发明实施例中逆变器电路的结构示意图一, 如图 1所示, 该 逆变器电路包括直流源 DC、 交流电路和一个以上的逆变电桥, 所述逆变电 桥包括第一桥臂和第二桥臂, 其中, 该直流源 DC的输出电压是 Udc, 第一 桥臂和第二桥臂并联设置, 都分别连接在直流源的正极和负极之间, 具体 的, 上述的第一桥臂包括串联的第一开关元件组 11和第二开关元件组 12, 其中第一开关元件组 11包括第一开关管 Q1和第一二极管单元 D1 ,第一二 极管单元 D1的负极与所述直流源 DC的正极连接, 第二开关元件组 12包 括第二开关管 Q2和第二二极管单元 D2, 第二二极管单元 D2的负极与所 述直流源 DC的正极连接; 第二桥臂也包括串联的第三开关元件组 13和第 四开关元件组 14, 第三开关元件组 13包括第三开关管 Q3和第三二极管单 元 D3 , 第三二极管单元 D3的负极与所述直流源 DC的正极连接, 第四开 关元件组 14 包括第四开关管 Q4和第四二极管单元 D4, 第四二极管单元 D4的负极与所述直流源 DC的正极连接; 上述交流电路包括并联的第一交流 源 AC和第一电容 C,或包括并联的第一交流负载 AC和第一电容 C,上述交流电 路的一端通过第一电感连接在第一桥臂的第一开关元件组 11与第二开关元件 组 12之间, 所述交流电路的另一端通过第二电感连接在第二桥臂的第三开关 元件组 13与第四开关元件组 14之间。 本发明上述实施例中, 包括一个或多个逆变电桥, 各逆变电桥包括并 联设置的第一桥臂和第二桥臂, 另外还包括交流电路, 该交流电路的一端连 接在第一桥臂的第一开关元件组与第二开关元件组之间, 交流电路的另一端 连接在第二桥臂的第三开关元件组与第四开关元件组之间, 且该交流电路 包括并联的第一电容和第一交流源; 或者包括并联的第一交流负载和第一电容, 上述的逆变器电路, 可以通 逆变电桥进 当的控制实现软开关的功能。 例如在对上述的逆变器电路的任一逆变电桥的控制过程中, 可以先闭合第一 开关管和第四开关管, 并令第二开关管和第三开关管断开, 以使流经第一电 感和第二电感的电感电流逐渐增大, 上述电感电流分为流经第一交流源或 第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时 间段后使所述流经第一电感和第二电感的电感电流达到峰值后, 断开第一 开关管; 如上所述的, 在第一开关管断开后, 第二二极管单元先导通并处 于导通状态, 而第二开关管与第二二极管单元并联, 此时第二二极管单元 两端的电压等于二极管的正向导通压降, 对于硅二极管而言, 其值仅为
0.3V~0.7V之间, 因此可以第二开关管两端的电压降至 4艮低, 闭合第二开关 管实现第二开关管的低电压导通。 进一步的, 在闭合上述的第二开关管后, 流经第一电感和第二电感的电 感电流逐渐减小到零, 被充电的第一电容放电产生流经所述第一电感和第 二电感的反向电感电流并逐渐增大, 在此过程中, 若检测到流经第一电感 和第二电感的电感电流或反向电感电流的绝对值小于第一门限时, 断开所 述第二开关管, 上述的第一门限值可以根据实际的需要设置, 例如可以设 置其值为 1A, 也就可以实现第二开关管的低电流关断。 另外, 在断开第二开关管后, 使得第一二极管单元先导通, 导致第一二 极管单元的两端电压接近为零, 此时闭合第一开关管, 即可以实现第一开 关管的低电压导通。 本发明上述实施例中, 通过对逆变器电路的适当控制, 可以实现各个 开关管的低电压导通和低电流关断, 从而达到软开关的效果。 对于逆变器 电路中的逆变电桥为两个以上时, 控制不同逆变电桥上流经第一电感和第 二电感的电感电流达到峰值的时间错开。 与上述技术方案中, 保持第四开关管的闭合状态, 高频转换第一开关 管和第二开关管的状态实现软开关不同, 还可以是保持第二开关管的闭合 状态, 高频转换第三开关管和第四开关管的状态实现软开关不同, 具体的, 闭合第二开关管和第三开关管, 令第一开关管和第四开关管断开, 以使流 经第一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经第一 交流源或第一交流负载的电流, 以及对第一电容充电的电流, 在持续设定 的第一时间段使所述流经第一电感和第二电感的电感电流达到峰值后, 断 开所述第三开关管; 在断开第三开关管后, 第四二极管单元首先导通, 并 处于导通状态, 此时闭合第四开关管, 能够实现第四开关管的低电压导通。 进一步的, 闭合第四开关管后, 流经第一电感和第二电感的电感电流 逐渐减小到零, 被充电的第一电容放电产生流经所述第一电感和第二电感 的反向电感电流并逐渐增大, 在此过程中, 对流经所述第一电感和第二电 感的电感电流和反向电感电流进行检测 , 若检测到流经所述第一电感和第 二电感的电感电流或反向电感电流的绝对值小于第一门限时, 断开所述第 四开关管, 能够所述第四开关管的低电流关断。
另外, 在断开所述第四开关管后, 所述第三二极管单元处于导通状态, 第三二极管单元两端的电压较小, 此时闭合所述第三开关管, 能够实现所 述第三开关管的低电压导通。 另外, 本发明上述实施例中, 对于逆变器电路中的逆变电桥为两个以 上时, 控制不同逆变电桥上流经第一电感和第二电感的电感电流达到峰值 的时间错开。 另外, 如图 2所示, 可以进一步的设置电阻 R与第一电容 C串联, 该 电阻 R能够起到抑制谐振的作用, 以及设置第三电感 L3和第四电感 L4, 在所述交流电路包括第一交流源时, 所述第一电容 C与所述第三电感 L3、 所述第一交流源和第四电感 L4组成的电路并联; 在所述交流电路包括第一 交流负载时, 所述第一电容 C与所述第三电感 L3、 所述第一交流负载和第 四电感 L4组成的电路串联。 具体的, 上述通过对逆变器电路的逆变电桥中开关管的状态控制, 可 以实现逆变器电路的单极性调制输出。 该单极性调制输出的工作原理图可 如图 3和图 4所示, 其一个输出周期内的步骤流程可以如图 5所示, 包括 如下的步骤: 步骤 101、 闭合第一开关管 Q1和第四开关管 Q4, 令 Q1和 Q4导通, 同时令第二开关管 Q2和第三开关管 Q3断开, 此时在一个回路中, 电流从 直流源的正极输出, 会经过 Ql、 Ll、 C、 L2、 Q4回到直流源的负极, 并可 以进一步的设置电阻 R与第一电容 C串联, 该回路的电流会对第一电容 C 充电, 其设置的电阻 R能够起到抑制谐振的作用; 另外还包括一个回路, 即电流从直流源的正极输出, 会经过 Ql、 Ll、 AC、 L2、 Q4回到直流源的 负极,并可以进一步的在 AC的两端分别串联上第三电感 L3和第四电感 L4 , 也能够起到抑制谐振的作用。 本步骤中, 在 Q1和 Q4闭合初期, 流经 L1 和 L2的电感电流逐渐增大,在持续第一时间段使所述流经 L1和 L2的电流 达到峰值后, 断开 Q1 , 该第一时间段可以根据输出需要而预先设定, 其电 感电流的峰值与第一时间段的长短相关;
步骤 102、 闭合 Q2, 此时,电感 L1和 L2会起到续流作用, 在关断 Q1 后 D2会随之导通, 并处于导通状态, 导通后 D2和 Q2两端的电压很小, 接近为零, 此时闭合 Q2, 能够实现 Q2的低电压导通, 在此过程中, 电感 续流,流经 L1和 L2的电感电流逐渐变小到零;
步骤 103、 在上述步骤 101和步骤 102中都会对 C充电, 待流经 L1和 L2的电感电流逐渐变小到零后, 电容也会放电, 产生流经 L1和 L2的反向 电感电流, 此过程中可以对流经所述 L 1和 L2的电感电流和反向电感电流 进行检测, 待检测到流经 L1和 L2的电感电流小于第一门限时断开 Q2, 实 现 Q2的零电流关断, 该第一门限可以设置为 1A; 步骤 104、 闭合 Q1 , 由于在 Q2关断后, 第一电容 C的放电电流流经 D1 , 此时 D1处于导通状态, D1和 Q1两端的接近为零, 此时闭合 Q1 , 即 可实现 Q1的零电压导通,回到上述步骤 101中 Q1和 Q4同时闭合的阶段; 在前半个逆变器电路输出周期, 重复执行上述的步骤 101〜步骤 104, 可以通过控制 Q1由闭合到关断的时间,即第一时间段来控制流经 L1和 L2 的电感电流的峰值,在第一个 1/4周期内,可以逐渐增大上述的第一时间段, 并在第二个 1/4周期内, 逐渐减小上述的第一时间段, 具体的, 流经 L1和 L2的电感电流可如图 6A所示, 其中的锯齿波表示流经 L1和 L2的实际电 感电流变化情况, 而上述实施例中使用了第一电容 C, 该第一电容 C能够 起到平滑电感电流的作用。
在上述过程中, 在八、 B两点间的电压 Uab在 Udc和 0之间高频切换, 通过对 Q1闭合和关断时间的控制, 使得 Uab电压在 Q4开通的半个周期内 与正弦正半波在面积上等效, Uab的高频电压脉冲经过 L1和 L2的滤波作用 , 与交流源的正弦正半波同相位, 实现电压跟随。 上述前半个周期,是通过对 Ql、 Q2和 Q4的控制实现,在后半个周期, 如图 4所示, 可以通过对 Q2、 Q3和 Q4的控制实现, 此过程中, Q2—直 保持闭合状态。 具体的包括如下的步骤:
步骤 105、 闭合第二开关管 Q2和第三开关管 Q3 , 令第二开关管 Q2和 第三开关管 Q3导通, 并令第一开关管 Q1和第四开关管 Q4断开, 此时在 一个回路中, 电流从直流源的正极输出, 会经过 Q3、 L2、 C、 Ll、 Q2回到 直流源的负极, 并可以进一步的设置电阻 R与第一电容 C串联, 该回路的 电流会对第一电容 C充电, 其设置的电阻 R能够起到抑制谐振的作用; 另 外还包括一个回路, 即电流从直流源的正极输出, 会经过 Q3、 L2、 AC、 Ll、 Q2回到直流源的负极, 并可以进一步的在 AC的两端分别串联上电感 L3和 L4, 也能够起到抑制谐振的作用。 本步骤中, 在 Q2和 Q3闭合初期, 流经第一电感和第二电感的电感电流逐渐增大, 在持续第一时间段使所述 流经第一电感和第二电感的电感电流达到峰值后关断 Q3 , 该第二时间段可 以根据输出需要而预先设定, 其电感电流的峰值与第一时间段的长短相关; 步骤 106、 闭合 Q4, 此时, 步骤 105 中关断 Q3后, 由于电感 L2和 L1会起到续流作用, 在关断 Q1后 D4会随之导通, 并处于导通状态, 导通 后 D4和 Q4两端的电压 4艮小, 接近为零, 此时开通 Q4, 能够实现 Q4的零 电压导通, 在此过程中, 流经 L1和 L2的电流逐渐变小到零; 步骤 107、 在上述步骤 105和步骤 106中都会对 C充电, 待流经 L2和 L1的电感电流逐渐变小到零后, 第一电容也会放电, 产生流经 L1和 L2的 反向电感电流, 此过程中可以对流经所述 L1和 L2的电感电流和反向电感 电流进行检测, 待检测到流经 L1和 L2的电流小于第一门限时断开 Q4, 实 现 Q4的零电流关断; 步骤 108、 闭合 Q3 , 在 Q4关断后, 第一电容 C的放电电流流经 D3 , 此时 D3处于导通状态, D3和 Q3两端的接近为零, 此时闭合 Q3 , 即可实 现 Q3的零电压导通, 回到上述步骤 105中 Q3和 Q2同时导通的阶段; 在后半个逆变器电路输出周期内, 重复执行上述的步骤 105〜步骤 108, 可以通过控制 Q3由闭合到关断的时间来控制流经 L1和 L2的电感电流的 峰值, 在第三个 1/4周期内, 可以逐渐增大 Q3由闭合到关断的时间, 并在 第四个 1/4周期内, 逐渐减小 Q3由闭合到关断的时间, 具体的流经 L1和 L2的电感电流可如图 6B所示。 上述在实现逆变器电路单极性输出的过程中, 是通过每次单独控制一 个开关管的闭合或断开实现的, 另外, 还可以通过同时控制两个开关管的 闭合或断开的方式, 实现双极性调制输出, 同时也能够实现软开关的效果。 具体的, 首先, 闭合第一开关管和第四开关管, 令第二开关管和第三开关 管断开, 以使流经第一电感和第二电感的电感电流逐渐增大, 该电感电流 分为流经第一交流源或第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时间段使流经第一电感和第二电感的电感电流达到 峰值后, 断开所述第一开关管和所述第四开关管;
第二二极管单元和第三二极管单元处于导通状态, 闭合第二开关管和 第三开关管, 以实现第二开关管和第三开关管的低电压导通。 进一步的, 在闭合所述第二开关管和所述第三开关管后, 流经第一电 感和第二电感的电感电流逐渐减小到零, 被充电的第一电容放电, 以及直 流源供电产生流经所述第一电感和第二电感的反向电感电流并逐渐增大, 在此过程中, 若检测到流经所述第一电感和第二电感的电感电流或反向电 感电流的绝对值小于第一门限时断开所述第二开关管和所述第三开关管, 实现所述第二开关管和所述第三开关管的低电流关断。
另外, 在断开所述第二开关管和所述第三开关管后, 所述第一二极管 单元和所述第四二极管单元处于导通状态, 其两端的电压都很低, 此时闭 合所述第一开关管和所述第四开关管, 能够实现所述第一开关管和所述第 四开关管的低电压导通。 本发明上述实施例中实现了开关管的低电压导通和低电流关断, 能够 实现软开关的效果, 对于所述逆变器电路中的逆变电桥为两个以上时, 控 制不同逆变电桥上流经第一电感和第二电感的电感电流达到峰值的时间错 开。
与上述实施例中先闭合第一开关管和第四开关管不同, 还可以先闭合 第二开关管和第三开关管, 令第一开关管和第四开关管断开, 以使流经第 一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经第一交流 源或第一交流负载的电流, 以及对第一电容充电的电流, 在持续设定的第 一时间段使流经第一电感和第二电感的电感电流达到峰值后关断所述第二 开关管和所述第三开关管;
第一二极管单元和第四二极管单元处于导通状态, 闭合第一开关管和 第四开关管, 以实现第一开关管和第四开关管的低电压导通。 进一步的, 闭合所述第一开关管和所述第四开关管后, 流经第一电感 和第二电感的电感电流逐渐减小到零, 被充电的第一电容放电, 以及直流 源供电产生流经所述第一电感和第二电感的反向电感电流并逐渐增大, 在 此过程中 , 对流经所述第一电感和第二电感的电感电流和反向电感电流进 行检测 , 待检测到流经所述第一电感和第二电感的电感电流或反向电感电 流的绝对值小于第一门限时断开所述第一开关管和所述第四开关管, 实现 所述第一开关管和所述第四开关管的低电流关断。 另外, 断开所述第一开关管和所述第四开关管后, 所述第二二极管单 元和所述第三二极管单元处于导通状态, 闭合所述第二开关管和所述第三 开关管, 实现所述第二开关管和所述第三开关管的低电压导通。 本发明上述实施例中实现了开关管的低电压导通和低电流关断, 能够 实现软开关的效果, 对于所述逆变器电路中的逆变电桥为两个以上时, 控 制不同逆变电桥上流经第一电感和第二电感的电感电流达到峰值的时间错 开。 该双极性调制输出方式, 其与单极性调制的不同之处在于, 双极性调 制会使 Ql、 Q4同时闭合或断开, Q3和 Q2也会同时断开, 具体的参考图 7 和图 8, 其调制步骤可以参考图 9。 具体的, 如图 9所示, 该双极性调制输出可以包括如下的步骤: 步骤 201、 闭合第一开关管 Q1和第四开关管 Q4, 并令第二开关管 Q2 和第三开关管 Q3断开, 此时在一个回路中, 电流从直流源的正极输出, 会 经过 Ql、 L l、 C、 R、 L2、 Q4回到直流源的负极, 该回路的电流会对第一 电容 C充电, 其设置的电阻 R能够起到抑制谐振的作用; 另外还包括一个 回路, 即电流从直流源的正极输出, 会经过 Ql、 Ll、 L3、 AC、 L4、 L2、 Q4回到直流源的负极, 其中的第三电感 L3和第四电感 L4, 也能够起到抑 制谐振的作用。 本步骤中, 在 Q1和 Q4闭合初期, 流经电感 L1和 L2的电 感电流会逐渐增大, 在持续第一时间段使所述流经 L1和 L2的电流达到峰 值后, 断开 Q1和 Q4, 该第一时间段可以根据输出需要而预先设定, 其电 感电流的峰值与第一时间段的长短相关; 步骤 202、 闭合第二开关管 Q2和第三开关管 Q3 , 此时, 电感 L1 和 L2会起到续流作用,在关断 Q1后 D2和 D3会随之导通,并处于导通状态, 导通后 D2和 Q2两端的电压,以及 D3和 Q3两端的电压会很小,接近为零, 此时闭合 Q2和 Q3 , 能够实现 Q2和 Q3的低电压导通, 在此过程中, 电感 续流,流经 L1和 L2的电感电流逐渐变小到零; 步骤 203、 在上述步骤 201和步骤 202中都会对 C充电, 当流经 L1和 L2的电流逐渐变小到零后, 第一电容会放电, 以及直流源 DC的供电都会 产生流经电感 L1和 L2的反向电感电流, 此过程中可以控制流经 L1和 L2 的电感电流较小 , 因此可以对流经 L1和 L2的电感电流进行测量, 待检测 到流经 L1和 L2的电流小于第一门限时断开 Q2和 Q3 ,实现 Q2和 Q3的低 电流关断;
步骤 204、 闭合 Q1和 Q4, 由于在 Q2和 Q3关断后, 第一电容 C的放 电电流流经 D1和 D4, 此时 D1和 D4导通, 此时 Q1和 Q4两端的接近为 零, 因此可以实现 Q1和 Q4的低电压导通, 回到上述步骤 201中 Q1和 Q4 同时导通的阶段继续执行; 在前半个交流电的输出周期, 重复执行上述的步骤 201〜步骤 204, 可 以通过控制 Q1和 Q4由闭合到断开的时间, 即第一时间段来控制流经 L1 和 L2的电感电流的峰值, 在第一个 1/4周期内, 可以逐渐增大上述的第一 时间段, 即增大 Q1和 Q4由闭合到断开的时间, 并在第二个 1/4周期内, 逐渐减小上述的第一时间段, 即减小 Q1和 Q4由闭合到断开的时间。 在后半个周期, 可以重复执行如下的步骤: 步骤 205、 闭合第二开关管 Q2和第三开关管 Q3 , 并令第一开关管 Q1 和第四开关管 Q4断开, 此时在一个回路中, 电流从直流源的正极输出, 会 经过 Q3、 L2、 R、 C、 Ll、 Q2回到直流源的负极, 该回路的电流会对第一 电容 C充电, 其设置的电阻 R能够起到抑制谐振的作用; 另外还包括一个 回路, 即电流从直流源的正极输出, 会经过 Q3、 L2、 L4、 AC、 L3、 Ll、 Q2回到直流源的负极, 其中的第三电感 L3和第四电感 L4, 也能够起到抑 制谐振的作用。 本步骤中, 流经电感 L1和 L2的电感电流会逐渐增大, 在 持续第一时间段后断开 Q2和 Q3 , 该第一时间段可以根据实际需要而预先 设定, 其电感电流的峰值与第一时间段的长短相关;
步骤 206、 闭合 Q1和 Q4, 此时, 步骤 205中关断 Q2和 Q3后, 由于 电感 L2和 L1会起到续流作用, 并且 D1和 D4会随之导通, 导通后 D1和 D4两端的电压, 以及 Q1和 Q4两端的电压会 4艮小, 接近为零, 此时闭合 Q1和 Q4, 能够实现 Q1和 Q4的低电压导通, 在此过程中, 电感续流, 流 经 L2和 L1的电流逐渐变小到零;
步骤 207、 在上述步骤 205和步骤 206中都会对 C充电, 待流经 L2和 L1的电流逐渐变小到零后, 第一电容会放电, 以及直流源 DC的供电, 使 得产生流经电感 L2和 L1方向的电感电流, 此过程中可以控制流经 L1和 L2的电感电流较小, 因此可以对流经 L 1和 L2的电感电流进行测量 , 待检 测到流经 L1和 L2的电感电流小于第一门限时断开 Q1和 Q4 , 实现 Q1和 Q4的低电流关断;
步骤 208、 闭合 Q2和 Q3 , 在 Q1和 Q4关断后, C的放电电流流经 D2 和 D3 , D2和 D3导通, 此时 Q2和 Q3两端的接近为零, 此时闭合 Q2和 Q3 , 即可实现 Q2和 Q3的低电压导通, 回到上述步骤 201中 Q2和 Q3同 时导通的阶段继续执行;
重复执行上述的步骤 205〜步骤 208, 可以通过控制第一时间段, 即 Q2 和 Q3由闭合到断开的时间来控制流经 L1和 L2的电感电流的峰值, 在第 一个 1/4周期内, 可以逐渐增大 Q2和 Q3由闭合到断开的时间, 并在第二 个 1/4周期内, 逐渐减小 Q2和 Q3由闭合到断开的时间。 由以上分析可知, 在前 1/2个周期和后 1/2个周期, 其中流经 L1和 L2 的电流方向不同, 可以实现双极性输出。 同时在上述的技术方案中, 也实 现了开关管的零电流断开和零电压导通, 达到了软开关的效果。 本发明上述实施例中提供的逆变器电路, 可以由直流源供电实现开关 管作为软开关的效果, 另外, 对于交流电路中包括交流源的情况, 也可以 实现由交流源供电的情况下, 通过对任一逆变电桥的控制, 也实现软开关 的效果。 具体的, 可以参照图 14, 首先闭合第二开关管 Q2和第四开关管 Q4, 令第一开关管 Q1和第三开关管 Q3断开, 第一交流源 AC供电, 以使流经 第一电感 1和第二电感 L2的电感电流逐渐增大,在持续设定的第一时间段 使流经第一电感 L1和第二电感 L2的电感电流达到峰值后, 断开所述第二 开关管 Q2;流经第一电感 L1和第二电感 L2的电感电流达到峰值的大小和 设定的第一时间段的长短有关;
在第二开关管 Q2断开后, 第一二极管单元 D1处于导通状态, 其两端 的电压很低, 此时闭合第一开关管 Q1 , 能够实现第一开关管 Q1的低电压 导通。
进一步, 在闭合第一开关管 Q1后, 在流经第一电感 L1和第二电感 L2 的电感电流逐渐减小到零, 直流源 DC供电产生流经所述第一电感 L1和第 二电感 L2的反向电感电流并逐渐增大的过程中,若检测到流经所述第一电 感 L1和第二电感 L2的电感电流或反向电感电流的绝对值小于第一门限时 , 断开所述第一开关管 Q1 , 实现所述第一开关管 Q1的低电流关断。
另外, 在断开所述第一开关管 Q1后, 第二二极管单元 D2处于导通状 态, 两端的电压 4艮低, 此时闭合第二开关管 Q2, 实现第二开关管 Q2的低 电压导通。
在执行上述步骤的过程中, 可以实现开关管的软开关性能, 并且进一 步的可以在逆变器电路的前半个输出周期内重复执行上述的步骤, 对于一 个 1/4周期, 依次增大第一时间段的长度, 对于第二个 1/4周期, 依次减小 第一时间段的长度, 使得流经 L1和 L2的电流的峰值先增大后减小。 另夕卜, 对于所述逆变器电路中的逆变电桥为两个以上的情况, 可以控制不同逆变 电桥上流经第一电感和第二电感的电感电流达到峰值的时间错开。 另外, 对于逆变器电路的后半个输出周期, 可以按照如下的方法进行 控制, 参考图 16, 首先, 闭合第二开关管 Q2和第四开关管 Q4, 令第一开 关管 Q1和第三开关管 Q3断开, 第一交流源 AC供电, 流经第一电感 L1 和第二电感 L2的电感电流逐渐增大,持续设定的第一时间段使流经第一电 感 L1和第二电感 L2的电感电路达到峰值后断开所述第四开关管 Q4; 断开第四开关管 Q4后, 第三二极管单元 D3处于导通状态, 此时闭合 第三开关管 Q3 , 实现第三开关管 Q3的低电压导通。 进一步的, 上述实施例中还可以闭合在第三开关管 Q3后, 在流经第一 电感 L1和第二电感 L2的电感电流逐渐减小到零, 直流源供电产生流经所 述第一电感 L1和第二电感 L2的反向电感电流并逐渐增大的过程中, 对流 经所述第一电感 L1和第二电感 L2的电感电流和反向电感电流进行检测, 待检测到流经所述第一电感 L1和第二电感 L2的电感电流或反向电感电流 的绝对值小于第一门限时断开所述第三开关管 Q3 , 实现所述第三开关管 Q3的低电流关断。 另外, 在断开第三开关管 Q3后, 第四二极管单元 D4处于导通状态, 两端的电压 4艮低, 此时闭合第四开关管 Q4, 能够实现第四开关管 Q4的低 电压导通。
在逆变器电路的前半个输出周期内重复执行上述的步骤, 其可以实现 开关管的软开关性能, 并且进一步的可以在第一个 1/4周期内,依次增大第 一时间段的长度, 对于第二个 1/4周期, 依次减小第一时间段的长度, 使得 流经 L1和 L2的电流的峰值先增大后减小。 另外, 对于所述逆变器电路中 的逆变电桥为两个以上的情况, 可以控制不同逆变电桥上流经第一电感和 第二电感的电感电流达到峰值的时间错开。
本发明上述实施例中是以一个逆变电桥为例, 对如何实现软开关进行 了说明, 另外在逆变器电路中还可以包括两个以上的逆变电桥, 如图 9所
具体的, 如图 10所示, 该逆变器电路在第一个逆变电桥的基础上, 还 包括另一个逆变电桥, 即第三桥臂和第四桥臂, 该第三桥臂分别连接所述 直流源的正极和负极, 所述第三桥臂和所述第一桥臂并联设置; 第四桥臂 分别连接所述直流源的正极和负极, 所述第四桥臂和所述第二桥臂并联设 置。
上述第三桥臂包括串联的第五开关元件组 15和第六开关元件组 16,所 述第五开关元件组 15包括并联的第五开关管 Q12和第五二极管单元 D12, 所述第五二极管单元 D12的负极与所述直流源的正极连接, 所述第六开关 元件组 16包括并联的第六开关管和 Q22第六极管单元 D22,所述第六极管 单元 D22的负极与所述直流源的正极连接; 所述第四桥臂包括串联的第七开关元件组 21和第八开关元件组 22,所 述第七开关元件组 21包括并联的第七开关管 Q32和第七二极管单元 D32, 所述第七二极管单元 D32的负极与所述直流源的正极连接, 所述第八开关 元件组 22包括并联的第八开关管 Q42和第八二极管单元 D42,所述第八二 极管单元 D42的负极与所述直流源的正极连接; 对于交流电路,其一端通过电感 L5连接在第三桥臂的第五开关元件组 15和第六开关元件组 16之间, 另一端通过电感 L6连接在第四桥臂的第七 开关元件组 21和第八开关元件组 22之间, 上述的电感 L5和 L6的作用和 连接关系与第一个逆变电桥中的第一电感 L1和第二电感 L2相同。 对于图 10所示的逆变器电路的各个逆变电桥分别利用上述实施例提供 的控制方法进行控制, 且控制不同逆变电桥上流经第一电感和第二电感的 电感电流达到峰值的时间错开, 也就是对于图 10中两个逆变电桥的技术方 案中,流经 L1和 L2的电流,与流经 L5和 L6的电流达到峰值的时间错开。 如图 11所示, 其中上半部分可以看作是流经 L1和 L2的电流, 下半部分可 以看作是流经 L5和 L6的电流, 二者的电流叠加, 并且到达峰值的时间错 开。
另外, 在本发明上述各个实施例中, 其中每个开关元件组中的开关管 和二极管单元可以是独立设置的电子元件,即并联的第一开关管 Q1和第一 二极管单元 D1为独立设置的电子元件, 所述第二开关管 Q2和第二二极管 单元 D2 为独立设置的电子元件, 所述第三开关管 Q3 和第三二极管单元 D3为独立设置的电子元件,以及所述第四开关管 Q4和第四二极管单元 D4 为独立设置的电子元件。 并且, 本发明具体实施例中, 其中的第一二极管单元 Dl、 所述第二二 极管单元 D2、 所述第三二极管单元 D3或所述第四二极管单元 D4可以由 如下任一方式构成: 单独设置的一个二极管或串联的两个以上的二极管组 成的二极管组。 而上述的第一开关管 Ql、 第二开关管 Q2、 第三开关管 Q3 和所述第四开关管 Q4可以为如下的任一开关管: MOSFET管、三极管、 IGBT 管或晶闸管。
另外, 由于在 MOSFET管中会寄生一个二极管,该 MOSFET管本身就 包括了开关管和二极管的功能, 因此可以使用该寄生的二极管作为二极管 单元, 此时的第一开关元件组、 所述第二开关元件组、 所述第三开关元件 组和所述第四开关元件组可以直接使用 MOSFET管。 第一开关管 Q1和所 述第一二极管单元 D1 , 所述第二开关管 Q2和所述第二二极管单元 D2, 所 述第三开关管 Q3和所述第三二极管单元 D3 , 以及所述第四开关管 Q4和 所述第四二极管单元 D4均为相应的 MOSFET管中的元件。 本发明上述实施例提供的技术方案也能够实现功补偿功能, 具体的无 功补偿可以如图 12所示, 在 T1和 T3时间段, 其中电压源 AC的输出电压 和电流同相; 其工作同正常纯有功输出一样, 仅电流的设置会稍有不同, 其可以通过控制导通时间达到, 而在 T2和 T3时间段内, 可以控制使电压 源 AC的输出电压和电流反相, 即可实现无功补偿。 具体的, 针对上述提及的单极性调制输出方式, 其电路可以如图 13、 14、 15和 16所示, 分别对应时间段 Tl、 Τ2、 Τ3和 Τ4, 以下对其工作原 理进行说明: a) , 对应图 12的时间段 T1 , 如图 13所示, 可以控制电流方向从 L1 , 经第一电压源 AC至 L2, 并且控制第一电压源 AC的正极在 L1一侧, 第一 电压源的负极在 L2—侧, 该工作周期内 Q4—直保持导通状态, Q3—直断 开, Q1作为主控开关管, 使 Q1和 Q2高频交替工作, T1时间段内第一电 压源的输出电压与电流方向同相, 可以实现正常纯有功输出;
b)、 对应图 11中的时间段 T2, 如图 14所示, 第一电压源 AC的正极 在 L1一侧, 第一电压源的负极在 L2—侧, 第一电压源 AC供电, 电流从 L2, 经电压源 AC至 L1 , 可以先闭合 Q2和 Q4, 其中一个电流回路为 L3、 Ll、 Q2、 Q4 ( D4 ) 、 L2、 L4 和交流源, 另一个回路由 C、 Ll、 Q2、 Q4
( D4 ) 、 R、 C构成, 当关断 Q2时, L1和 L2上的电流流经 Dl ( Q1 ) 、 第一电流源 DC, 再到 Q4 ( D4 )形成回路。 通过调节 Q2的导通时间可以 得到不同的输出电流。 在电压过零点时, 为了维持电流, 可以增加 Q2 和 Q4的导通时间。 该时间段 T2内, 第一电流源 AC的输出电压和电流反向, 可以实现市电向逆变器电路灌入无功, 改时间段 T2 内, Q4—直处于导通 状态, Q1和 Q2高频交替工作, Q2作为主控开关管; c)、 对应图 11的时间段 T3 , 如图 15所示, 可以控制电流方向从 L2, 经电压源 AC至 L1 , 并且控制电压源 AC的正极在 L2—侧, 电压源的负极 在 L1一侧,使得电压源的输出电压与电流同相,可以实现正常纯有功输出, 该时间段 T3内, Q2—直处于导通状态, Q3和 Q4进行高频交替工作, Q3 作为主控开关管使用; d)、对应图 11中的时间段 T4,如图 16所示,电压源的正极在 L2—侧, 电压源的负极在 L1一侧, 电流从 L1 , 经电压源至 L2 , 可以先闭合 Q2和 Q4, 其中一个电流回路为 L4、 L2、 Q4、 Q2 ( D2 ) 、 L2、 L3和交流源, 另 一个回路由 C、 R、 L2、 Q4 ( D4 ) 、 Ll、 C构成, 该时间段 T4内, 电流源的 输出电压和电流反相, 可以实现市电向逆变电路灌入无功, 期间 Q2—直处 于导通状态, Q3和 Q4高频交替工作, Q4作为主控管。 另外, 对于双极性调制方案, 也可以实现无功补偿, 具体的可如图 17、 图 18、 图 19和图 20所示。 如图 17所示,在 T1时间段内,首先闭合 Q1和 Q4,令 Q3和 Q2关断, 电流从 L1 , 经第一电压源 AC至 L2 , 第一电压源的正极位于 L1一侧, 第 一电压源的负极位于 L2—侧, 在之后的控制过程中, 重复同时断开 Q1和 Q4, 以及同时闭合 Q3和 Q3 , 使得在 T1时间段内, 输出电压和电流同相, 可以实现纯有功输出;
如图 18所示,在 T2时间段内,首先闭合 Q2和 Q3 ,令 Q1和 Q4关断, 电流从 L2 , 经第一电压源 AC至 L1 , 第一电压源的正极位于 L1一侧, 第 一电压源的负极位于 L2—侧, 在之后的控制过程中, 重复同时断开 Q2和 Q3 , 以及同时闭合 Q1和 Q4 , 使得在 T2时间段内, 输出电压和电流反相, 可以实现市电向逆变电路灌入无功。
如图 19所示,在 T3时间段内,首先闭合 Q1和 Q4,令 Q3和 Q2关断, 电流从 L1 , 经第一电压源 AC至 L2 , 第一电压源的正极位于 L1一侧, 第 一电压源的负极位于 L2—侧, 在之后的控制过程中, 重复同时断开 Q1和 Q4, 以及同时闭合 Q3和 Q3 , 使得在 T3时间段内, 输出电压和电流同相, 可以实现纯有功输出; 如图 20所示,在 T4时间段内,首先闭合 Q2和 Q3 ,令 Q1和 Q4关断, 电流从 L2 , 经第一电压源 AC至 L1 , 第一电压源的正极位于 L1一侧, 第 一电压源的负极位于 L2—侧, 在之后的控制过程中, 重复同时断开 Q2和 Q3 , 以及同时闭合 Q1和 Q4 , 使得在 T4时间段内, 输出电压和电流反相, 可以实现市电向逆变电路灌入无功。 本领域普通技术人员可以理解: 实现上述方法实施例的全部或部分步 骤可以通过程序指令相关的硬件来完成, 前述的程序可以存储于一计算机 可读取存储介质中, 该程序在执行时, 执行包括上述方法实施例的步骤; 而前述的存储介质包括: ROM、 RAM, 磁碟或者光盘等各种可以存储程序 代码的介质。 最后应说明的是: 以上实施例仅用以说明本发明的技术方案, 而非对 其限制; 尽管参照前述实施例对本发明进行了详细的说明, 本领域的普通 技术人员应当理解: 其依然可以对前述各实施例所记载的技术方案进行修 改, 或者对其中部分技术特征进行等同替换; 而这些修改或者替换, 并不 使相应技术方案的本质脱离本发明各实施例技术方案的范围。

Claims

权利要求
1、 一种逆变器电路, 其特征在于, 包括直流源、 交流电路和一个以上 的逆变电桥, 所述逆变电桥包括第一桥臂和第二桥臂, 所述第一桥臂分别 连接所述直流源的正极和负极, 所述第二桥臂分别连接所述直流源的正极 和负极, 所述第一桥臂和所述第二桥臂并联设置; 所述第一桥臂包括串联的第一开关元件组和第二开关元件组, 所述第 一开关元件组包括并联的第一开关管和第一二极管单元, 所述第一二极管 单元的负极与所述直流源的正极连接, 所述第二开关元件组包括并联的第 二开关管和第二二极管单元, 所述第二二极管单元的负极与所述直流源的 正极连接;
所述第二桥臂包括串联的第三开关元件组和第四开关元件组, 所述第 三开关元件组包括并联的第三开关管和第三二极管单元, 所述第三二极管 单元的负极与所述直流源的正极连接, 所述第四开关元件组包括并联的第 四开关管和第四二极管单元, 所述第四二极管单元的负极与所述直流源的 正极连接; 所述交流电路包括并联的第一交流源和第一电容,或包括并联的第一交流负 载和第一电容, 所述交流电路的一端通过第一电感连接在所述第一桥臂的第一开 关元件组和第二开关元件组之间, 所述交流电路的另一端通过第二电感连接在所 述第二桥臂的第三开关元件组和第四开关元件组之间。
2、 根据权利要求 1所述的逆变器电路, 其特征在于, 所述电路还包括 第三电感和第四电感, 在所述交流电路包括第一交流源时, 所述第一电容 与所述第三电感、 所述第一交流源和所述第四电感组成的电路并联; 在所 述交流电路包括第一交流负载时, 所述第一电容与所述第三电感、 所述第 一交流负载和所述第四电感组成的电路串联。
3、 根据权利要求 1或 2所述的逆变器电路, 其特征在于, 所述电路还 包括一电阻, 所述电阻与所述第一电容串联。
4、 根据权利要求 1至 3任一项所述的逆变器电路, 其特征在于, 所述 第一开关管和所述第一二极管单元为独立设置的电子元件, 所述第二开关 管和所述第二二极管单元为独立设置的电子元件, 所述第三开关管和所述 第三二极管单元为独立设置的电子元件, 以及所述第四开关管和所述第四 二极管单元为独立设置的电子元件。
5、 根据权利要求 4所述的逆变器电路, 其特征在于, 所述第一二极管 单元、 所述第二二极管单元、 所述第三二极管单元或所述第四二极管单元 由如下任一方式构成: 单独设置的一个二极管或串联的两个以上的二极管 组成的二极管组; 所述第一开关管、 所述第二开关管、 所述第三开关管和所述第四开关 管为如下任一开关管: MOSFET管、 三极管、 IGBT管或晶闸管。
6、 根据权利要求 1至 5任一项所述的逆变器电路, 其特征在于, 所述 第一开关元件组、 所述第二开关元件组、 所述第三开关元件组和所述第四 开关元件组为 MOSFET管, 所述第一开关管和所述第一二极管单元, 所述 第二开关管和所述第二二极管单元, 所述第三开关管和所述第三二极管单 元, 以及所述第四开关管和所述第四二极管单元均为相应的 MOSFET管中 的元件。
7、 一种针对权利要求 1-6任一所述的逆变器电路的控制方法, 其特征 在于, 对于所述逆变器电路中的任一逆变电桥, 根据如下的方法进行控制: 闭合第一开关管和第四开关管, 令第二开关管和第三开关管断开, 以 使流经第一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经 第一交流源或第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时间段使流经所述第一电感和所述第二电感的电感 电流达到峰值后, 断开所述第一开关管;
第二二极管单元处于导通状态, 闭合所述第二开关管, 实现所述第二 开关管的低电压导通。
8、 根据权利要求 7所述的逆变器电路的控制方法, 其特征在于, 在所 述闭合第二开关管后, 所述方法还包括: 在流经所述第一电感和所述第二电感的电感电流逐渐减小到零, 被充 电的所述第一电容放电产生流经所述第一电感和所述第二电感的反向电感 电流并逐渐增大的过程中, 若检测到流经所述第一电感和所述第二电感的 电感电流或反向电感电流的绝对值小于第一门限时, 断开所述第二开关管, 实现所述第二开关管的低电流关断。
9、 根据权利要求 8所述的逆变器电路的控制方法, 其特征在于, 在所 述断开第二开关管后, 所述方法还包括: 第一二极管单元处于导通状态, 闭合所述第一开关管, 实现所述第一 开关管的低电压导通。
10、根据权利要求 7-9任一所述的逆变器电路的控制方法,其特征在于, 在所述逆变器电路中的逆变电桥为两个以上时, 控制不同逆变电桥上流经 所述第一电感和所述第二电感的电感电流达到峰值的时间错开。
11、一种针对权利要求 1-6任一所述的逆变器电路的控制方法,其特征 在于, 对于所述逆变器电路中的任一逆变电桥, 根据如下的方法进行控制: 闭合第一开关管和第四开关管, 令第二开关管和第三开关管断开, 以 使流经第一电感和第二电感的电感电流逐渐增大, 所述电感电流分为流经 第一交流源或第一交流负载的电流, 以及对第一电容充电的电流; 在持续设定的第一时间段使流经所述第一电感和所述第二电感的电感 电流达到峰值后, 断开所述第一开关管和所述第四开关管;
第二二极管单元和第三二极管单元处于导通状态, 闭合所述第二开关 管和所述第三开关管, 以实现所述第二开关管和所述第三开关管的低电压 导通。
12、 根据权利要求 11所述的逆变器电路的控制方法, 其特征在于, 在 所述闭合第二开关管和第三开关管后, 所述还包括: 在流经所述第一电感和所述第二电感的电感电流逐渐减小到零, 被充 电的第一电容放电, 以及直流源供电产生流经所述第一电感和所述第二电 感的反向电感电流并逐渐增大的过程中, 若检测到流经所述第一电感和所 述第二电感的电感电流或反向电感电流的绝对值小于第一门限时, 断开所 述第二开关管和所述第三开关管, 实现所述第二开关管和所述第三开关管 的低电流关断。
13、 根据权利要求 12所述的逆变器电路的控制方法, 其特征在于, 在 所述断开第二开关管和所述第三开关管后, 所述方法还包括: 第一二极管单元和第四二极管单元处于导通状态, 闭合所述第一开关 管和所述第四开关管, 实现所述第一开关管和所述第四开关管的低电压导 通。
14、根据权利要求 11-13任一所述的逆变器电路的控制方法,其特征在 于, 在所述逆变器电路中的逆变电桥为两个以上时, 控制不同逆变电桥上 流经所述第一电感和所述第二电感的电感电流达到峰值的时间错开。
15、一种针对权利要求 1-6任一所述的逆变器电路的控制方法,其特征 在于, 在交流电路包括第一交流源时, 对于所述逆变器电路中的任一逆变电 桥, 根据如下的方法进行控制:
闭合第二开关管和第四开关管, 令第一开关管和第三开关管断开, 第 一交流源供电, 以使流经第一电感和第二电感的电感电流逐渐增大; 在持续设定的第一时间段使流经所述第一电感和所述第二电感的电感 电流达到峰值后, 断开所述第二开关管;
第一二极管单元处于导通状态, 闭合所述第一开关管, 实现所述第一 开关管的低电压导通。
16、 根据权利要求 15所述的逆变器电路的控制方法, 其特征在于, 在 所述闭合第一开关管后, 所述方法还包括: 在流经所述第一电感和所述第二电感的电感电流逐渐减小到零, 直流 源供电产生流经所述第一电感和所述第二电感的反向电感电流并逐渐增大 的过程中, 若检测到流经所述第一电感和所述第二电感的电感电流或反向 电感电流的绝对值小于第一门限时, 断开所述第一开关管, 实现所述第一 开关管的低电流关断。
17、 根据权利要求 16所述的逆变器电路的控制方法, 其特征在于, 在 所述断开第一开关管后, 所述方法还包括:
第二二极管单元处于导通状态, 闭合所述第二开关管, 实现所述第二 开关管的低电压导通。
18、根据权利要求 15-17任一所述的逆变器电路的控制方法,其特征在 于, 在所述逆变器电路中的逆变电桥为两个以上时, 控制不同逆变电桥上 流经所述第一电感和所述第二电感的电感电流达到峰值的时间错开。
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CN108321834B (zh) * 2017-01-16 2020-03-06 浙江昱能科技有限公司 一种并网逆变器的控制方法及控制器
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