WO2006002994A1 - Etage d'entree de recepteur radio et procede pour la conversion en frequence d'un signal d'entree - Google Patents

Etage d'entree de recepteur radio et procede pour la conversion en frequence d'un signal d'entree Download PDF

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Publication number
WO2006002994A1
WO2006002994A1 PCT/EP2005/007282 EP2005007282W WO2006002994A1 WO 2006002994 A1 WO2006002994 A1 WO 2006002994A1 EP 2005007282 W EP2005007282 W EP 2005007282W WO 2006002994 A1 WO2006002994 A1 WO 2006002994A1
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WO
WIPO (PCT)
Prior art keywords
radio receiver
receiver front
mixer
signal
frequency
Prior art date
Application number
PCT/EP2005/007282
Other languages
English (en)
Inventor
Henrik Sjöland
Original Assignee
Telefonaktiebolaget L M Ericsson (Publ)
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from EP04015808A external-priority patent/EP1615336B1/fr
Application filed by Telefonaktiebolaget L M Ericsson (Publ) filed Critical Telefonaktiebolaget L M Ericsson (Publ)
Priority to CN2005800296635A priority Critical patent/CN101010872B/zh
Priority to US11/571,614 priority patent/US20070287403A1/en
Priority to JP2007519711A priority patent/JP4634448B2/ja
Publication of WO2006002994A1 publication Critical patent/WO2006002994A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0007—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage
    • H04B1/0014—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at radiofrequency or intermediate frequency stage using DSP [Digital Signal Processor] quadrature modulation and demodulation
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1206—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification
    • H03B5/1209—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier having two current paths operating in a differential manner and a current source or degeneration circuit in common to both paths, e.g. a long-tailed pair.
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1206—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification
    • H03B5/1212—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair
    • H03B5/1215—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device using multiple transistors for amplification the amplifier comprising a pair of transistors, wherein an output terminal of each being connected to an input terminal of the other, e.g. a cross coupled pair the current source or degeneration circuit being in common to both transistors of the pair, e.g. a cross-coupled long-tailed pair
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1231—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device the amplifier comprising one or more bipolar transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B5/00—Generation of oscillations using amplifier with regenerative feedback from output to input
    • H03B5/08—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance
    • H03B5/12—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device
    • H03B5/1237—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator
    • H03B5/124—Generation of oscillations using amplifier with regenerative feedback from output to input with frequency-determining element comprising lumped inductance and capacitance active element in amplifier being semiconductor device comprising means for varying the frequency of the generator the means comprising a voltage dependent capacitance
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1433—Balanced arrangements with transistors using bipolar transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1441—Balanced arrangements with transistors using field-effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/145—Balanced arrangements with transistors using a combination of bipolar transistors and field-effect transistors
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1458—Double balanced arrangements, i.e. where both input signals are differential
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/14—Balanced arrangements
    • H03D7/1425—Balanced arrangements with transistors
    • H03D7/1483—Balanced arrangements with transistors comprising components for selecting a particular frequency component of the output
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
    • H03D7/16—Multiple-frequency-changing
    • H03D7/165—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
    • H04B1/0028—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage
    • H04B1/0032—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage with analogue quadrature frequency conversion to and from the baseband
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/16—Circuits
    • H04B1/26—Circuits for superheterodyne receivers
    • H04B1/28—Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
    • H03B2200/006—Functional aspects of oscillators
    • H03B2200/0078—Functional aspects of oscillators generating or using signals in quadrature
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/0043—Bias and operating point
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/0047—Offset of DC voltage or frequency
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
    • H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
    • H03D2200/0041—Functional aspects of demodulators
    • H03D2200/0088—Reduction of intermodulation, nonlinearities, adjacent channel interference; intercept points of harmonics or intermodulation products

Definitions

  • the present invention relates to an N-phase radio receiver front-end for converting an input signal having a first frequency to an output signal having a second frequency.
  • the invention also relates to a method for converting an input signal having a first frequency to an output signal having a second frequency.
  • the radio receiver front-end incorporates conversion of incoming radio frequency (RF) signals to one or more intermediate frequency (IF) signals, the last of which is then converted to base band.
  • the radio receiver front-end may comprise a low noise amplifier (LNA) with a substantial voltage gain.
  • LNA low noise amplifier
  • one or several mixers is/are provided for converting the input signal to the IF signal (s) , which is/are provided at the output of the mixer(s) .
  • a quadrature radio receiver front-end is designed to mix a differential or single-ended input signal with four local oscillator signals having different phase, and provide two output signals, one for I-channel and one for Q-channel .
  • the input signal may comprise superimposed out-of- band interference.
  • one or several filters for processing the input signal is/are provided.
  • a pre-filter such as a band-select filter, is provided before the LNA to suppress the out-of- band interference. Additional filters may also be provided for processing the input signal.
  • filters are difficult to implement with on-chip design. Thus, the filters must often be implemented off-chip. This is a disadvantage as off-chip components make the radio receiver front-end more expensive, larger and complex. Consequently, in the development towards smaller and less expensive radio receivers most of the off-chip filters have been removed.
  • band-select filter In today's homodyne receivers, one off-chip filter remaining is the band-select filter. If also the band- select filter could be removed, substantial costs and space could be saved. This is especially true for multi-band radio receiver front-ends requiring one band-select filter per band. If also multiple antennas are used, the impact is even higher.
  • a conventional radio receiver front-end does not fulfill this requirement without a pre-filter, such as a band-select filter.
  • the band- select filter may be integrated on-chip.
  • this solution does not fulfill the maximum out-of-band requirements of different mobile communication standards, such as the GSM or the UMTS (Universal Mobile Telecommunication Standard) standards.
  • radio receiver front-end which is less complex than radio receiver front-ends known in the art, and which may be implemented with on-chip technology. It is also an object of the invention to provide a method for converting an input signal having a first frequency to an output signal having a second frequency. According to a first aspect of the invention, these objects are achieved by an N-phase radio receiver front-end according to the invention, which neither has an on-chip nor an off-chip band-select filter.
  • the N-phase radio receiver front-end comprises a low noise amplifier, a mixer arrangement, and a signal generator.
  • An input port of the N-phase radio receiver front-end is directly connected to an input port of the low noise amplifier.
  • the mixer arrangement is a current mode mixer arrangement, as the input signal has not been converted to voltage before mixing.
  • An output port of the low noise amplifier is directly connected to the input port of the mixer arrangement .
  • the signal generator is adapted to generate N phase shifted local oscillator signals. The phase shifted local oscillator signals may be used for selectively activating mixer cores of the mixer arrangement .
  • the mixer arrangement may comprise N/2 mixer cores. Each mixer core may have an input terminal directly connected to the input port of the mixer arrangement .
  • the mixer cores may be single-balanced or double-balanced mixer cores .
  • the low noise amplifier may be a single ended or a differential amplifier.
  • the output port of the mixer arrangement may be connected to an active or passive frequency selective load.
  • the frequency selective load may comprise N/2 current to voltage conversion means, whereby out-of-band interference of a signal input to the radio receiver front-end may be suppressed.
  • Each current to voltage conversion means may comprise a mixer load connected to a respective output terminal of the mixer cores and signal grounding means, respectively.
  • Each mixer load may be a resistor connected in parallel with a capacitor.
  • the capacitor of each mixer load has a value, which is effective for suppressing out-of-band interference of a signal input to the radio receiver front- end when said signal has been mixed.
  • the capacitance of the capacitor of each mixer load may be variable for suppressing out-of-band interference of input signals having different bandwidths.
  • the signal generator may be an oscillator for providing signals for driving the mixer cores.
  • the oscillator may be a voltage controlled oscillator
  • the mixer arrangement may be connected to the voltage controlled oscillator by means of transformers for providing local oscillator signals, such as quadrature local oscillator signals. Supplying the local oscillator signals by means of transformers is an advantage as no low- frequency noise will be introduced to the local oscillator terminals of the mixer arrangement.
  • the local oscillator may comprise quadrature oscillators with LC-tanks.
  • Inductors of the LC-tanks may provide primary windings of the transformers, and inductors connected to local oscillator input terminals of the mixer may provide secondary windings of said transformers. Thus, no additional components are needed for providing the transformers except the inductors for providing the secondary windings.
  • the capacitor of each LC-tank may be a variable capacitor for adjusting the frequency of the local oscillator signals.
  • the signal generator may be provided by a high frequency oscillator and a frequency divider arranged to provide N non-overlapping local oscillator signals having a duty cycle of substantially 1/N. With quadrature oscillator signals, the duty cycle should be substantially 25% for each signal.
  • the objects are achieved by the use of the N-phase radio receiver front-end according to the invention in a wireless electronic communication apparatus for converting an input signal having a first frequency to a signal having a second frequency.
  • the objects are achieved by a wireless electronic communication apparatus comprising a N-phase radio receiver front-end according to the invention.
  • the objects are achieved by a method for converting an input signal having a first frequency to an output signal having a second frequency in an N-phase radio receiver front-end.
  • the method comprises the steps of receiving the input signal at an input port of the radio receiver front-end; amplifying the input signal comprising out-of-band interference in a low noise amplifier; mixing the input signal and the out-of-band interference with a plurality of phase shifted local oscillator signals having a second frequency in a current mode mixer arrangement to generate a mixed signal having the second frequency.
  • the mixed signal may comprise out-of-band interference.
  • the method may further comprise the step of suppressing the out-of-band interference of the mixed signal .
  • the step of suppressing may comprise supplying the mixed input signal comprising the out-of-band interference to a passive or active frequency selective load.
  • the frequency selective load may be a mixer load, which is connected to a respective output terminal of the mixer arrangement and signal grounding means, respectively.
  • the mixed signal may be an IF signal .
  • the step of suppressing may comprise suppressing by means of a capacitor of the mixer load, which has a value that is effective for suppressing out-of-band interference of the mixed signal .
  • the method may comprise adjusting the capacitance of the capacitor, which may be a variable capacitor, of the frequency selective load for suppressing out-of-band interference of mixed input signals having different bandwidths.
  • the method may also comprise the steps of generating the local oscillator signals, and supplying said generated local oscillator signals to N/2 mixer cores of the mixer arrangement.
  • the method may comprise adjusting the capacitance of a capacitor of an oscillator connected to the mixer arrangement for adjusting the frequency of said local oscillator signals.
  • Fig. 1 is a front view of a mobile communication apparatus comprising a N-phase radio receiver front-end according to the invention
  • Fig. 2 is a block-diagram of the N-phase radio receiver front-end according to the invention.
  • Fig. 3 is a circuit-diagram of an embodiment of the N-phase radio receiver front-end according to the invention.
  • Fig. 4a is a circuit-diagram of a first embodiment of a voltage controlled oscillator for generating low noise local oscillator signals
  • Fig. 4b is a signal diagram of local oscillator signals
  • Fig. 5a is a block-diagram of a high frequency oscillator connected to frequency dividers for generating low noise local oscillator signals;
  • Fig. 5b is a signal diagram of local oscillator signals
  • Fig. 6 is a circuit-diagram of another embodiment of the N-phase radio receiver front-end according to the invention.
  • Fig. 7 is a flow-chart of one embodiment of the method according to the invention.
  • Fig. 1 illustrates a mobile telephone 1 as one exemplifying wireless electronic communication apparatus, in which an N-phase radio receiver front-end according to the present invention may be utilized.
  • the invention is not limited to implementation in a mobile telephone 1.
  • the invention may be implemented in a wide variety of electronic equipment wherein a radio receiver front-end is required for receiving and processing radio frequency (RF) input signals, such as a mobile radio terminal, a pager, a communicator, an electronic organizer or a smartphone.
  • the mobile telephone 1 may comprise a first antenna 10 and a second auxiliary antenna 11 for receiving input signals.
  • a microphone 12, a loudspeaker 13, a keypad 14, and a display 15 provide a man-machine interface for operating the mobile telephone 1.
  • the mobile telephone may in operation be connected to a radio station 20 (base station) of a mobile communication network 21, such as a GSM, UMTS, PCS (Personal
  • the mobile telephone 1 may in operation establish a second wireless link to a peripheral device 30 via a second radio link 31 by means of the auxiliary antenna 11.
  • the second radiolink 31 is e.g. a Bluetooth ® link, which is established in the 2.4 (2.400- 2.480) GHz frequency range.
  • the mobile telephone 1 comprises radio resources, which are adapted according to the relevant technologies that are used.
  • the mobile telephone 1 comprises a first radio access means, such as a transceiver, for communicating radio signals with the base station 20, and a second radio access means for communicating radio signals with the peripheral device 30.
  • one radio access means may be switchable to communicate radio signals with either the base station 20 or the peripheral device 30.
  • the peripheral device 30 may be any device having wireless communicating capabilities, such as according to Bluetooth ® technology or any other wireless local area network (WLAN) technology. It comprises an antenna 32 for exchanging signals over the second link 31, and a transceiver (not shown) adapted according to the communication technology that the peripheral device 30 uses.
  • the device may be a wireless headset, a remote server, a fax machine, a vending machine, a printer, a computer etc.
  • a wide variety of electronic equipment may have such communication capabilities and have a need for wirelessly transferring data.
  • Received input signals having radio frequencies (RF) may be processed by the radio receiver front-end according to the invention.
  • the input signals may be single-ended or differential.
  • the input signals are converted to intermediate frequency (IF) signals before further signal processing is applied.
  • the radio receiver front-end of the mobile telephone 1 may comprise a mixer arrangement comprising one or several mixer cores for converting a signal having a first frequency to signals having a second frequency as will be disclosed in the following.
  • Fig. 2 illustrates the radio receiver front-end according to the invention.
  • the antenna 10 may be directly connected to an input port of a low noise amplifier (LNA) 50.
  • LNA 50 is inherently linear or linearized, such that it can handle out-of band interference, e.g. according to the GSM standard, wherein out-of-band interference up to at least 0 dBm should be handled.
  • the RF signal input to the LNA 50 comprises both the desired signal and superimposed out-of-band interference, which are amplified by the gain of the LNA 50.
  • a current input port of an N-phase mixer arrangement 50a is connected to an output port of the LNA 50.
  • the mixer arrangement 50a may comprise N/2 mixer cores 51, 52. In the embodiments of Figs. 2, 3 and 6, quadrature radio receiver front-ends are described. These mixer arrangements 50a comprises a first and a second mixer core 51, 52 having input terminals. Each input port and output port of the mixer arrangement 50a and the LNA 50 may comprise one or several terminals.
  • the first mixer core 51 may be used for the I-channel of the input signal and the second mixer core 52 may be used for the Q-channel of the input signal.
  • the output port of the LNA 50 is directly connected to the input port of the mixer arrangement 50a, i.e.
  • the signal current from the LNA 50 is not converted to a voltage by a load impedance.
  • a signal converted to a voltage would be too large to handle.
  • the mixing is therefore performed in the current domain according to the invention by controlling the mixer by means of the phase shifted LO signals for selectively activating the mixer arrangement 50a, e.g. by selectively activating the mixer cores 51, 52.
  • the interferer may be handled.
  • Each mixer core 51, 52, and thus the mixer arrangement 50a also comprises local oscillator (LO) input terminals for receiving LO signals, which are generated by an LO signal generating means or LO signal generator, to be mixed with the amplified input signal .
  • LO local oscillator
  • the second mixer core 52 is adapted to receive and be responsive to a second LO signal LOQ having a second phase, which is different from the first phase.
  • Output ports of the mixer arrangement 50a e.g. output terminals of the first and the second mixer core 51,
  • the frequency selective load may comprise a first and a second current to voltage conversion means 53, 54.
  • the input signal which now is amplified and mixed to lower frequency signals, may be converted to voltage by the current to voltage conversion means.
  • I- and Q-channel output signals IFi, IF Q may be provided at output ports of the frequency selective load.
  • Each output port of the frequency selective load may comprise first and second terminals.
  • the frequency selective load will also function as a suppression means for suppressing the out-of-band interference.
  • the radio receiver front-end according to Fig. 3 is a quadrature radio receiver front- end.
  • the linearity of the LNA is sufficiently high to handle out-of-band interference of e.g. up to 0 dBm, as described above.
  • the LNA is a common gate or common base LNA provided by an amplifier transistor 60, which may be an input transistor.
  • the transistor 60 may be an FET (Field Effect Transistor) , such as an MOS (Metal Oxide Semiconductor) transistor or a BJT (Bipolar Junction Transistor) transistor.
  • FET Field Effect Transistor
  • MOS Metal Oxide Semiconductor
  • BJT Bipolar Junction Transistor
  • the LNA 50 is provided by an FET transistor.
  • the input port of the quadrature radio receiver front-end is connected to the source terminal of transistor 60.
  • the gate of amplifier transistor 60 is connected to a bias voltage V b i as i- Alternatively, a bias input (gate) of the amplifier transistor 60 is connected to a common mode feedback circuit for controlling the bias of said amplifier transistor 60.
  • the first mixer core 51 may comprise first and second mixer transistors 61a, 62a connected to the input terminal of the first mixer core 51.
  • the mixer transistors 61a, 62a may be FET transistors or BJT transistors. It is an advantage of the BJT transistor that it is quicker than the FET transistor, which provides a higher linearity.
  • the mixer transistors 61a, 62a are provided by BJT transistors. The emitter of each mixer transistor 61a, 62a is connected to the input terminal of the first mixer core 51.
  • each mixer transistor is connected to an LO (Local Oscillator) input terminal of the first mixer core 51.
  • Each mixer transistor 61a, 62a is responsive to a respective quadrature LO signal.
  • the first mixer transistor 61a is responsive to a first quadrature LO signal LOi + having a first phase.
  • the second mixer transistor 62a is responsive to a second LO signal LOi- having a second phase, which is phase shifted 180° relative the first phase.
  • the collectors of the mixer transistors 61a, 62a are connected to first and second output terminals, respectively, of the first mixer core 51.
  • the frequency selective load e.g. the current to voltage conversion means, may comprise a capacitor 67a provided between the input terminals of the frequency selective load.
  • the frequency selective load will be operative for filtering of out-of-band interference and to provide some channel filtering.
  • the mixer arrangement 50a and the mixer cores 51, 52 are current mode mixers operating in the current domain.
  • the output signals from the first mixer core 51 are supplied to the frequency selective load.
  • the frequency selective load may comprise the first current to voltage conversion means 53, which may convert the output signals from the first mixer core 51 to voltage.
  • the first current to voltage conversion means 53 may comprise separate conversion means for each output signal.
  • Each conversion means may comprise passive components, such as resistors 63a, 65a and capacitors 64a, 66a connected in parallel to the output terminals of the first mixer core 51 and signal grounding means, such as supply voltage.
  • the first mixer transistor 61a is connected to resistor 63a and capacitor 64a
  • the second mixer transistor 62a is connected to resistor 65a and capacitor 66a.
  • the first and second current to voltage conversion means may also comprise active components.
  • a transistor connected as a resistor may replace resistor 63a and/or resistor 65a.
  • the first and second current to voltage conversion means 53, 54 may comprise transimpedance amplifiers to convert the current signal output from the mixer arrangement 50a.
  • the transfer function of such a transimpedance amplifier can be made frequency selective.
  • a first IF (Intermediate Frequency) output signal IFi for the I-channel may be generated between the output terminals of the first mixer core 51.
  • the desired signal which may be centered at low frequencies, is not significantly attenuated by the capacitors 64a, 66a and 67a.
  • the out-of-band interference may be heavily attenuated by choosing suitable values of capacitors 64a, 66a and 67a.
  • LO to IF leakage is suppressed by capacitors 64a, 66a and 67a, enabling the use of single balanced mixer cores and a single-ended LNA.
  • a single-ended LNA removes the need of an external balun. An external filter may perform the balun function. Thus, if a differential LNA is utilized, a stand alone external balun may need to be provided.
  • the signals after the radio receiver front-end are differential, which is suitable for further processing on-chip.
  • the LNA 50 may alternatively be provided by a feedback LNA, which is sufficiently linear for handling out-of-band interference of up to 0 dBm for satisfying the GSM standard.
  • a feedback LNA which is sufficiently linear for handling out-of-band interference of up to 0 dBm for satisfying the GSM standard.
  • the linearity requirement has to be considered in each specific case.
  • the second mixer core 52 may comprise first and second mixer transistors 61b, 62b, and is configured as the first mixer core 51.
  • the second current to voltage conversion means 54 comprises a mixer load provided by resistors 63b, 65b and capacitors 64a, 66b, and a capacitor 67b arranged between the input terminals of the second current to voltage conversion means 54.
  • the first mixer transistor 61b of the second mixer core 52 is responsive to a third quadrature LO signal L0 Q+ having a third phase, which is phase shifted 90° relative the first phase.
  • the second mixer transistor 62b of the second mixer core 52 is responsive to a fourth LO signal L0 Q - having a fourth phase, which is phase shifted 270° relative the first phase.
  • the collectors of the mixer transistors 61b, 62b of the second mixer core 52 are connected to first and second output terminals of the second mixer core 52.
  • a second IF (Intermediate Frequency) output signal IF Q for the Q-channel may be outputted between the output terminals of the second mixer core 52.
  • a current device 68 is connected to the input port of the radio receiver front-end and the input port of the LNA 50.
  • the current device 68 may e.g. be provided by a resistor, an inductor, or a transistor connected as a current source.
  • An inductor has the advantage that it causes a lower voltage drop than a resistor or transistor connected as a current source. Also, if current device 68 is provided by an inductor it can tune out parasitic capacitance appearing at the source of transistor 60.
  • the LO input terminals of the first and second mixer cores 51, 52 are connected to an LO signal generator.
  • the LO signal generator is a quadrature LO signal generating means. Since the signal to out-of band interference ratio is not improved by filtering before the input signal is supplied to the mixer arrangement 50a, the phase-noise of the LO signals must be very low at large offset frequencies, e.g. above 20 MHz in a GSM implementation. If the phase-noise is too high, reciprocal mixing of strong out-of-band interference can block the reception of weak signals. In the GSM case, the requirement of the phase-noise will be similar to what is needed in a transmitter.
  • the same or a similar oscillator may be used for generating the LO signals LOi + , LOi., LO 0+ , L0 Q - for the transmitter and the radio receiver front-end.
  • the low- frequency local oscillator noise must also be low, since it is directly transferred to the IF outputs.
  • the signal generator may comprise an oscillator, such as a VCO (Voltage Controlled Oscillator) .
  • VCO Voltage Controlled Oscillator
  • Fig. 4a illustrates one embodiment of a VCO (Voltage Controlled Oscillator) , which may be used for generating quadrature LO signals.
  • VCO Voltage Controlled Oscillator
  • a possibility to generate low phase- noise local oscillator signals substantially free from low- frequency noise is to use oscillators with LC-tanks.
  • the LC-tanks may be part of transformers having secondary windings connected to the mixer cores 51, 52. No local oscillator buffers are needed in this case, and the DC- level of the local oscillator signal fed to the mixer cores may easily be set.
  • the VCO comprises four pairs of transistors 71a, 71b, 72a, 72b, 73a, 73b, 74a, 74b.
  • Said transistors may be provided by FET or BJT transistors.
  • the source of transistor 71a is connected to the drain of transistor 71b.
  • the gate of transistor 71a is connected to the drain of transistor 73a, and the drain of transistor 71a is connected to a first LC-tank comprising an inductor 75 connected in parallel with a capacitor 76.
  • the center tap of inductor 75 is connected to the supply voltage.
  • the value of capacitor 76 will set the frequency of the VCO.
  • the gate of transistor 71b is connected to the drain of transistor 72a and to the gate of transistor 73a.
  • the source of transistor 71b is connected to the drain of a bias transistor 79.
  • the gate of bias transistor 79 will in operation receive a bias voltage Vbias3•
  • the source of bias transistor 79 is connected to grounding means.
  • the drain of transistor 72a is connected to second terminals of inductor 75 and capacitor 76, and to the gate of transistor 73a and 71b.
  • the gate of transistor 72a is connected to the drain of transistor 74a.
  • the source of transistor 72a is connected to the drain of transistor 72b.
  • the gate of transistor 72b is connected to the drain of transistor 71a and to the gate of transistor 74a.
  • the source of transistor 72b is connected to the drain of bias transistor 79.
  • the source of transistor 73a is connected to the drain of transistor 73b.
  • the gate of transistor 73a is connected to the drain of transistor 72a, and the drain of transistor 73a is connected to a second LC-tank comprising an inductor 77 connected in parallel with a capacitor 78, and to the gate of transistor 71a.
  • the center tap of inductor 77 is connected to the supply voltage.
  • the value of capacitor 78 should track that of capacitor 76 and it will set the frequency of the VCO.
  • the gate of transistor 73b is connected to the drain of transistor 74a and to the gate of transistor 72a.
  • the source of transistor 73b is connected to the drain of a bias transistor 80.
  • the gate of bias transistor 79 will in operation receive the bias voltage V b i aS3 •
  • the source of bias transistor 80 is connected to grounding means.
  • the drain of transistor 74a is connected to second terminals of inductor 77 and capacitor 78, and to the gate of transistor 72a and 73b.
  • the gate of transistor 74a is connected to the drain of transistor 71a.
  • the source of transistor 74a is connected to the drain of transistor 74b.
  • the gate of transistor 74b is connected to the gate of transistor 71a and to the drain of transistor 73a.
  • the source of transistor 74b is connected to the drain of bias transistor 80.
  • the VCO is magnetically coupled to the LO input terminals of the mixer cores 51, 52 by means of first and second transformers.
  • the first transformer comprises inductor 75 and an inductor 81 connected to the gate of transistor 61a and the gate of transistor 62a.
  • the primary winding of the first transformer is provided by inductor 75, and the secondary winding thereof is provided by inductor 81.
  • the second transformer comprises inductor 77 and an inductor 82 connected to the gate of transistor 61b and to the gate of transistor 62b.
  • Fig. 4b illustrates the LO signals, which may be generated by the VCO according to the embodiment of Fig. 4a.
  • the LO signal having the highest voltage level will dominate the other LO signals, except at the crossings of the LO signals, due to the phase shifting of the signals. This means that it is possible to interconnect the input terminals of the mixer cores 51, 52.
  • the transistor receiving the LO signal having the highest voltage level will be conducting and thus operative.
  • the transistor receiving the LO signal having the highest voltage level will dominate over the other transistors of the mixer arrangement 50a even if any of the other transistors are conducting to a certain extent.
  • FIG. 5a illustrates an alternative solution for generating the LO signals LOi + , LOi-, LO Q+ , L0 Q _ with sufficiently low phase-noise and low-frequency noise, which are phase shifted relative each other.
  • the LO signals are phase shifted in this embodiment such that substantially only one signal will be in the high state at the time.
  • a high frequency oscillator 90 is connected to a digital frequency divider 91.
  • the frequency divider is arranged to generate quadrature LO signals, i.e. the four LO signals LOi + , LO 1 -, L0 Q+ , LO Q -, of which only one is active at a time.
  • the frequency of the high frequency oscillator should at least be twice the frequency of the output signals from the digital frequency divider 91. It is important to avoid time overlaps when more than one of the four local oscillator signals are simultaneously high. The overlaps can be avoided by arranging the frequency divider 91 to provide an approximate duty cycle of 1/N, i.e. 25% for quadrature signals, for each of the output signals. If overlaps exist, additional noise is generated and the sensitivity to matching inaccuracy of the mixer transistors is increased. However, if the noise requirement is less strict some overlap may be allowed. It is an advantage of the high frequency oscillator 90 and the digital frequency divider 91 that they provide a more compact design, albeit with increased current consumption compared to the VCO implementation of Fig. 4a.
  • the frequency divider 91 may be provided by a Johnson counter with N flip-flops in series, where the output signal of the last flip-flop is fed back to the input terminal of the first one. All flip-flops should be clocked by the same clock signal of N times the frequency of the output signal. The flip-flops have to be forced to a state where only one output is high at the time to avoid loops of false states. The N LO signals can then be extracted at the outputs of the N flip-flops.
  • the LO signals may be square waves, which are substantially non- overlapping.
  • the input signal RFi n is single ended. However, the input signal may equally well be differential, wherein the LNA 50 will be arranged to amplify the differential signal, which is then supplied to double balanced mixer cores instead of single balanced mixer cores as described above.
  • the radio receiver front-end as described above may be adapted to dual mode mobile communication, wherein it may handle incoming signals from at least two mobile communication networks applying different communication standards, such as GSM and UMTS.
  • a dual mode radio receiver front-end may be provided by arranging two radio receiver front-end circuits as disclosed above in parallel, wherein each front-end is adapted according to a specific standard.
  • the parallel-connected circuits may be selectively activated by biasing the LNA of each radio receiver front- end circuit selectively.
  • a controller may be arranged to control the biasing of the LNA of each circuit .
  • a dual mode radio receiver front-end may be provided by changing the bandwidth of the frequency selective load, e.g. of the current to voltage conversion means 53, 54.
  • the capacitors 64a, 64b, and 67a are variable capacitors having selectively variable capacitance values
  • a controller may be arranged to set specific values of the capacitors 64a, 64b, and 67a. The value set will be chosen such that out-of-band interference of input signals of different received signal bandwidths will be suppressed and the signal to be received is essentially unaffected.
  • a topology for making the LNA 50 and the mixer arrangement 50a sufficiently linear for handling out-of-band interference is chosen. If the LNA and the mixer arrangement were not sufficiently linear, the out-of-band interference would cause intermodulation distortion and compression of the input signal as the band- select -filter is removed according to the invention.
  • the radio receiver front-end illustrated in Fig. 6 comprises a double balanced mixer arrangement with a differential LNA.
  • the differential amplifier comprises a first and a second amplifier means, e.g. provided by a first and a second amplifier transistor 160a, 160b, such as a MOS or BJT transistor.
  • the input port of the quadrature radio receiver front-end is connected to the input port of the LNA 50, which is directly connected to source terminals of transistors 160a, 160b, to which an input signal RFi n may be applied.
  • the gates of transistors 160a and 160b are connected to a bias voltage V b i as - Alternatively, the bias input (gate) of transistors 160a and 160b may be connected to a common mode feedback circuit for controlling the bias of said transistors 160a, 160b (not shown) .
  • the first and the second mixer cores 51, 52 each comprise four mixer transistors 161a, 161b, 161c, 161d, 162a, 162b, 162c, 162d.
  • the gates of transistors 161a and 162c are connected to receive the local oscillator signal LOi + .
  • the gates of transistors 161b and 162d are connected to receive the local oscillator signal L0 Q+ .
  • the gates of transistors 161c and 162a are connected to receive the local oscillator signal L0 Q ..
  • the gates of transistors 161d and 162b are connected to receive the local oscillator signal LOi-.
  • the drains of transistors 161a and 162b are connected to a first terminal of capacitor 64a, resistor 63a and capacitor 67a.
  • the drains of transistors 161b and 162a are connected to a first terminal of capacitor 64b, resistor 63b and capacitor 67b.
  • the drains of transistors 161c and 162d are connected to a first terminal of capacitor 66b and resistor 65b and to a second terminal of capacitor 67b.
  • the drains of transistors 161d and 162c are connected to a first terminal of capacitor 66a and resistor 65a, and to a second terminal of capacitor 67a.
  • the first output signal IFi will during operation be generated between output terminals connected to the terminals of capacitor 67a, and the second output signal IF Q will be generated between output terminals connected to the terminals of capacitor 67b.
  • the local oscillator signals LOi + , LOi-, L0 Q+ , L0 Q . may be provided according to the principles as described in relation to Figs. 4a-4b or 5a-5b.
  • Fig. 7 illustrates the method according to the invention.
  • the input signal which comprises out-of-band interference
  • the input signal is received at the input port of the N-phase radio receiver front-end.
  • the input signal comprising the out-of-band interference is amplified in the LNA 50.
  • the amplified input signal and the out-of-band interference is mixed with the phase shifted LO signals, as discussed above, to generate a mixed signal comprising out-of-band interference.
  • step 103 the out-of-band interference of the mixed signal is suppressed, e.g. as discussed above by supplying the mixed signal to the frequency selective load, e.g. to the mixer loads comprising resistors and capacitors.
  • the capacitors 64a, 66a, 64b, 66b, 67a, 67b of the mixer loads may have values that are effective for suppressing the out-of-band interference.
  • the method may comprise the step of setting the value of said capacitors.
  • the method may also comprise the step of supplying the LO signals to the mixer cores, 51, 52.
  • the capacitors 76, 78 of the LC-tanks of the VCO are variable, the method may comprise the step of setting the value of said capacitors .
  • the N-phase radio receiver may be a quadrature radio receiver front end.
  • the N-phase radio receiver may be a quadrature radio receiver front end.
  • virtually any number of phases may be processed by suitably arranging the front- end.
  • six phases may be processed by adding an additional mixer core to the mixer arrangement 50a according to the embodiment of Fig. 4a.
  • the number of different LO signals to be generated will thus be 6.
  • the appropriate number of LO signals may be generated by the frequency divider or designing a VCO according to the principles of the embodiment of Fig. 4a.
  • the number of phases to process may be denoted N.
  • the number of different LO signals to generate will be N.
  • the LO signals will thus be phase shifted by 360°/N relative each other.
  • the present invention may be e.g. be used for down converting RF signals to zero IF or low IF signals without using any band-select filter.
  • the front-end according to the invention will have a compact design, and be cheap to manufacture.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Superheterodyne Receivers (AREA)
  • Amplifiers (AREA)
  • Noise Elimination (AREA)

Abstract

Cette invention concerne un étage d'entrée de récepteur radio à N phases qui sert à convertir un signal d'entrée ayant une première fréquence en signaux de sortie ayant une seconde fréquence, ainsi qu'un procédé qui sert à convertir un signal d'entrée dans un étage d'entrée de récepteur radio à N phases. A cet effet, un port d'entrée de l'étage d'entrée de récepteur radio à N phases est connecté directement à un port d'entrée d'un amplificateur à faible bruit (LNA) (50). Un système mélangeur (50a) est constitué par un système mélangeur de mode courant. Un port de sortie de l'amplificateur à faible bruit est directement connecté à un port d'entrée du système mélangeur. Un générateur de signaux, connecté en mode opérationnel au système mélangeur, est conçu pour générer N signaux d'oscillateur local déphasés.
PCT/EP2005/007282 2004-07-06 2005-07-06 Etage d'entree de recepteur radio et procede pour la conversion en frequence d'un signal d'entree WO2006002994A1 (fr)

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CN2005800296635A CN101010872B (zh) 2004-07-06 2005-07-06 用于对输入信号进行频率变换的无线电接收机前端及方法
US11/571,614 US20070287403A1 (en) 2004-07-06 2005-07-06 Radio-Receiver Front-End and A Method For Frequency Converting An Input Signal
JP2007519711A JP4634448B2 (ja) 2004-07-06 2005-07-06 無線受信機フロントエンド、及び入力信号を周波数変換する方法

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EP04015808.1 2004-07-06
EP04015808A EP1615336B1 (fr) 2004-07-06 2004-07-06 Circuits d'entrée de récepteur radio et procédé pour supprimer l'interférence hors-bande
US58924704P 2004-07-20 2004-07-20
US60/589,247 2004-07-20

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8145155B2 (en) 2005-09-06 2012-03-27 Mediatek, Inc. Passive mixer and high Q RF filter using a passive mixer

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR101325196B1 (ko) * 2012-10-22 2013-11-04 지씨티 세미컨덕터 인코포레이티드 임피던스 쉐이핑을 이용한 수신기

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5507036A (en) * 1994-09-30 1996-04-09 Rockwell International Apparatus with distortion cancelling feed forward signal
US20030062960A1 (en) * 2001-10-01 2003-04-03 Bushman Michael L. Multiphase voltage controlled oscillator
US20030137358A1 (en) * 2002-01-21 2003-07-24 Zheng-Dao Lee Multi-phase oscillator and multi-phase oscillation signal generation method
US20030162521A1 (en) * 1998-11-12 2003-08-28 Broadcom Corporation System and method for on-chip filter tuning

Family Cites Families (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5789799A (en) * 1996-09-27 1998-08-04 Northern Telecom Limited High frequency noise and impedance matched integrated circuits
US6147559A (en) * 1998-07-30 2000-11-14 Philips Electronics North America Corporation Noise figure and linearity improvement technique using shunt feedback
JP2001053553A (ja) * 1999-08-17 2001-02-23 Sony Corp 位相反転回路、ドライブ回路および受信機
FR2811827B1 (fr) * 2000-07-12 2002-10-11 St Microelectronics Sa Dispositif d'amplification a faible bruit, en particulier pour un telephone mobile cellulaire
JP2002353741A (ja) * 2001-03-23 2002-12-06 Rf Chips Technology Inc ミキサ回路

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5507036A (en) * 1994-09-30 1996-04-09 Rockwell International Apparatus with distortion cancelling feed forward signal
US20030162521A1 (en) * 1998-11-12 2003-08-28 Broadcom Corporation System and method for on-chip filter tuning
US20030062960A1 (en) * 2001-10-01 2003-04-03 Bushman Michael L. Multiphase voltage controlled oscillator
US20030137358A1 (en) * 2002-01-21 2003-07-24 Zheng-Dao Lee Multi-phase oscillator and multi-phase oscillation signal generation method

Non-Patent Citations (7)

* Cited by examiner, † Cited by third party
Title
"LOW COST HBT UPCONVERTERS FOR CDMA TRANSMITTER ICS", November 1996, MICROWAVE JOURNAL, HORIZON HOUSE. DEDHAM, US, PAGE(S) 154,156, ISSN: 0192-6225, XP000681574 *
BEFFA F ET AL INSTITUTE OF ELECTRICAL AND ELECTRONICS ENGINEERS: "A 6.5-MW RECEIVER FRONT-END FOR BLUETOOTH IN 0.18-MUM CMOS", 2002 IEEE MTT-S INTERNATIONAL MICROWAVE SYMPOSIUM DIGEST.(IMS 2002). SEATTLE, WA, JUNE 2 - 7, 2002, IEEE MTT-S INTERNATIONAL MICROWAVE SYMPOSIUM, NEW YORK, NY : IEEE, US, vol. VOL. 1 OF 3, 2 June 2002 (2002-06-02), pages 501 - 504, XP001099538, ISBN: 0-7803-7239-5 *
CHEUNG V S L ET AL: "A 1-V 10-mw monolithic bluetooth receiver in a 0.35-/spl mu/m cmos process", ESSCIRC '03 - 2003 CONFERENCE ON EUROPEAN SOLID-STATE CIRCUITS, 16 September 2003 (2003-09-16), pages 687 - 690, XP010677180 *
DARABI H ET AL: "A 4.5-MW 900-MHZ CMOS RECEIVER FOR WIRELESS PAGING", August 2000, IEEE JOURNAL OF SOLID-STATE CIRCUITS, IEEE INC. NEW YORK, US, PAGE(S) 1085-1096, ISSN: 0018-9200, XP001075109 *
KWAN K W ET AL: "A bipolar IF device for GSM mobile telephone", BIPOLAR/BICMOS CIRCUITS AND TECHNOLOGY MEETING, 1992., PROCEEDINGS OF THE 1992 MINNEAPOLIS, MN, USA 7-8 OCT. 1992, NEW YORK, NY, USA,IEEE, US, 7 October 1992 (1992-10-07), pages 55 - 58, XP010104589, ISBN: 0-7803-0727-5 *
PIAZZA F ET AL: "FP 20.1: A 170MHZ RF FRONT-END FOR ERMES PAGER APPLICATIONS", 1 February 1995, IEEE INTERNATIONAL SOLID STATE CIRCUITS CONFERENCE, IEEE INC. NEW YORK, US, PAGE(S) 324-325,389, ISSN: 0193-6530, XP000557658 *
WASSENER H-J: "A Small Package Integrated Two Band Mixer-oscillator SI-IC", CONSUMER ELECTRONICS, 1993. DIGEST OF TECHNICAL PAPERS. ICCE., IEEE 1993 INTERNATIONAL CONFERENCE ON ROSEMONT, IL, USA 8-10 JUNE 1993, NEW YORK, NY, USA,IEEE, US, 8 June 1993 (1993-06-08), pages 106 - 107, XP010107192, ISBN: 0-7803-0843-3 *

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US8145155B2 (en) 2005-09-06 2012-03-27 Mediatek, Inc. Passive mixer and high Q RF filter using a passive mixer

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