WO2002080375A2 - Code spatio-temporel de ponderation complexe non nulle destine a une transmission par antenne a usage multiple - Google Patents

Code spatio-temporel de ponderation complexe non nulle destine a une transmission par antenne a usage multiple Download PDF

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Publication number
WO2002080375A2
WO2002080375A2 PCT/IB2002/000939 IB0200939W WO02080375A2 WO 2002080375 A2 WO2002080375 A2 WO 2002080375A2 IB 0200939 W IB0200939 W IB 0200939W WO 02080375 A2 WO02080375 A2 WO 02080375A2
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WO
WIPO (PCT)
Prior art keywords
signals
antenna
transmitter
block code
transmitted
Prior art date
Application number
PCT/IB2002/000939
Other languages
English (en)
Other versions
WO2002080375A3 (fr
Inventor
Kiran Kuchi
Yrjo Kaipanen
Markku Kuusela
Olav Trikkonen
Ari Hottinen
Original Assignee
Nokia Corporation
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US09/819,573 external-priority patent/US6748024B2/en
Application filed by Nokia Corporation filed Critical Nokia Corporation
Priority to CA002440033A priority Critical patent/CA2440033C/fr
Priority to AU2002251395A priority patent/AU2002251395A1/en
Priority to KR1020037012490A priority patent/KR100914233B1/ko
Priority to EP02720328A priority patent/EP1512256A4/fr
Priority to BRPI0208208-0A priority patent/BR0208208A/pt
Priority to JP2002578663A priority patent/JP2005503045A/ja
Priority to MXPA03008030A priority patent/MXPA03008030A/es
Publication of WO2002080375A2 publication Critical patent/WO2002080375A2/fr
Publication of WO2002080375A3 publication Critical patent/WO2002080375A3/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L1/00—Arrangements for detecting or preventing errors in the information received
    • H04L1/02—Arrangements for detecting or preventing errors in the information received by diversity reception
    • H04L1/06—Arrangements for detecting or preventing errors in the information received by diversity reception using space diversity
    • H04L1/0618—Space-time coding
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/06—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station
    • H04B7/0613—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission
    • H04B7/0667—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of delayed versions of same signal
    • H04B7/0669—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission of delayed versions of same signal using different channel coding between antennas
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/06—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station
    • H04B7/0613—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission
    • H04B7/0682—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas at the transmitting station using simultaneous transmission using phase diversity (e.g. phase sweeping)
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J13/00—Code division multiplex systems
    • H04J13/0007—Code type
    • H04J13/004—Orthogonal
    • H04J13/0048—Walsh
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J13/00—Code division multiplex systems
    • H04J13/0074—Code shifting or hopping

Definitions

  • This invention relates to a method and apparatus for achieving transmit diversity in telecommunication systems and, more particularly, to a method and apparatus for non-zero complex weighting and space-time coding signals for transmission on multiple antennas.
  • Time-varying multi-path fading is an effect in wireless systems whereby a transmitted signal propagates along multiple paths to a receiver causing fading of the received signal due to the constructive and destructive summing of the signals at the receiver.
  • Several methods are known for overcoming the effects of multi-path fading, such as time interleaving with error correction coding, implementing frequency diversity by utilizing spread spectrum techniques, or transmitter power control techniques.
  • Each of these techniques has drawbacks in regard to use for third and fourth generation wireless systems. Time interleaving may introduce unnecessary delay, spread spectrum techniques may require large bandwidth allocation to overcome a large coherence bandwidth, and power control techniques may require higher transmitter power than is desirable for sophisticated receiver-to-transmitter feedback techniques that increase mobile terminal complexity. All of these drawbacks have negative impact on achieving the desired characteristics for third and fourth generation mobile terminals.
  • Antenna diversity is another technique for overcoming the effects of multi-path fading in wireless systems.
  • two or more physically separated antennas are used to receive a transmitted signal, which is then processed by combining and switching to generate a received signal.
  • a drawback of diversity reception is that the physical separation required between antennas may make diversity reception impractical for use on the forward link in the new wireless systems where small mobile terminal size is desired.
  • a second technique for implementing antenna diversity is transmit diversity. In transmit diversity a signal is transmitted from two or more antennas and then processed at the receiver by using e.g. maximum likelihood sequence estimator (MLSE), minimum mean square error (MMSE) receivers, Maximum-a Posteriori receivers, or their approximations. Transmit diversity has more practical application to the forward link in wireless systems in that it is easier to implement multiple antennas in the base station than in the mobile terminal.
  • MSE maximum likelihood sequence estimator
  • MMSE minimum mean square error
  • Transmit diversity has more practical application to the forward link in wireless systems in that it is easier to implement multiple
  • Alamouti has proposed a method of transmit diversity for two antennas that offers second order diversity for complex valued signals.
  • S. Alamouti "A Simple Transmit Diversity Technique for Wireless Communications, " IEEE Journal on Selected Areas of Communications, pp. 1451-1458, October 1998.
  • the Alamouti method involves simultaneously transmitting two signals from two antennas during a symbol period. During one symbol period, the signal transmitted from a first antenna is denoted by SO and the signal transmitted from the second antenna is denoted by S1. During the next symbol period, the signal -S1* is transmitted from the first antenna and the signal SO* is transmitted from the second antenna, where * is the complex conjugate operator.
  • a similar diversity transmission system may also be realized in code domain. As an example, two copies of the same symbol can be transmitted in parallel using two orthogonal Walsh codes. Similar techniques can be also used to construct a space-frequency coding method.
  • This method requires an outer code and offers second order diversity due to the STTD block (Alamouti block) and a second order interleaving gain from use of the OTD block.
  • Another rate 1 method is proposed in O. Tirkkonen, A. Boariu, and A. Hottinen, "Minimal non-orthogonality rate 1 space-time block code for 3+ Tx antennas," in Proc. ISSSTA 2000, September 2000. The method proposed in this publication attains high performance but requires a complex receiver. [0007] It would be advantageous, therefore, to have a method and apparatus that provided the advantage of transmit diversity on greater than two antennas while at the same time not greatly increasing the complexity of system design.
  • the present invention presents a method and apparatus for non-zero complex weighting and space-time coding signals for transmission on multiple antennas.
  • the method and apparatus provides expansion of an N x N' space- time block code, where N is the number of transmit paths and N' is the number of output symbols per transmit path, to a M x M' space-time block code, where M>N, generated by using repetition and non-zero complex weighting of the symbols within the N x N' space time block code, to allow transmission of the space time block code on a number M of diversity transmit paths.
  • the diversity transmit paths may comprise separate antennas or beams.
  • the temporal length of the larger code M' may equal the temporal length of the original code, N'.
  • a transform is performed on an input symbol stream, to generate a transform result comprising a space-time block code.
  • the N output streams of the space-time block code, each consisting of N' output symbols, are then repeated and at least one of the repeated streams non-zero complex weighted over time to generate M streams of N' output symbols for transmission on M diversity transmit paths.
  • the non-zero complex weighting may include phase shifting.
  • N is at least 2 and M is at least 3. At least two of the N streams of N' output symbols, corresponding to the original N streams of N' output symbols, are then each transmitted on a first at least one antenna and at least one of the M-N non-zero complex weighted streams of N' symbols are transmitted on one of a second at least one antenna.
  • the first at least one antenna and second at least one antenna may comprise of any one of the M antennas.
  • the method and apparatus may be implemented in a transmitter having common or dedicated pilot channels that enable efficient channel estimation of the coefficients that are required to decode the space-time code.
  • the common and dedicated pilot channels may be implemented alone or both together in the transmitter.
  • training symbols are transmitted on N transmit diversity paths, making it possible to estimate the N independent diversity transmit paths.
  • a dedicated pilot channel code sequence may be multiplexed into each of the N streams of N' output symbols of the original space-time block code, to generate N streams of N' output symbols and pilot channel sequence. Repetition and non-zero complex weighting may then be applied to generate M phase shifted streams of N' symbols and pilot channel sequence.
  • At least two of the N original streams of N' output symbols and pilot channel sequence are then transmitted on one of the first at least one antenna and at least one of the M-N complex weighted streams of N' output symbols and pilot channel sequence are transmitted on one of the second at least one antenna.
  • Another way of enabling estimation of N channels is to transmit common pilot channels so that N common pilot channel are transmitted on each of the first at least one antenna, and M-N complex weighted copies of some of the N common pilot channels are transmitted on each of the second at least one antenna.
  • the complex weighting factors used for the common channels on each of the second at least one antenna are the same as the ones used to construct the M-N additional complex weighted streams of N' output symbols from the original N streams of N' output symbols.
  • the receiver may or may not know the method used to expand the N x N' space- time block code to an M x N' space-time block code, and the temporal weighting sequences employed.
  • the pilot channels may be arranged to enable estimation of at least N+1 diversity transmit paths. At least one of the N streams of N' output symbols, corresponding to the original N streams of N' output symbols, are then each transmitted on a first at least one antenna and at least one of the M-N complex weighted streams of N' symbols are each transmitted on one of a second at least one antenna. Different common pilot channels are transmitted on each of the first at least one antenna and on at least one of the second at least one antenna.
  • the receiver needs at least partial knowledge of the method used to expand the N x N' space-time block code to an M x N' space-time block code, and the temporal weighting sequences employed.
  • Complex weighting in the various embodiments may be applied by applying a periodic or random complex weighting pattern to each of the symbol streams that are complex weighted.
  • the relationship between the complex weights of the symbol streams transmitted on the various antennas may also be predefined.
  • FIG 1 a shows a block diagram of a transmitter according to an embodiment of the invention
  • FIG. 1b shows a block diagram of portions of a common pilot channel STTD transmitter according to an embodiment of the invention
  • FIG. 2 shows a block diagram of portions of a common pilot channel STTD transmitter according to another embodiment of the invention.
  • FIG. 3 shows a block diagram of portions of a dedicated pilot channel STTD transmitter according to a further embodiment of the invention
  • FIG. 4 shows a block diagram of portions of an embodiment of a receiver for use with the transmitter of FIG.1 ;
  • FIG. 5 shows a block diagram of portions of an embodiment of a receiver for use with the transmitter of FIG.2 or the transmitter of FIG. 3;
  • FIG.7 shows a block diagram of portions of an STS transmitter according to an embodiment of the invention.
  • FIG. 8 shows a block diagram of portions of an OTD transmitter according to an embodiment of the invention
  • FIG 9 shows a block diagram of portions of an embodiment of a receiver for use with the transmitter of FIG. 7;
  • FIG. 10 shows a block diagram of portions of an embodiment of a receiver for use with the transmitter of FIG. 8;
  • FIG. 11 shows a block diagram of portions of a long ST block code transmitter according to an embodiment of the invention.
  • FIG. 12 shows a block diagram of portions of a common/dedicated pilot channel STTD transmitter according to another embodiment of the invention.
  • FIG. 13 shows a block diagram of portions of a receiver for use with the transmitter of FIG. 12.
  • FIG. 14 shows a block diagram of portions of a receiver for use in power control of the transmitter of FIG. 12.
  • FIG. 15 shows a constellation defining a phase shifting pattern that may be used in various embodiments of the invention.
  • Transmitter 150 includes input 152 for receiving an input symbol stream, block code processor 154 for performing a transform on the input symbol stream to generate a transform result representable by an orthogonal space time block code and outputting 2 streams of symbols of the transform result, non-zero complex weighter 156 for non-zero complex weighting a first one of the two streams of symbols, a non-zero complex weighter 158 for non-zero complex weighting the second of the two streams of symbols, an RF transmitter 160 for transmitting the first stream of symbols on Ant. 1 , RF transmitter 162 for transmitting the non-zero complex weighted stream of symbols on Ant.
  • each of the 4 transmit diversity paths includes a separate antenna, Ant. 1 - Ant. 4.
  • the non-zero complex weighting may be all performed on selected ones of the transmission paths to create relative phase shifts between the transmissions on Ant. 1 and Ant. 2 or on Ant. 3 and Ant. 4.
  • non-zero complex weighting could also be applied before the inputs to RF transmitters 160 and 164, creating a non-zero complex weighted version of each of the symbol streams, but maintaining a relative phase shift between the transmitted signals.
  • An alternative of the embodiment of transmitter 150 may be implemented using less than 4 antennas, to implement the 4 diversity paths.
  • the signals input to RF transmitters 164 or 166 may be connected together and transmitted on a single antenna.
  • the non-zero complex weighing operation may be performed after RF transmitter blocks 160, 162, 164, 166, i.e., non-zero complex weighing could be implemented as a continuous phase sweep after the modulation, and baseband filtering of Space-Time coded symbols.
  • the non-zero complex weighting for these transmissions on Ant. 2 and Ant. 4 may be performed according to various alternatives.
  • Examples of this would be a phase pattern of shifts in degrees of ⁇ 0, 90, 180, 270 ⁇ on Ant. 2 and ⁇ 180, 270, 0, 90 ⁇ on Ant. 4 for 4PSK constellation.
  • FIG. 15 shows a constellation defining another phase shifting pattern that may be used in various embodiments of the invention.
  • This sequence of shifts in degrees of ⁇ 0, 135, 270, 45, 180, 315, 90, 225 ⁇ may be transmitted on antenna 2 while using the pattern of shifts in degrees of ⁇ 180, 315, 90, 225, 0, 135, 270, 45 ⁇ on antenna 4.
  • the phase shifting may be periodic or random. Periodic phase shifting refers to a predefined phase pattern for e.g. complex weight W1(t) repeated periodically.
  • the complex weights can be defined so that the sequence of complex weights defines a maximal length path, to make successive samples of the effective channel as independent as possible. This can make interleaving redundant and thus enable low delay transmission.
  • Pseudo-random phase shifting used may be a sequence of random phase selections from a MPSK constellation.
  • another non-zero complex weighting scheme where the phase difference between successive phase states is as small as possible is advantageous when estimating channel coefficients or metrics related to power control from a non-zero complex weighted channel.
  • the phase states may still cover 360 degrees during the duration of one encoding block.
  • Channel interleaving may be used in the embodiments as in conventional systems.
  • the non-zero complex weighting sequence and the interleaver jointly, so that the symbols at the output of the interleaver are as independent as possible.
  • the method can be implemented so that there is a phase shift or sweep in all antenna elements, but relative phase shifts between antennas 1 and 2, and 3 and 4 are maintained.
  • phase sweep one may have a 50 Hz phase sweep on antenna 1 and -50 Hz phase sweep on antenna 2, in order to implement a 100 Hz effective sweep.
  • antennas 3 and 4 may have a 50 Hz phase sweep on antenna 1 and -50 Hz phase sweep on antenna 2, in order to implement a 100 Hz effective sweep.
  • the phase rotation may be changed every T seconds.
  • T depends on total time duration of the data symbols and the method used for estimating the channel coefficients.
  • the phase may be kept constant for the total time duration occupied by the data symbols within at least one space-time encoding block and the corresponding dedicated or common pilot sequence/training sequence can be used to enable proper channel estimation.
  • the pilot sequence could be a walsh code, as used in CDMA systems, or sequence of training symbols with good correlation properties used for channel estimation in TDMA.
  • the pilot symbols may apply the same non-zero complex weighting coefficients as the data within the space-time block. Alternatively, the pilots may be transmitted without phase hopping.
  • the effective channel for the data can be derived jointly from the a priori known hopping pattern and the channel estimate obtained from a non-hopping channel.
  • the same or different phase pattern may be applied to both data and common pilots.
  • Channel estimation using non-hopping pilot or training sequences provides better channel estimates as the channel is more stationary.
  • Transmitter 100 may operate as a 4-antenna transmit diversity extension to release 99 of the wideband CDMA (WCDMA) third generation system standard.
  • WCDMA wideband CDMA
  • Transmitter 100 comprises input 126, block code processor 124, traffic channel symbol stream processing branch inputs 102a - 102d, antenna gain blocks 104a, - 104d, phase shifters 106a and 106b, phase shifter inputs 112a and 112b, Code multipliers 108a - 108d, pilot sequence processing branch inputs 114a - 114d, antenna gain blocks 116a -116d, code multipliers 118a - 1.18d, RF transmitter 128, including RF transmitters 128z - 128d, and antennas Ant.1 - Ant. 4.
  • Block code processor 124 performs a transform on every two received symbols S1S2 to generate a transform result comprising a 2x2 orthogonal space-time block code.
  • block code processor 124 may perform an Alamouti transform to generate the block code in the form represented by the following matrix:
  • the matrix is then divided into 4 streams of 2 symbols with each of the streams being input to one of the traffic channel symbol stream processing branch inputs 102a - 102d.
  • the stream S1S2 is input to 102a
  • S1S2 is input to 102b
  • -S2*S1 * in input to 102c
  • to -S2*S1* is input to 102d.
  • the non-zero complex weighting is performed by antenna gain blocks 104a - 104d and phase shifters 106a and 106b.
  • Antenna gain for each of the processing branches is adjusted in antenna gain blocks 104a - 104d.
  • phase shifters 106a and106b apply a phase shift to the stream S1S2 output from antenna gain block 104b and stream -S2*S1* output from antenna gain block 104d.
  • the phase shifter control blocks 112a and 112b may control phase shifters 106a and 106b by causing shifting using a continuous or discrete phase hopping pattern.
  • a CDMA scrambling code is then input to code multipliers 108a - 108d to generate the stream S1S2 to RF transmitter 128a for transmission on Ant. 1 , S1S2 (exp(j ⁇ k1)) to RF transmitter 128b for transmission Ant. 2, -S1*S2* to RF transmitter 128c for transmission on Ant.
  • the RF transmitters may perform of baseband pulse shaping, modulation, and carrier up conversion. In some implementations one may choose to apply phase hopping or sweep after baseband pulse shaping and modulation steps.
  • pilot channel sequences X1 - X4 are input to pilot sequence processing branch inputs 114a - 114d.
  • the pilot sequences are then separately processed through antenna gain blocks 116a - 116d, and code multipliers 118a - 118d.
  • the coded outputs from code multipliers 118a - 118d are then input to RF transmitters 128a - 128d, respectively, of RF transmitter 130.
  • the pilot sequence X1 is then transmit on Ant. 1
  • the pilot sequence X2 is transmit on Ant. 2
  • the pilot sequence X3 is transmit on Ant. 3
  • the pilot sequence X4 is transmit on Ant. 4.
  • FIG. 4 therein is a block diagram of portions of a receiver for use with transmitter 100 of FIG. 1 b.
  • FIG. 4 shows the signal processing for one rake finger receiver section of a receiver.
  • the received pilot sequences X1 - X4 transmit from transmitter 100 are received and input to channel estimation processing branch 402a - 402d, respectively.
  • Channel estimator 404 then performs a channel estimation function, for example a low pass filter moving average function, for each of channel 1 - channel 4.
  • the estimates of channel 1 - channel 4 are then output from outputs 406a - 406d to summer 410a, phase shifter 408a, summer 410b and phase shifter 408b.
  • Phase shifter 408a receives input from phase shifter control block 414a and shifts the estimate for channel 2 by the same phase shift used on the traffic channel symbols S1 S2 transmit from Ant. 2 in transmitter 100.
  • Phase shifter 408b receives input from phase shifter control block 414b shifts the estimate for channel 4 by the same phase shift used on the traffic channel symbols -S2* S1* transmit from Ant. 4 in transmitter 100.
  • the phase shifted version of the estimate for channel 2 is combined with the estimate for channel 1 by summer 410a, and the phase shifted version of the estimate for channel is combined with the estimate for channel 3 in summer 410b.
  • the combined estimate for channels 1 and 2 (412a) and the combined estimate for channel 3 and 4 (412b) are then input to STTD demodulator 418, which processes the received traffic signals from input 416 using the channel estimates.
  • the demodulated signal is then processed in rake combiner, deinterleaver and channel decoder 420 to generate the received symbols S1S2.
  • FIG. 2 therein is a block diagram of portions of a common pilot channel STTD transmitter 200 according to another embodiment of the invention.
  • Transmitter 200 comprises input 226, block code processor 224, traffic channel symbol stream processing branch inputs 202a - 202d, antenna gain blocks 204a, - 204d, phase shifters 206a and 206b, phase shifter inputs 212a and 212b, Code multipliers 208a - 208d, Code multiplier input 210, pilot sequence processing branch inputs 214a - 214d, antenna gain blocks 216a - 216d, phase shifters 218a and 218b, phase shifter control blocks 224a and 224b, code multipliers 220a - 220d, code multiplier input 222, RF transmitter 228, including RF transmitters 228a - 228d, and antennas Ant 1 - Ant.4.
  • the traffic channel processing and transmission in transmitter 200 is performed in the same manner as used for the traffic channel processing in transmitter 100 of FIG. 1.
  • Transmitter 200 uses common pilot channels, which are phase shifted.
  • Common pilot channel sequence P1 is input to pilot sequence processing branch inputs 214a and 214b and common pilot channel sequence P2 is input to pilot sequence processing branch inputs 214c and 214d.
  • the pilot sequences are then separately processed through antenna gain blocks 216a - 216d.
  • the pilot sequence P1 output from antenna gain block 216a is input to code multiplier 220a.
  • the pilot sequence P2 output form antenna gain block 216c is input to code multiplier 220c. Pilot sequence P1 output from antenna gain block 216b is input to phase shifter 218a.
  • Pilot sequence P2 output from antenna gain block 216d is input to phase shifter 218b.
  • Phase shifter 218a and 218b apply a phase shift under the control of phase shifter control block 224a and 224b, respectively.
  • the phase shift may be the same continuous or discrete phase hopping pattern used for the traffic channels.
  • the phase shifted pilot sequence P1 output from phase shifter 218a is then input to code multiplier 220b and the phase shifted pilot sequence P2 output from phase shifter 218b is then input to code multiplier 220d.
  • the coded pilot sequence P1 output from code multiplier 220a is then input to RF transmitter 228a for transmission on Ant. 1.
  • the coded phase shifted pilot sequence P1 output from code multiplier 220b is input to RF transmitter 228b for transmission on Ant. 2
  • the coded pilot sequence P2 output from code multiplier 220c is input to RF transmitter 228c for transmission on Ant. 3
  • the coded phase shifted pilot sequence P2 output from code multiplier 220d is input to RF transmitter 228d for transmission on Ant. 4.
  • phase shifting performed by phase shifters 218a and 218b may according to various alternatives, for example, as described for the phase shifting performed in the embodiment of FIG. 1.
  • Receiver 500 comprises channel 1 and channel 2 estimate processing branch input 502a and channel 3 and channel 4 estimate processing branch input 502b, channel estimator 504, STTD demodulator 508, traffic signal input 510 and rake combiner, deinterleaver and channel decoder 512.
  • the received pilot sequence P1 (ch1+ch20) received on channels 1 and 2 from Ant. 1 and Ant. 2, respectively, of transmitter 200 is input to input 502a.
  • the received pilot sequence P2 (ch3+ch40) received on channels 3 and
  • Channel estimator 504 performs channel estimation using, for example, a low pass filter moving average function, and outputs combined estimate for channels 1 and 2 (chest 1 ,2), and a combined estimate for channels 3 and 4 (chest 3,4).
  • the channel estimates are then input to STTD demodulator 508, which processes the received traffic signals from input 510 using the channel estimates.
  • the demodulated signal is then processed in rake, combiner, deinterleaver and channel decoder 512 to generate the received symbols S1S2.
  • FIG. 6 shows an embodiment of a rake finger of STTD demodulator 508 of FIG.
  • Transmitter 300 comprises input 318, block code processor 316, channel symbol stream processing branch inputs 302a - 302d, antenna gain blocks 304a, - 304d, phase shifters 306a and 306b, phase shifter inputs 312a and 312b, code multipliers 308a - 308d, code multiplier input 310, and antennas Ant. 1 - Ant 4.
  • Transmitter 300 of FIG. 3 is an implementation that uses dedicated pilot channels that are transmitted by embedding pilot sequences in the traffic channel symbol stream.
  • Input 318 and block code processor 316 function in the same manner as input 126 and block code processor 124 of FIG. 1.
  • pilot channel sequence U1 is input into inputs 302a and 302b multiplexed between the symbol sets of S1S2.
  • -S2*S1* is input to symbol stream processing branch inputs 302c and 302d
  • pilot channel sequence U2 is input onto inputs 302c and 302d, and multiplexed between the symbol sets of -S2*S1*.
  • Another possibility is to define 4 different dedicated pilot sequences, one for each transmit antenna
  • the multiplexed symbol streams at inputs 302a - 302d are then input to antenna gain blocks 304a - 304d, respectively.
  • Channel gain is applied in antenna gain blocks 304a - 304d.
  • the stream comprising S1S2 and pilot sequence U1 is output from antenna gain block 304a to code multiplier 308a.
  • the stream comprising S1S2 and pilot sequence U1 is output from antenna gain block 304b to phase shifter 306a, where it is phase shifted according to input from phase shifter control block 312a and then input to code multiplier 308b.
  • the stream comprising -S2*S1* and pilot sequence U2 is output from antenna gain block 304c to code multiplier 308c, and the same stream, -S2*S1* and pilot sequence, is output from antenna gain block 304d to phase shifter 306b, where it is phase shifted according to input from phase shifter control block 312b and then input to code multiplier 308d.
  • Code multipliers 308a - 308d multiply the appropriate stream by a scrambling code.
  • the code multiplied stream S1S2 and pilot sequence U1 is then input to RF transmitter 314a for transmission on Ant. 1.
  • the code multiplied phase shifted stream S1S2 and pilot sequence U1 is input to RF transmitter 314b for transmission on Ant. 2.
  • the code multiplied stream -S2*S1* and pilot sequence U2 is input to RF transmitter 314c for transmission on Ant. 3, and the code multiplied phase shifted stream -S2*S1* and pilot sequence U2 is input to RF transmitter 314d for transmission on Ant. 4.
  • RF transmitter 314a - 314d perform modulation and carrier up conversions before transmitting the streams on Ant. 1 - Ant. 4.
  • the RF transmitters may perform of baseband pulse shaping, modulation, and carrier up conversion. In some implementations one may choose to apply the non-zero weighting after baseband pulse shaping and modulation.
  • receiver 500 may be modified for use with transmitter 300 of FIG. 3.
  • receiver 500 would function similarly but inputs 502a and 502b would input U1 (Ch1+Ch20) and U2(Ch3+Ch40), respectively, to channel estimator 504c.
  • FIG. 12 therein is a block diagram of portions of a dedicated/common pilot channel STTD transmitter 1200 according to another embodiment of the invention.
  • Transmitter 1200 functions essentially in the same manner as transmitter 300 of FIG. 3 with the exception being that common pilot channels are added on Ant. 1 and Ant. 3.
  • Common pilot channel sequences P1 and P2 are input to pilot sequence processing branch inputs 1218a and 1218b, respectively.
  • the pilot sequences are then separately processed through antenna gain blocks 1220a and 1220b, and code multipliers 1222a and 1222b.
  • the coded outputs from code mu!tipliers1222a and 1222b are then input to RF transmitters 1214a and 1214c, respectively, of RF transmitter 1214.
  • the RF transmitters may perform baseband pulse shaping, modulation, and carrier up conversions. In some implementations one may choose to apply the non-zero weighting after baseband pulse shaping and modulation.
  • Transmitter 1200 of FIG. 12 provides non-hopped common pilot channels on Ant. 1 and Ant. 3 and dedicated pilot channels on Ant. 1 , Ant. 2, Ant. 3, and Ant. 4.
  • the pilot sequences may be multiplexed within one slot, for example in an embodiment where there are 15 slots in a transmission frame.
  • Antenna gains may be set different for the common and dedicated control channels.
  • the antenna gains can also be time varying.
  • Receiver 1300 comprises channel 1 and channel 2 processing branch having inputs 1302a and 1302b, and channel 3 and channel 4 processing branch having inputs 1302c and 1302d.
  • Channel estimator 1306 performs channel estimation using, for example, a low pass filter having average function, and outputs a combined estimate for channels 1 and 2 (chest 1 ,2) 1308a, and a combined estimate for channels 3 and 4 (chest 3,4) 1308b.
  • the channel estimates are then input to STTD demodulator 1310, which processes the received traffic signals from input 1312 using the channel estimates.
  • the demodulated signal is then processed in rake, combiner, deinterleaver, and channel decoder 1314 to generate the received symbols S1 , S2.
  • Receiver 1400 includes channel estimator 1402, channel estimating branch inputs 1404a-1404d, phase shifter inputs 1408a and 1408b, phase shifter 1406a and 1406b, channel estimate output 1410a and 1410b, squaring blocks 1412a and 1412b, and power control processor 1414.
  • Channel estimator 1402 computes channel coefficients from the common or dedicated channels from, for example transmitter 1200, for all four antennas during a given slot "t". This may be a channel prediction for slot t+1 , alternatively the channel estimate for slot t may be used in slowly fading channels . These channel coefficients are denoted by chanest#1 (t), chanest#2(t), chanest#3(t), and chanest#4(t) at inputs 1404a-1404d, respectively. For multiple rake fingers, e.g. chanest#1(t) is a vector channel estimate corresponding to all rake fingers from Ant. 1.
  • chanest#l2(t + 1) chanest#l(t) + chanest#2(t)e ⁇ M
  • chanest#34(t + ⁇ ) chanest#3(t) + chanest#4(t)e ⁇ 34(M) (2)
  • Received signal power estimate for slot (t+1) can be done based on chanest #12(1*1) and chanest #12(t+1); received _ power(t + 1) + 1)
  • a power control command is generated by processor 1414 using the received power estimate.
  • FIG.7 therein is a block diagram of portions of an space time spreading (STS) transmitter 700 according to an embodiment of the invention.
  • STS space time spreading
  • Transmitter 700 is a STS embodiment of transmitter 150 of FIG. 1a in which the space time block processor performs the transform in the Walsh Code domain.
  • the STS block code matrix used may be represented as:
  • each row of the matrix and its phase shifted version are each transmitted on separate antennas Ant. 1 - Ant. 4.
  • the symbols S1 and S2 in each row are each transmitted simultaneously over two symbol periods, rather than sequentially.
  • Data symbols are input to transmitter 700 at input 718 of channel coder 720.
  • Channel coder 720 codes, punctures, interleaves, and formats the input data symbols and outputs every other coder output symbol S1 as even data and every other coder output symbol S2 as odd data.
  • the even data is then processed through symbol repetition blocks 702a,b,e,f, Walsh function blocks 704b and 704d, Walsh multipliers 706a,b,e,f, summers 708a - 708d and complex adders 710a and 710b.
  • the odd data is processed through symbol repetition blocks 702c,d,g,h, Walsh function blocks 704b and 704d, Walsh multipliers 706c,d,g,h, summers 708a - 708d, and complex adders 710a and 710b.
  • the result at the output of complex adder 710a is the matrix row
  • SW l - S2 * W 2 is then input to complex multiplier 712a to generate
  • Transmitter 700 comprises input 912, Walsh function blocks 902b and 902d, Walsh multipliers 902a and 902c, channel multipliers 904a - 904d, complex adders 906a and 906b, multiplexer (Mux) 908, and output 910.
  • a received input signal is received at input 912, is processed by the STS demodulator.
  • the pilot channel transmission and channels estimation procedures can be same as explained in STTD case.
  • the channel estimates 904c and 904b can be same as 412a, 412b from Fig. 4 for non-hopping common pilot channel case.
  • the channel estimates can be obtained from channel estimation block 504 from Fig. 5. These channel estimates are input to the STS demodulator in Fig. 9 as hi and h2. hi corresponds to combined channel estimate from Ant. 1 , Ant. 2 and h2 corresponds to channel estimate from Ant. 3, Ant 4. After STS demodulation using 902a,b,c,d and 904a,b,c,d, and 906a, b the output from 908 is the STS demodulated signal to be sent to rake combiner, deinterleaver, and channel decoder block 512 from Fig5.
  • Transmitter 800 comprises input 822, channel coder 820, symbol repetition blocks 802a - 802d, Walsh function blocks 804a and 804b, Walsh multipliers 806a - 806d, complex adders 808a - 808b, complex multipliers 810a and 810b, RF transmitters 812a - 812d.
  • Transmitter is an orthogonal transmit diversity (OTD) embodiment of transmitter 150 of FIG. 1a in which the space time block processor performs the transform in the Walsh code domain.
  • OTD block code matrix used may be represented as:
  • each row of the matrix and its phase shifted version are each transmitted on separate antennas Ant. 1 - Ant. 4.
  • Data symbols are input to transmitter 800 at input 822 of channel coder 820.
  • Channel coder 820 codes, punctures, interleaves, and formats the input data symbols and outputs every other coder output symbol S1 as even data and every other coder output symbol S2 as odd data.
  • the even data is then processed through symbol repetition blocks 802a and 802b, Walsh function block 804a, Walsh multipliers 806a and 806b, and complex adder 808a.
  • the odd data is processed through symbol repetition blocks 802c and 802d, Walsh function block 804b, Walsh multipliers 806c and 806d, and complex adder
  • Transmitter 800 comprises input 1010, Walsh function blocks 1002a and 1002b, Walsh multipliers 1010a and 1010b, multipliers 1004a and 1004b, multiplexers 1006 and output 1008.
  • a received input signal is received at input 912 is demodulated using a OTD demodulator 1000 using the knowledge of channel coefficients hi* and h2*.
  • the channel coefficients hi and h2 for this OTD block are derived in the same as explained in Fig4 and Fig5.
  • the OTD demodulator 1000 is implemented using 1010, 1010a,b and 1012a,b and 1004a,b and 1006.
  • the OTD demodulated output 1008 is sent to rake combiner, deinterleaver, and channel decoder block 512 from Fig5.
  • Transmitter 1100 comprises input 1118, 1120, symbol stream processing branch inputs 1116a - 1116d, time reversal blocks 1102 and 1104, complex conjugate blocks 1106a and 1106b, multiplier 1108, phase multiplier 1110a and 1110b, phase multiplier control blocks 1112a and 1112b, and antennas Ant. 1 , Ant. 2, Ant. 3, and Ant. 4.
  • Channel coder 1120 codes, punctures, interleaves, and formats a symbol stream received at input 1118.
  • Channel coder 1120 also splits the input symbol stream into odd and even data streams.
  • the even data stream is input to branch input 1116a and RF transmitter 1122a for transmission on Ant. 1 during the first half of a data burst and the odd data stream is input to branch input 1116c and RF transmitter 1112c for transmission on Ant. 2 during the first half of the data burst.
  • the even data stream is input to branch input 116b, time received on time reversal block 1102, complex conjugated in complex conjugate block 1106a and sent to RF transmitter 1122c for transmission on Ant. 3.
  • the odd data stream is input to branch input 1116d, time reversed in time reversal block 1104, complex conjugated in complex conjugate block 1106b, multiplied by a negative in multiplier 1108 and sent to RF transmitter 1122d for transmission on Ant. 4 during the second half of the data burst.
  • a training sequence SEQ1 is embedded in the middle of the burst transmit on Ant. 1 and a training sequence SEQ2 is embedded in the middle of burst transmit on Ant. 2.
  • Phase multipliers 1112a and 1112b phase shifts the inputs to RF transmitters 1122b and 1122d, using the multiplication blocks 1110a and 1110b respectively.
  • the output of phase multiplier 1112a is then input to RF transmitter 1122b for transmission on Ant.
  • the RF transmitters may perform of baseband pulse shaping, modulation, and carrier up conversion. In some implementations one may choose to apply the phase multiplication after baseband pulse shaping and modulation steps.
  • phase rotation applied in phase multipliers 1122a and 1122b is kept constant during the burst length, with the phase being changed on a burst by burst basis.
  • the phase can be chosen periodically or randomly from a MPSK constellation as explained previously.
  • the phase rotation on the Ant. 4 is kept same as phase rotation on ant2 with a 180 degree shift or multiplied by -1.
  • the phase multiplication may be done before or after the base band pulse shaping.
  • the transmission on Ant. 1 and Ant. 3 may be intercharged.
  • the transmitter shown FIG. 3 can also be applied to EDGE with some modification.
  • the space-time code described in 316 is applied blockwise instead of symbol wise for an EDGE application.
  • the block length can be chosen as first half of the burst. In EDGE the length of the first half and second half of the bursts are equal to 58 symbols.
  • S1 and S2 denotes a block of symbols and ()* denotes time reversal of a block of symbols and complex conjugation operation.
  • S1* denotes the block of symbols S1 is time reversed and complex conjugated.
  • -S2* denotes that the block of symbols S2 is time reversed, complex conjugated and multiplied by -1.0.
  • the pilot sequences U1 and U2 can be chosen as two training sequences such as well-known CAZAC sequences.
  • the spreading codes 308a,b,c,d will not be applied in EDGE.
  • the phase multiplication blocks 306a and 306b are retained.
  • a receiver designed for a 2-antenna space time block code may be used as a receiver for the embodiments of FIG. 1 or FIG. 2.

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  • Signal Processing (AREA)
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Abstract

La présente invention concerne un procédé et un système de pondération complexe non nulle et de codage spatio-temporel de signaux destinés à une transmission sur des antennes à usage multiple (160,162,164,166). Le procédé et le système de l'invention effectuent l'extension d'un code de bloc spatio-temporel N x N' à un code de bloc spatio-temporel M x M' (M>N) en utilisant une pondération complexe non nulle des symboles intégrés au code de bloc spatio-temporel N x N' afin d'assurer la transmission du code de bloc spatio-temporel à un nombre d'antennes à réception simultanée supérieur à N'. Un résultat de M antennes à réception simultanée peut être obtenu pour M antennes d'émission.
PCT/IB2002/000939 2001-03-28 2002-03-26 Code spatio-temporel de ponderation complexe non nulle destine a une transmission par antenne a usage multiple WO2002080375A2 (fr)

Priority Applications (7)

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CA002440033A CA2440033C (fr) 2001-03-28 2002-03-26 Code spatio-temporel de ponderation complexe non nulle destine a une transmission par antenne a usage multiple
AU2002251395A AU2002251395A1 (en) 2001-03-28 2002-03-26 Non-zero complex weighted space-time code for multiple antenna transmission
KR1020037012490A KR100914233B1 (ko) 2001-03-28 2002-03-26 다중 안테나 전송을 위한 0이 아닌 복소수 가중치가부과된 시공간 코드
EP02720328A EP1512256A4 (fr) 2001-03-28 2002-03-26 Code spatio-temporel de ponderation complexe non nulle destine a une transmission par antenne a usage multiple
BRPI0208208-0A BR0208208A (pt) 2001-03-28 2002-03-26 método e aparelho para codificar os sinais em salto de fase e especial temporal para transmissão por uma pluralidade de antenas
JP2002578663A JP2005503045A (ja) 2001-03-28 2002-03-26 複数アンテナ送信用の非ゼロ複素重み付けした空間−時間符号
MXPA03008030A MXPA03008030A (es) 2001-03-28 2002-03-26 Codigo de espacio-tiempo de ponderacion compleja no nula para transmision en antenas multiples.

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US09/819,573 US6748024B2 (en) 2001-03-28 2001-03-28 Non-zero complex weighted space-time code for multiple antenna transmission
US09/819,573 2001-03-28
US10/078,840 US6816557B2 (en) 2001-03-28 2002-02-20 Non-zero complex weighted space-time code for multiple antenna transmission
US10/078,840 2002-02-20

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US8068570B2 (en) 2003-07-21 2011-11-29 Broadcom Corporation Weight generation method for multi-antenna communication systems utilizing RF-based and baseband signal weighting and combining based upon minimum bit error rate
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US11916627B2 (en) 2010-12-10 2024-02-27 Sun Patent Trust Signal generation method and signal generation device
RU2651805C2 (ru) * 2012-05-22 2018-04-24 Йен Стюарт Макрае Нигель Способ и система передачи определенных данных на одной частоте с использованием нескольких линейных поляризованных сигналов
EP3104637A4 (fr) * 2014-02-05 2017-09-13 Mitsubishi Electric Corporation Station de base, système de communication et procédé de communication

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CA2440033C (fr) 2007-01-09
AU2002251395A1 (en) 2002-10-15
JP2005503045A (ja) 2005-01-27
MXPA03008030A (es) 2005-04-29
CN1611047A (zh) 2005-04-27
BR0208208A (pt) 2006-12-12
RU2003131399A (ru) 2005-05-27
CN100536450C (zh) 2009-09-02
RU2276463C2 (ru) 2006-05-10
CA2440033A1 (fr) 2002-10-10
EP1512256A4 (fr) 2009-12-02
EP1512256A2 (fr) 2005-03-09
WO2002080375A3 (fr) 2005-01-13

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