Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L5/00—Arrangements affording multiple use of the transmission path
  • H04L5/003—Arrangements for allocating sub-channels of the transmission path
  • H04L5/0044—Arrangements for allocating sub-channels of the transmission path allocation of payload
  • H04L5/0046—Determination of how many bits are transmitted on different sub-channels
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L5/00—Arrangements affording multiple use of the transmission path
  • H04L5/0091—Signaling for the administration of the divided path
  • H04L5/0094—Indication of how sub-channels of the path are allocated
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L5/00—Arrangements affording multiple use of the transmission path
  • H04L5/0001—Arrangements for dividing the transmission path
  • H04L5/0003—Two-dimensional division
  • H04L5/0005—Time-frequency
  • H04L5/0007—Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT

Definitions

• the present invention relates broadly to telecommunications systems and methods. More particularly, the present invention relates to a handshake for an xDSL (Digital Subscriber Line type) modem.
• xDSL Digital Subscriber Line type
• Digital subscriber line (DSL) systems are a new and fast- growing data transmission service which provide significantly higher data rates than conventional V.34 and V.90 type modems.
• the abbreviation "xDSL” is an integrated designation for different DSL services including ADSL (asymmetric DSL) , SDSL (symmetric DSL) , RADSL (rate-adaptive DSL) , HDSL (high speed DSL) , and VDSL (very high speed DSL) , UDSL (universal DSL) , and their modifications such as ADSL-LITE (also known as G.lite).
• the xDSL services typically provide data rates of several Mbits/s downstream and several hundred Kbits/s upstream, although SDSL provides the same upstream and downstream rates .
• DSL discrete multitone
• G.hs handshake procedure
• the requirements for G.hs are set forth in several documents such as "Proposal for G.hs Modulation Technique and Message Protocol", ITU-T Telecommunication Standardization Sector. Cl-068 Chicago, USA 6-9 April 1998, and "Handshake procedures for Digital Subscriber Line (DSL) transceivers", ITU-T Draft G.994.1 (February 3, 1999) which are both hereby incorporated by reference herein in their entireties.
• the main requirements of the handshake are: transmission of several tens of bytes during the handshake; signal compatibility with all types of DSL receivers; and interworking with the plain old telephone service (POTS) , the integrated services digital network (ISDN) , and time compression multiplexing ISDN (TCM-ISDN) .
• POTS plain old telephone service
• ISDN integrated services digital network
• TCM-ISDN time compression multiplexing ISDN
• Meeting these main requirements is not a trivial task because of considerable noise and crosstalk impairments, and lack of knowledge regarding the frequency characteristics of the channel, all of which is described in various papers such as: Matsushita Electric Industrial Co. Ltd, "Proposed Working Text for G.hs Based on V.8bis", ITU- Telecommunication Standardization Sector. NF-044. Nice, France, 11-14 May 1998; Matsushita Electric Industrial Co.
• signal attenuation across lines carrying xDSL signals is a non-monotonic function of frequency, and may have several deep notches, while noise power spectral density (PSD) is also not a flat function of frequency.
• PSD noise power spectral density
• SNR signal to noise ratio
• the SNR is subject to random and cyclic variations in time.
• FEXT far-end cross-talk
• NEXT near-end cross-talk
• Another object of the invention is to provide modems and methods for implementing the above-listed objects.
• handshake information for xDSL services are transmitted utilizing a spread spectrum modulated system where a plurality (n) of carrier tones (n > 2) are summed and utilized as a spread spectrum carrier (SSC) , and data is modulated onto the carrier (at all utilized frequencies) .
• SSC spread spectrum carrier
• PSK phase shift keying
• BPSK - binary PSK or DBPSK - differential binary PSK
• the SSC is transmitted with sign "+" if the handshake bit is a +1 and with sign "-" if the handshake bit is a "-1".
• DPSK the same modulation procedure is used for differentially encoded handshake bits.
• the handshake symbol rate is set equal to .8A symbols/msec, where A is a positive integer.
• symbols are preferably repeated at least four times.
• a preamble can be provided for timing recovery purposes.
• Further aspects of the invention include different receiver systems, including a quasicoherent receiver, an autocorrelation receiver, and a presently preferred incoherent receiver which utilizes coherent accumulation of FFT components for a DBPSK spread spectrum handshake signal.
• Fig. 1 is a block diagram of the preferred transmitter of the invention.
• Fig. 2 is a diagram showing the signal structure of the preferred handshake signal of the invention.
• Fig. 3a is a block diagram of an autocorrelation receiver of DBPSK signals according to the invention.
• Fig. 3b is a block diagram of a quasicoherent receiver of DBPSK signals according to the invention.
• Fig. 3c is a block diagram of an incoherent receiver which utilizes coherent accumulation of FFT components for a DBPSK spread spectrum handshake signal according to the invention.
• handshake information for xDSL services is transmitted by modulating the handshake information on a spread spectrum carrier (SSC) , where the SSC is a sum of tones conventionally used by xDSL for the data transmission mode.
• the transmitter 10 includes a phase initialization (PI) unit 15, an inverse fast Fourier transformation (IFFT) unit 20, a spread spectrum carrier (SSC) memory 25, a modulator 30, a differential encoder 35 and a block frame unit 40.
• the phase initialization unit 15 generates complex numbers indicating a desirable amplitude and initial phase distribution for a plurality of multitone signals.
• the amplitude distribution is chosen to be flat (uniform) .
• the initial phases of different tones are generated randomly or selected specifically in order to minimize the crest-factor of the generated tones.
• the IFFT transforms a set of complex numbers into a set of time-domain samples which are stored in memory 25. If, for example, all or substantially all two hundred fifty-six DMT tones (such as might be utilized in ITU-T Standard G.992.2) are generated by the PI unit 15, a five hundred twelve sample set may be stored in the memory 25. Additional repetitive samples may also be stored in the memory, if desired as a prefix which can be used by the receiver to reduce distortion. If desired, the samples may be generated in other manners (e.g., without the PI and IFFT, or in another apparatus) and loaded and stored in the transmitter memory for use as described below.
• a carrier may be considered a spread spectrum carrier if three or more distinct tones are modulated together.
• the SSC for a down-stream connection may contain a full or partial set of down-stream tones
• the SSC for an up-stream connection could contain a full or partial set of up-stream tones.
• a G.Lite ADSL up-stream SSC might utilize allowed tones from the set six through thirty-two (25.875 kHz...
• the downstream SSC might utilize allowed tones from the set thirty-three through one hundred twenty-eight (142.3125 kHz ... 552 kHz) .
• the SSC may contain only even or odd tones to reduce the processing at the receiver.
• Handshake information (as described below with reference to Fig. 2) which is to be modulated onto the spread spectrum carrier is provided to the differential encoder 35 and differentially encoded bits are written to the block frame unit 40.
• the handshake information is provided to the differential encoder at a speed of .8 kbps, and differentially encoded 4-bit subblocks are written into registers of the block frame unit 40.
• each 4-bit subblock is read four times such that each block frame is provided to the modulator 30 with a speed of 3.2 kbps.
• the modulation technique is preferably is a binary phase shift keying (BPSK) .
• the modulation technique is preferably a differential BPSK.
• the modulator 30 uses the output of the block framer unit 40 to select whether the samples stored in the memory 25 are to be transmitted as is, or inverted (i.e., multiplied by -1 or 180 degrees out of phase) .
• the samples stored in the memory 25 are sequentially read out of the memory so that all samples are modulated (i.e., transmitted as is or inverted) at the proposed symbol rate discussed below.
• the SSC samples are transmitted with sign "+” if the handshake bit is a "+1", and transmitted with sign "-” if the handshake bit is "-1" (or vice versa) .
• DBPSK the same modulation procedure is used for differentially encoded handshake bits.
• BPSK or DBPSK modulation is preferred, other modulation techniques such as QPSK (quadrature PSK) , DQPSK (differential QPSK) , frequency modulation, amplitude modulation, and quadrature amplitude modulation could be utilized.
• QPSK quadrature PSK
• DQPSK differential QPSK
• the handshake includes a preamble and a G.hs message.
• the preamble comprises N subblocks of a distinct four bit sequence "1,1,1,-1” followed by four subblocks of a four bit divider sequence "-1,-1,-1,-1", followed by eight subblocks of a pseudorandom sequence (as specified) .
• Each subblock is preferably generated at a 1.25 millisecond rate (i.e., each subblock has a duration of 1.25 ms) , with bits being generated at a .3125 millisecond rate.
• the G.hs message is provided and preferably includes N blocks which are generated at a 5 millisecond rate.
• Each block preferably includes four subblocks of four information bits (symbols) each (bl, b2, b3, b4) , with the four information bits being repeated four times (i.e., each subblock within the block contains the same material).
• Each symbol carries one information bit. So each block of duration 5 milliseconds, carries four information bits with redundancy 3/4.
• each bit of the preamble and G.hs message is preferably modulated onto a spread spectrum carrier.
• the preamble is preferably provided to permit the receiver to detect G.hs transmission, to recover the spread spectrum carrier for coherent processing, and for symbol and block synchronization (timing recovery) . While the preamble is preferably modulated, an unmodulated preamble (all +ls) can be utilized.
• each symbol of the G.hs message at least four times, at least two symbol time-separated blocks will occur within the 1.25 ms high SNR FEXT areas in a TCM-ISDN cross-talk environment .
• a noiseless time window may be found by calculating the correlation between N-symbol blocks delayed by 2.5 ms relative to each other. If the delayed blocks coincide with each other (i.e., they have not been corrupted by noise), the time window has a "high enough" SNR (i.e., it is "noiseless” for the purpose of the handshake) and can be used for receiving the handshake message.
• the structure of the preamble is particularly arranged to permit this determination.
• the noiseless time window has a random time position relative to the transmission of the preamble and handshake message
• received N-symbol blocks may be cyclically shifted.
• the block frame may not correspond to the noiseless time window frame. It is therefore preferred that this shift be estimated and eliminated.
• the cyclic shift may be estimated and eliminated by transmitting an N-symbol reference block.
• the preamble is provided with a series of reference blocks having the form "1,1,1,-1". It should be appreciated that any shift of the reference block will be distinct (-1,1,1,1; 1,-1,1,1; 1,1,-1,1) and detectable, and may therefore be detected and eliminated at the receiver. This pattern therefore allows for symbol synchronization and subblock synchronization.
• the autocorrelation receiver 100a for DBPSK spread spectrum handshake signals is seen in Fig. 3a.
• the autocorrelation receiver 100a includes an autocorrelation demodulator 102a, a timing signal extractor 103a, and preferably further includes a noiseless time window (TW) determination unit 104a and a transmitted bit selection (BS) unit 106a.
• the autocorrelation demodulator 102a includes a delay line (DL) 110a, a multiplier 112a, a low pass filter (LPF) 114a, and a binary slicer (Sgn) 116a.
• DL delay line
• LPF low pass filter
• Sgn binary slicer
• Incoming SSC modulated signals are provided to the delay line 110a and the multiplier 112a.
• the delay ⁇ t of the delay line is preferably set equal to 1/.8A ms (i.e., the handshake symbol duration).
• the multiplier 112a multiplies the incoming signal with the delayed signal.
• the output of the low pass filter 114a reflects the modulation function in the transmitter, and the sign function of the low pass filter output, as generated by the binary slicer 116a which compares the output to a zero threshold, corresponds to the transmitted bits.
• the autocorrelation receiver 100a calculates (at the multiplier 112a) a sealer product (S n (t) *S n _ 1 (t) ) between a given spread spectrum signal S n (t) and a previous spread spectrum signal S n _ x (t) .
• the binary slicer 116a requires timing information which is preferably extracted from the low pass filter output by bandpass filtering of a frequency component responding to the baud (symbol) frequency.
• the timing information can be extracted from the incoming signal by a variety of well-known methods; e.g., as taught in Jan W. M. Bergmans, Digital Baseband Transmission and Recording. Chapters 9 and 10, “Basics of Timing Recovery”, and “A Catalog of Timing Recovery Schemes", Kluwer Academic Publishers, Boston (1996) pp. 451-587.
• the autocorrelation demodulator 102a in conjunction with the timing extractor 103a suffices as a G.hs receiver in situations which do not require carrier recovery or other special synchronization, additional circuitry can be utilized if desired.
• the channel noise has a steady power spectral density, robustness can be increased by accumulating signals at the output of the low pass filter, taking into account that every symbol may be repeated several times.
• the spread spectrum signal may be passed through a corresponding filter (not shown) at the input of the receiver in order to emphasize components of the spread spectrum signal having a higher SNR.
• a noiseless time window determination unit 104a can be provided to compare the signal subblocks containing N symbols and delayed relative to each other by 2.5 ms. If the delayed N bit combination coincides within a certain time window, it indicates that this window has a sufficiently high SNR and can be used for receiving handshake bits. Regardless, the window determination unit 104 finds the time window of interest and generates an output signal indicating the time position of the desired window which is provided to the bit selection unit 106a. The demodulated bits provided at the output of the slicer during the noiseless window are also provided to the bit selection unit 106a, which determines from the bits and the window information the cyclic shift in effect. Thus, during receipt of the G.hs message, the bit selection unit 106a selects the correct portion of the received bits and eliminates the cyclic shift in the received information blocks. The bit selection unit 106a produces for output N bits every 5 milliseconds.
• the quasicoherent receiver 100b includes an autocorrelation demodulator 102b, a timing signal extractor 103b, and preferably further includes a noiseless time window determination unit 104b and a transmitted bit selection unit 106b.
• the quasicoherent demodulator 102b includes a spread spectrum recovery (SSCR) unit 111b, a multiplier 112b, a low pass filter 114b, a binary slicer 116b, a delay line 118b, and a sign multiplier 120b.
• SSCR spread spectrum recovery
• Incoming SSC modulated signals are provided to the spread spectrum carrier recovery unit 111b and the multiplier 112b.
• the spread spectrum carrier recovery unit 111b accumulates SSC samples during the preamble and extracts a spread spectrum reference signal R(t) therefrom.
• the multiplier 112b multiplies the incoming signal with the output of the SSC recovery unit.
• the output is forwarded to the low pass filter 114b which is preferably provided with a frequency bandwidth ⁇ f approximately equal to N/1.25 kHz.
• the output of the low pass filter 114b is fed to slicer 116b which compares the output to a threshold (typically zero) .
• the output of slicer 116b is a binary signal which is fed to the delay line 118b and to the sign multiplier 120b.
• the sign of the output of the sign multiplier 120b corresponds to the transmitted bits.
• the average unmodulated SSC preferably extracted from the preamble by the SSC recovery unit 111b, is used as a spread spectrum reference signal R(t) for the coherent demodulation.
• R(t) the spread spectrum reference signal
• J n _ ! sgn (S ⁇ (t) *R(t) )
• the quasicoherent receiver 100b provides excellent results, but is substantially more complicated to implement than the autocorrelation receiver 100a because of the SSC recovery unit 111b.
• timing signal extractor 103b and the time window determination unit 104b and bit selection unit 106b of the quasicoherent receiver 100b are substantially as described above with respect to corresponding elements of Fig. 3a.
• the incoherent receiver includes a fast Fourier transform block 130, a quadrature component accumulation unit 135, a multichannel incoherent demodulator 140, a DMT accumulation unit 145, and a binary slicer 150.
• the FFT block 130 receives the time domain handshake signal and converts the signal into a frequency domain signal.
• the output of the FFT block are signals F cnkm and F snkm which are respectively, the real and complex parts for the k-th DMT tone at the m-th DMT symbol interval of the n-th handshake symbol.
• the quadrature component accumulation (QCA) unit 145 separately sums the real parts together and the imaginary parts together according to F snkm .
• component accumulation unit 145 are then demodulated by the incoherent demodulator 140 according to
• F nk F cnk * F c(n _ 1)k + F snk * F s(n _ 1)k .
• the output of the DMT accumulator 145 is provided to the binary slicer 150 in order to compare the output F n to a zero threshold.
• the decoded binary symbol I n sgn (F n ) .
• the incoherent receiver 100c is relatively simple to implement because it is based on the use of a FFT which is already available in DMT-based systems. In addition, no frequency equalization (carrier phase recovery) is required, and the performance of the incoherent receiver 100c is nearly as good as the quasicoherent receiver 100b of Fig. 3b.

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• 1999
  • 1999-06-18 EP EP99931832A patent/EP1090463A1/de not_active Withdrawn
  • 1999-06-18 CA CA002332941A patent/CA2332941A1/en not_active Abandoned
  • 1999-06-18 JP JP2000556452A patent/JP2002519886A/ja active Pending
  • 1999-06-18 WO PCT/US1999/013817 patent/WO1999067890A1/en not_active Application Discontinuation
  • 1999-06-18 AU AU48255/99A patent/AU4825599A/en not_active Abandoned

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