Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
  • H04N7/00—Television systems
  • H04N7/12—Systems in which the television signal is transmitted via one channel or a plurality of parallel channels, the bandwidth of each channel being less than the bandwidth of the television signal
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
  • H04N21/00—Selective content distribution, e.g. interactive television or video on demand [VOD]
  • H04N21/20—Servers specifically adapted for the distribution of content, e.g. VOD servers; Operations thereof
  • H04N21/23—Processing of content or additional data; Elementary server operations; Server middleware
  • H04N21/238—Interfacing the downstream path of the transmission network, e.g. adapting the transmission rate of a video stream to network bandwidth; Processing of multiplex streams
  • H04N21/2383—Channel coding or modulation of digital bit-stream, e.g. QPSK modulation
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
  • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
  • H04L25/03006—Arrangements for removing intersymbol interference
  • H04L25/03343—Arrangements at the transmitter end
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
  • H04L25/40—Transmitting circuits; Receiving circuits
  • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
  • H04L25/497—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems by correlative coding, e.g. partial response coding or echo modulation coding transmitters and receivers for partial response systems
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/38—Synchronous or start-stop systems, e.g. for Baudot code
  • H04L25/40—Transmitting circuits; Receiving circuits
  • H04L25/49—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems
  • H04L25/497—Transmitting circuits; Receiving circuits using code conversion at the transmitter; using predistortion; using insertion of idle bits for obtaining a desired frequency spectrum; using three or more amplitude levels ; Baseband coding techniques specific to data transmission systems by correlative coding, e.g. partial response coding or echo modulation coding transmitters and receivers for partial response systems
  • H04L25/4975—Correlative coding using Tomlinson precoding, Harashima precoding, Trellis precoding or GPRS
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
  • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
  • H04L27/3405—Modifications of the signal space to increase the efficiency of transmission, e.g. reduction of the bit error rate, bandwidth, or average power
  • H04L27/3411—Modifications of the signal space to increase the efficiency of transmission, e.g. reduction of the bit error rate, bandwidth, or average power reducing the peak to average power ratio or the mean power of the constellation; Arrangements for increasing the shape gain of a signal set
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
  • H04N21/00—Selective content distribution, e.g. interactive television or video on demand [VOD]
  • H04N21/40—Client devices specifically adapted for the reception of or interaction with content, e.g. set-top-box [STB]; Operations thereof
  • H04N21/43—Processing of content or additional data, e.g. demultiplexing additional data from a digital video stream; Elementary client operations, e.g. monitoring of home network or synchronising decoder's clock; Client middleware
  • H04N21/438—Interfacing the downstream path of the transmission network originating from a server, e.g. retrieving encoded video stream packets from an IP network
  • H04N21/4382—Demodulation or channel decoding, e.g. QPSK demodulation

Definitions

• the present invention relates to the transmission of digital data and, more particularly, to high-definition television.
• a problem arises in the context of the transmission of high-definition television (HDTV) signals in channels that are allocated for standard, so-called NTSC TV transmission.
• the problem specifically arises in areas where channels that are unused for NTSC transmission — and are thus candidates for HDTV transmission — are used for NTSC transmissions in relatively proximate areas.
• HDTV high-definition television
• the HDTV signal may be corrupted by the channel 3 NTSC signal from Philadelphia or Hartford for New York City HDTV viewers.
• the NTSC signal may be corrupted by the HDTV signal.
• Such corruption is referred to as "co-channel interference.”
• the co-channel interference from the HDTV signal to the NTSC signal can be satisfactorily dealt with by specifying in the HDTV standards (still under consideration) a sufficiently low transmit power level.
• a sufficiently low transmit power level in order to deal with co-channel interference from the NTSC signal to the HDTV signal, it has been proposed to rely on the fact than an NTSC signal is dominated by energy concentrations at particular locations in the frequency spectrum.
• an NTSC rejection filter having nulls at those frequency locations, thereby removing a significant portion of the interfering NTSC signal, while only minimally degrading the HDTV signal.
• the proposed HDTV standards envision a digital transmission format in which the bit stream representing the baseband HDTV signal is mapped, via a so- called channel code, into channel symbols each comprised of one or more so-called signal points.
• a potential problem in such arrangements is that the NTSC rejection filter creates so-called forced intersymbol interference (ISI) in the receiver, which corrupts the received signal points.
• ISI forced intersymbol interference
• an overall transmission scheme embodying the principles of the present invention comprises a channel encoder which provides a more-than- minimal amount of shaping gain in conjunction with a Tomlinson precoder.
• the invention more particularly, contemplates the use of a channel code which provides such a high degree of shaping gain of its signal points that the shaping gain of the signal points that are output by the Tomlinson precoder is at least 1.0 dB, when compared to signal points generated by a channel encoder without shaping gain and without use of the Tomlinson precoder.
• a shaping gain of 1.0 dB is significant, and is achieved only via such a large tradeoff in bandwidth efficiency that, absent the teachings of the present invention, the prior art-because of the aforementioned commonly held belief-is not led to use, in a system which includes a Tomlinson precoder, a channel code having such a large amount of shaping gain.
• FIG. 1 is a block diagram of an HDTV transmitter embodying the principles of the present invention
• FIG. 2 is a chart helpful in describing the trellis code that can be used as part of the overall channel code in the HDTV system comprised of the transmitter of FIG. 1 and the receiver of FIG. 3;
• FIG. 3 is a block diagram of an HDTV receiver adapted to receive HDTV signals generated by the transmitter of FIG. 1;
• FIGS. 4-7 describe a first illustrative trellis code
• FIGS. 8-10 describe a second illustrative trellis code
• FIG. 11 is a table comparing the characteristics of the aforementioned first and second trellis codes and a third trellis code that can be used by the HDTV system.
• FIG. 1 shows a television transmitter embodying the principles of the invention.
• a television signal-illustratively an HDTV signal- is provided by signal source 11.
• Signal source 11 includes circuitry for compressing the TV signal and putting it into an HDTV format, as well as certain standard modem-type circuitry such as a circuitry which randomizes the bit stream-the so-called "scrambler.”
• the HDTV signal thus generated is applied to concatenated encoder 13, which includes the serial combination of Reed-Solomon encoder 131; an RS symbol interleaver 134, which reorders the sequence of Reed-Solomon symbols generated by encoder 131 in order to provide protection against so-called "bursty" errors introduced either in the television channel or the receiver; N-dimensional trellis encoder 136; and N-dimensional constellation mapper 139.
• the combination of trellis encoder 136 and constellation mapper 139 implement a type of channel code referred to as "coded modulation.”
• the output of trellis encoder 136 is a sequence of data words identifying a sequence of four-dimensional symbols. Each of the four- dimensional symbols is to be transmitted in the form of a sequence of four one- dimensional signal points.
• the data words identifying the four- dimensional symbols are applied within concatenated encoder 13 to four- dimensional constellation mapper 139, whose output for each identified symbol is a sequence of four one-dimensional signal points. Since the signal points are one- dimensional, each signal point is simply represented as a signed number. There are M possible signal point values.
• Each four- dimensional symbol generated by constellation mapper 139 is comprised of a sequence of four signal points.
• Each signal point is a point in a predetermined one- dimensional base constellation.
• the one-dimensional constellation illustratively has six signal points located at coordinates -5, -3, -1, 1, 3 and 5.
• the four-dimensional symbol is delivered during a so-called symbol interval, or equivalently, four so- called signaling intervals, one signal point in each signaling interval.
• the assemblage of all four-dimensional symbols that can be output by constellation mapper 139 is referred to as the four-dimensional constellation.
• a particular one four-dimensional symbol ⁇ (3,l,-l,5)--- is represented in FIG. 2 via an enlarged signal point of each of the one-dimensional constellations.
• the signal point values generated by constellation mapper 139 are applied to signal point interleaver 15 of the type described in my U.S. Patent No. 5,056,112 issued October 8, 1991, which reorders the sequence of signal point values.
• the combination of this interleaver with a corresponding de- interleaver in the receiver advantageously causes the noise in the received signal to be whitened prior to being Viterbi decoded in the receiver as noted below.
• the interleaved signal points are applied to Tomlinson precoder 17, which compensates in advance for so-called forced intersymbol interference (ISI) that is introduced in the receiver by the receiver's NTSC rejection filter.
• ISI forced intersymbol interference
• the output of the Tomlinson precoder is a sequence of values which no longer take on the finite number of values -5, -3, -1, 1, 3 and 5, but, rather, a continuous set of values within a bounded range between -6 and +6. These values are applied to vestigial-sideband, or VSB, modulator 19.
• the VSB signal generated by modulator 19 is applied to a television channel, such as an over-the-air channel or a cable, and is received by the receiver of FIG 3.
• the received VSB signal is applied to VSB demodulator 39. Its output is processed by NTSC rejection filter 37, which has notches corresponding to those regions of the NTSC television signal spectrum where the signal energy is concentrated. This filter thus advantageously removes a major source of interference in the HDTV signal when the HDTV transmitter of FIG. 1 is in relatively close proximity to an NTSC transmitter broadcasting on the same television channel.
• the resulting signal is equalized by equalizer 36 to compensate for intersymbol interference introduced by the channel.
• the output of equalizer 36 is the equalizer's best approximation of the values of the sequence of interleaved signal points that was generated by signal point interleaver 15.
• the equalizer output is first deinterleaved in signal-point deinterleaver 35 and then applied to concatenated decoder 33 which includes Viterbi decoder 336; RS symbol deinterleaver 334, which re-orders the Reed Solomon symbols; and Reed-Solomon decoder 331, which provides a recovered version of the HDTV signal.
• the latter is decompressed and deformatted within television set 31 and displayed thereon.
• Some of the values that are output by equalizer 36 may be offset by a constant value of "12" or "-12" as a result of the action of Tomlinson precoder 17. Such an offset, however, is compensated for by the Viterbi decoder.
• synchronization signal points are periodically inserted by the transmitter into the signal point stream that it receives from signal-point interleaver 15.
• the receiver recognizes these synchronization signal points and, in response, generates a synchronization control signal which is used, in conventional fashion, by various components of the receiver (e.g., deinterleaver 334) in order to synchronize their operations with those of corresponding components in the transmitter (e.g, interleaver 134).
• FIGS. 4-7 A first coded modulation scheme that can be implemented by the combination of trellis encoder 136 and constellation mapper 139 is shown in FIGS. 4-7. This scheme encodes 8 bits per four-dimensional symbol of a predetermined four-dimensional constellation.
• FIG.4 shows the structure of trellis encoder 136.
• trellis encoder 136 receives the stream of serial bits supplied by RS symbol interleaver 134 and collects them into groups of eight parallel bits (e.g. by a serial- to-parallel converter not shown), denoted Y 1 administrat through Y 8 hail , where "n" denotes the current signaling interval.
• Bits Y 1 n and Y2 n are applied to an eight-state, rate 2 3 convolutional encoder 41 which provides three output bits.
• One of them, denoted Y0 n is generated by the delay elements and exclusive-OR circuits which comprise the convolutional encoder.
• Encoder 41 is a so-called "systematic" convolutional encoder, so that its other two output bits are simply its two input bits Y 1 n and Y2 n .
• the operations of convolutional encoder 41 can be explicitly described as follows.
• the encoder makes a transition from its current state Wl n W2 D W3 n to a next state W l n+4 W2 n+4 W3 n+ and outputs three bits Y2 n , Y l n , andYO n , where Wlford , W2 n , andW3 stii are the bits stored in the delay elements at the beginning of the symbol interval, and W 1 n+4 , W2 n+4 , andW3 n+4 are the bits stored in the delay elements at the end of the symbol interval, and
• W2 n+4 W l n ⁇ W2 n ⁇ Y1 D
• W3 n+4 W2 n ⁇ Y2 n
• Each of the eight different possible bit patterns represented by the three output bits of convolutional encoder 41 identifies a respect subset of the symbols of the four-dimensional constellation.
• the remaining six, so-called "uncoded,” bits Y3via through Y8 shall further select a particular symbol from the identified four- dimensional subset.
• the symbols of the four-dimensional constellation are partitioned into the aforementioned eight subsets based on a partitioning of its constituent one-dimensional base constellations.
• FIG. 7 shows how the one- dimensional six-point constellation is partitioned into two subsets, A and B, each subset having three one-dimensional signal points.
• the four-dimensional constellation is then partitioned into eight four-dimensional subsets, 0, 1, ..., and 7. As shown in FIG.
• each four-dimensional subset consists of two sequences of four one-dimensional subsets.
• four-dimensional subset 2 consists of the two one-dimensional-subset sequences (A.A33) and (B3-A.A), meaning that if a four-dimensional symbol is a member of subset 2, then either one of two criteria are met.
• one possibility is that its first constituent one- dimensional signal point is taken from subset A of the first constituent one- dimensional constellation; its second constituent one-dimensional signal point is taken from subset A of the second constituent one-dimensional constellation; its third constituent one-dimensional signal point is taken from subset B of the third constituent one-dimensional constellation; and its fourth constituent one-dimensional signal point is taken from subset B of the fourth constituent one-dimensional constellation.
• its first constituent one-dimensional signal point is taken from subset B of the first constituent one-dimensional constellation; its second constituent one-dimensional signal point is taken from subset B of the second constituent one-dimensional constellation; its third constituent one-dimensional signal point is taken from subset A of the third constituent one-dimensional constellation; and its fourth constituent one-dimensional signal point is taken from subset A of the fourth constituent one-dimensional constellation.
• the selection of a particular symbol from the identified four- dimensional subset proceeds as follows:
• the three bits, Y2 n , Y 1 detox, and Y0 n , from convolutional encoder 41, along with an uncoded input bit Y3 n , are first converted by a one-dimensional-subset-sequence selector 42 into another four bits, Z0 n , Z0 n+ 1 , Z0 n+2 , and Z0 n+3 .
• FIG. 5 shows the detail of this conversion.
• the effect of this operation is to select one of the two possible one-dimensional subset sequences of the 4-dimensional subset identified by bits Y0 cough through Y2 n ⁇ for example (A 33) or (B,B,A,A) of subset 2. Bits Y4 n through Y8 n are then used to select a symbol from the identified one-dimensional subset sequence.
• the five bits Y4 radical through Y8 n can represent only 32 different bit patterns, not all of the 81 symbols will actually be used. Rather, it is advantageous for the lookup table of FIG. 6 to map the 32 input bit patterns into the 32 smallest-energy symbols (the energy of a symbol being simply given by the sum of the squares of the coordinates of its constituent one-dimensional signal points.)
• bits Y4 n through Y8 n are applied to bit converter 45 which implements the lookup table shown in FIG. 6.
• FIGS. 8-10 A second coded modulation scheme that can be implemented by the combination of trellis encoder 136 and constellation mapper 139 is shown in FIGS. 8-10. This scheme encodes an average 8.5 bits per four-dimensional symbol.
• FIGS. 8-10 The modulation scheme of FIGS. 8-10 is built on the same underlying convolutional code as the first coded modulation scheme.
• encoder 81 and selector 82 shown in FIG. 8 are identical to encoder 41 and selector 42 shown in FIG. 4, so that, again, three input bits are used to identify a sequence of four one- dimensional subsets for each symbol interval. The remaining 5.5 uncoded bits per symbol are used to select a symbol from the identified subset sequence.
• this second coded modulation scheme uses the invention set forth in my United States Patent No.4,941,154 issued July 10, 1990 in order to realize a non-integral, or so- called "fractional" bit rate.
• the trellis encoder 136 collects from RS symbol interleaver 134 17 bits over two successive symbol intervals. Two groups of three, i.e., six, of those bits are used to identify the one-dimensional subset sequence for a respective two successive symbol interval. The remaining eleven uncoded bits are used to jointly select a pair of four-dimensional symbols from the two identified one-dimensional subset sequences.
• FIG. 8 Of the 11 uncoded bits, three are applied to a fractional bit encoder 83, implemented as the lookup table of FIG.9. As shown in FIG. 9, the output of encoder 83 is in the form of two pairs of bits, each of which can take on one of three possible bit patterns-00, 01 and 10. For each symbol interval one of the two pairs of encoder 83 output bits-denoted Y9 n and Y8 personally ⁇ is combined with four of the remaining eight uncoded bits to provide a six-bit input to bit converter 85. The latter, similar to the first scheme, is used to select a four-dimensional symbol from the first identified one-dimensional subset sequence, per the lookup table shown in FIG. 10.
• the second of the two pairs of encoder 83 output bits-which will be bits Y9 n+4 and Y8 n+4 ⁇ will be combined with the four remaining uncoded bits to provide a second six-bit input to bit converter 85, so that a second four-dimensional symbol will be selected from the second identified one-dimensional subset sequence.
• bits Y9 D and Y8 do not take on the value 11
• the total number of different bit patterns represented by bits Y4 n through Y9 suitcase is 48.
• the lookup table of FIG. 10 similar to the lookup table of FIG. 6, maps the 48 input bit patterns into the 48 lowest-energy symbols.
• any number of further coded modulation schemes can be implemented by, for example, varying the number of bits that are collected per symbol interval and accommodating the varying number of resulting uncoded bits via different mapping strategies following the basic concepts illustrated above.
• an average bit rate of 8.25 bits per symbol can be accommodated by performing the encoding over four four-dimensional symbol intervals-that is 33 bits per four four-dimensional symbol intervals-by mapping nine of the 21 uncoded bits into four three-bit groups which, for each symbol interval, are combined with three of the remaining twelve uncoded bits to address a bit converter similar to bit converter 85.
• FIG. 11 presents various properties and characteristics of the two coded modulation schemes explicitly described above, as well as a third, 9-bits-per-symbol interval scheme based on the same convolutional code and base constellation, that code being described in my co-pending United States Patent application, serial no. 08/226,606 filed April 12, 1994, hereby incorporated by reference.
• the signaling rate is the same, namely 10.76 x 10 6 signal points per second, or 10.76 Mbaud.
• the trellis code input bit rates of 8, 8.5 and 9 bits per symbol interval correspond to so-called "payloads"-i.e bit rates at the input to concatenated encoder 13-of 19.5, 20.7 and 21.9 Mbps, given the use of the above-mentioned RS(208,188) Reed-Solomon code.
• the signal points of one-dimensional base constellation at the output of constellation mapper 139 do not appear with equal probability.
• the signal points at coordinates -5, -3, -1, 1, 3 and 5 appear with probabilities 0.06, 0.17, 0.27, 0.27, 0.17 and 0.06, respectively.
• the probability of use decreases with increasing signal point energy, i.e., squared coordinate value.
• the average energy of the selected signal points is less than it would be if they were used with equal probabilities.
• the shaping gain of a given scheme at the output of constellation mapper 139 is a function of a ratio X Y -typically expressed in dB- where X is the average signal point energy of the base constellation that would result if the signal points appeared with equal probability, and Y is the average signal point energy at the output of constellation output 139 for the scheme under consideration.
• X is the average signal point energy of the base constellation that would result if the signal points appeared with equal probability
• Y is the average signal point energy at the output of constellation output 139 for the scheme under consideration.
• the shaping gains for schemes ⁇ and m are 1.67 dB and 0.67 dB, respectively. It is thus seen that one can achieve, via the use of different ones of these coded modulation schemes, increased levels of noise immunity in exchange for a decreased payload.
• the varying amounts of shaping gain are all achieved using a particular one, i.e., 6-signal- point one-dimensional base constellation which is, advantageously, the smallest constellation that can be used for any of the schemes I, II and m to support the input bit rates of the trellis encoder.
• the actual amount of shaping gain achieved by, for example, schemes I and II is really quite significant in terms of real life system performance.
• the Tomlinson precoder input signal is received on lead 151 from signal point interleaver 15. That signal is added by combiner 175 to a feedback signal provided on lead 174.
• the resulting signal is processed by modulo device 171.
• the function of modulo device is to ensure that the output of the Tomlinson precoder as a whole on lead 177 is maintained within a predetermined range of values ⁇ in this case in the range between -6 and +6.
• it does this, quite simply, by performing the conventional modulo -12 function. That is, it repetitively adds or subtracts the value "12" from the combiner output until the result falls within that range.
• the aforementioned feedback signal on lead 174 is a function of past Tomlinson precoder outputs on lead 177. It is, more particularly, provided by filter 173, which is illustratively a finite impulse response (FIR) filter, having a Z-transfer function given by
• Z "1 denotes a delay element with a delay amount of i signaling intervals.
• k 36
• h i through h 36 have values ranging between -0.0865 and 0.0603.
• Tomlinson precoder 17 is provided in the transmitter of FIG. 1 in order to compensate for the forced intersymbol interference (ISI) that is created by NTSC filter 37 in the receiver.
• ISI intersymbol interference
• the use of a Tomlinson precoder will eliminate the forced ISI, it is commonly held belief that the Tomlinson precoder will remove any "shaping gain" that might be provided by the preceding encoder. This belief is a consequence of the fact that in the prior art's understanding of the Tomlinson precoder operation, it was assumed that each allowed precoder output value will appear with the same probability as any other allowed value.
• an overall transmission scheme embodying the principles of the present invention comprises a channel encoder which provides a more-man-minimal amount of shaping gain in conjunction with a Tomlinson precoder.
• the invention contemplates the use of a channel code which provides such a high degree of shaping gain of its signal points that the shaping gain of the signal points at the output of the Tomlinson precoder is at least 1.0 dB.
• a shaping gain of 1.0 dB is significant and is achieved only via such a large tradeoff in bandwidth efficiency that absent the teachings of the present invention, the prior art-because of the aforementioned commonly held belief-is not led to use, in a system which includes a Tomlinson precoder, a channel code having such a large amount of shaping gain.
• the quality of the received signal could be measured at a remote sensing site and fed back via a wired or wireless telemetry channel to the broadcast site. If the quality deteriorates significantly, the transmitter can "fall back" to a lower-bit-rate, more- noise-immune scheme.
• a broadcaster may decide to augment a simple HDTV broadcast service offering to include a relatively-low-bit- rate news or other data (e.g., stock price) service.
• the additional payload required in order to deliver such an auxiliary service can be accommodated. Indeed, it will be appreciated from a consideration of these two examples that not all transmitters operating in a given geographic area need to be using the same coding scheme at the same time. Different coding schemes can be used simultaneously for different channels.
• control lead 12 is a lead extending to TV signal source 11 and concatenated encoder 13. There is provided on this lead a control signal specifying the desired payload rate, thereby allowing the signal source and encoder to adapt their operations to the desired rate.
• the HDTV or other receivers need to be able to decode signals transmitted using any of the different coding schemes.
• the shaping gain diversity is implemented via coding schemes from the same "family" as defined above, a relatively small amount of additional circuitry is required in order to be able to do that decoding.
• the same receiver "front end,” e.g., demodulator, equalizer, etc. can be used.
• the major portion of the Viterbi decoder- notably the portion which carries out the so-called maximum likelihood computations-will also be the same.
• the only change is in the circuitry which converts the recovered sequence of signal points back into bits. This, however, is a simple matter of using receiver lookup tables corresponding to the lookup tables used in the transmitter to implement the bit converters and fractional bit encoders, as described above.
• an ability to determine which scheme is being used can be achieved in various different ways.
• One way would be to monitor the distribution of sliced versions of the outputs of equalizer 36 and infer which coding scheme was used by the transmitter based on an a priori knowledge of the signal point distributions of the different coding schemes as shown in FIG. 11.
• Another way would be for the transmitter to explicitly encode this information into the HDTV signal itself.
• the various components of the transmitter and receiver are shown as individual functional blocks, the functions of any one or more of them could be provided by, for example, a single processor operating until software control; by one or more digital signal processing (DSP) chips; or by special-purpose integrated circuitry.
• DSP digital signal processing
• trellis encoder 136, constellation mapper 139, and signal point interleaver 15 could be replaced by a bank of trellis encoders and associated constellation mappers and interleavers.
• Each of the trellis encoders would receive a successive output of the interleaver in a round-robin fashion and each of the interleavers would provide a one-dimensional signal point in a round-robin fashion to Tomlinson precoder 17.
• the aforementioned bands of trellis encoders and associated constellation mappers need not be physically separate elements. Rather, the effect of such a bank could be achieved by time-sharing a single trellis encoder and constellation mapper in a manner that will be apparent to those skilled in the art
• the Tomlinson precoder will be removed from the HDTV transmitter of FIG. 1, and its associated NTSC rejection filter will be removed from the HDTV receiver of FIG.3.

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• 1994
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