WO1988008228A2 - Procede et appareil pour attenuer les bruits occasionnels dans les circuits integres - Google Patents

Procede et appareil pour attenuer les bruits occasionnels dans les circuits integres Download PDF

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Publication number
WO1988008228A2
WO1988008228A2 PCT/US1988/001051 US8801051W WO8808228A2 WO 1988008228 A2 WO1988008228 A2 WO 1988008228A2 US 8801051 W US8801051 W US 8801051W WO 8808228 A2 WO8808228 A2 WO 8808228A2
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WIPO (PCT)
Prior art keywords
current
circuit
transistor
voltage
package
Prior art date
Application number
PCT/US1988/001051
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English (en)
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WO1988008228A3 (fr
Inventor
Timothy Glen O'shaughnessy
David Kyong-Sik Chung
Richard William Hull
Kenneth W. Ouyang
Victor G. Pierotti
Joseph Arthur Souza
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Western Digital Corporation
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Priority claimed from US07/161,469 external-priority patent/US4947063A/en
Application filed by Western Digital Corporation filed Critical Western Digital Corporation
Publication of WO1988008228A2 publication Critical patent/WO1988008228A2/fr
Publication of WO1988008228A3 publication Critical patent/WO1988008228A3/fr

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/16—Modifications for eliminating interference voltages or currents
    • H03K17/161—Modifications for eliminating interference voltages or currents in field-effect transistor switches
    • H03K17/165—Modifications for eliminating interference voltages or currents in field-effect transistor switches by feedback from the output circuit to the control circuit
    • H03K17/166—Soft switching
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/16—Modifications for eliminating interference voltages or currents
    • H03K17/161—Modifications for eliminating interference voltages or currents in field-effect transistor switches
    • H03K17/162—Modifications for eliminating interference voltages or currents in field-effect transistor switches without feedback from the output circuit to the control circuit
    • H03K17/163—Soft switching
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K19/00—Logic circuits, i.e. having at least two inputs acting on one output; Inverting circuits
    • H03K19/003—Modifications for increasing the reliability for protection
    • H03K19/00346—Modifications for eliminating interference or parasitic voltages or currents
    • H03K19/00361—Modifications for eliminating interference or parasitic voltages or currents in field effect transistor circuits

Definitions

  • This invention relates to noise reduction in integrated circuits and, more particularly, to a method and apparatus for reducing the transient noise generated during operation of the output drivers of an integrated circuit.
  • Digital data processing systems are typically built from integrated circuits that incorporate on a single chip thousands of binary circuit elements such as registers, logic gates and input and output circuits.
  • an integrated circuit chip processes a number of parallel channels of data, e.g. eight, sixteen or thirty-two.
  • the integrated circuits are mounted on printed circuit boards having conductive paths that interconnect the input and output circuits on the integrated circuit chips and supply thereto operating power supply voltage.
  • An integrated circuit chip is conventionally incorporated into a package that has a number of leads connecting the integrated circuits on the chip to the printed circuit board. Since the output driver channels of an integrated circuit chip are connected through the package leads "off chip" to the conductive paths on the printed circuit board, they must be designed to drive a large capacitive load.
  • An integrated circuit chip has parasitic inductance, called package inductance, which is representative of the inductance of the conductive paths from the integrated circuit chip to the power supply potentials on the printed circuit board in which the integrated circuit chip is mounted.
  • package inductance parasitic inductance
  • the inter «nal conductive path in the package between the integrated circuit chip and the package lead comprises most of the package inductance.
  • the transient noise generated at the output drivers of an integrated circuit chip is reduced by maintaining an increasing ramp shaped current through each output driver during the entire transition interval between binary states of a capacitive load. As a result, during transfer of the charge to or from the capacitive load, the transient noise generated due to package inductance is minimized.
  • a capacitor fed by a fixed current source is connected across the input of each output driver stage.
  • the fixed current source and capacitor are so selected as to generate across the input of each output driver stage a linear ramp shaped control voltage that regulates the charging/discharging current through the output driver stage and package inductance in the described manner.
  • a feature of the invention is a specially designed bias circuit that reduces the sensitivity of the resulting transient noise to process variations and operating conditions. Preferably this is implemented by circuitry that forms a pair of -current mirrors in unity gain feedback.
  • Another feature of the invention is a feedback connection from the package inductance to the bias control circuit for the fixed current source to adjust the fixed current inversely with the transient noise.
  • Another feature of the invention is a dynamic clamp to suppress voltage spikes extending outside the voltage supply operating range.
  • Another feature of the invention is a bias circuit arrangement to compensate for sheet resistivity of the integrated circuit chip. If the resistance value of the sheet drops below a prescribed value, the fixed current is limited so it cannot exceed its designed value.
  • FIG. 1 is a schematic circuit diagram of a prior art integrated circuit output driver channel
  • FIGS. 2 to 4 are wave forms illustrating the operation of the prior art output driver of FIG. 1;
  • FIG. 5 is a schematic circuit diagram of an integrated circuit output driver channel incorporating principals of the invention.
  • FIGS. 6 to 8 are wave forms illustrating the operation of the output driver of FIG. 5;
  • FIG. 9 is a schematic circuit diagram of an output driver channel incorporating the preferred bias circuitry for the fixed current source and the dynamic clamp;
  • FIG. 10 is a schematic circuit diagram of an output driver channel applying the principals of the invention to reduce transient noise due to package inductance associated with two power supply connections;
  • FIG. 11 is a wave form illustrating the operation of part of FIG. 10;
  • FIG. 12 is a schematic circuit diagram of an output driver channel illustrating negative feedback to suppress transient noise
  • FIG. 13 is a schematic circuit diagram of another embodiment of an output driver channel illustrating negative feedback to suppress transient noise
  • FIG. 14 is a schematic, partially block diagram of the embodiment of FIG. 13 for a plurality of output driver channels;
  • FIG. 15 is a schematic circuit diagram of an alternative output driver stage incorporating principles of the invention
  • FIG. 16 is a schematic circuit diagram of an alternative predriver for an output driver channel incorporating the principals of the invention.
  • FIG. 17 is a schematic circuit diagram of another alternative predriver for an output driver channel incorporating principals of the invention.
  • the present invention is generally applicable to binary electronic circuitry for driving capacitive loads, particulary in the environment of integrated circuits.
  • Integrated circuit chips are usually incorporated into packages that have a number of leads connecting the integrated circuits to each other.
  • the integrated circuits are generally mounted on circuit boards that have conductive paths to form the interconnections between integrated circuits, and to bus power supply potentials to the integrated circuits.
  • An integrated circuit usually processes a number of parallel channels of data, e.g., eight, sixteen or thirty-two. It is particularly advantageous to employ the invention for multiple output driver stages of an integrated circuit. Since these output stages are connected through the package leads "off chip" to conductive paths on a printed circuit board, they must be designed to drive large capacitive loads.
  • common reference numerals are used to identify the same components of the various embodiments of the prior art and the invention.
  • a typical prior art integrated circuit output driver channel has as an output driver stage, an N-channel MOS transistor 58, and a complementary predriver stage having an N-channel MOS transistor 56 and a P-channel MOS transistor 104 all formed on the integrated circuit chip.
  • Power supply potentials V ⁇ D and V ss are bussed from "off chip", i.e. f . rom the printed circuit board on which the chip is mounted, through package leads to the circuitry "on chip".
  • V DD and V ss could be at any different potentials, it is assumed for discussion that V D D is connected to a positive potential bus and V ss is connected to a ground bus.
  • a package inductance 22 connected between S s an( the ground bus off chip represents the parasitic inductance of the conductive path from the chip to the printed circuit bus.
  • the internal conductive path in the integrated circuit package between the chip and the package lead comprises most of package inductance 22.
  • This package inductance generally does not adversely effect the operation of the output drivers as much as the parasitic ground package inductance because the electrical specification of the logic-0 voltage level during the high to low output transitions have less margin to transient noise than the logic-I voltage level during the low to high transitions.
  • An input terminal 60 is connected to the gates of transistors 56 and 104 ⁇
  • the source of transistor 104 is connected to V D £ ) , the drain of transistor 104, the drain of tran ⁇ sistor 56 and . the gate of transistor 58 are connected together.
  • the source of transistor 56 and the source of transistor 58 are connected to V ss .
  • the drain of transistor 58 is connected to a charging circuit 24 to V D r j and to an output terminal 23.
  • Output terminal 23 is connected to a data output package lead on the integrated circuit. When connected to a circuit path off chip, output terminal 23 in effect sees a large capacitive load 28 which comprises the package lead conductive paths on the printed circuit board and the input capacitance of the circuitry being driven.
  • a binary signal V IN representing a channel of data is applied to input terminal 60.
  • transistor 104 turns on and transistor 56 turns off.
  • V DD is applied to the gate of transistor 58, which has an internal gate to source capacitance represented as 21.
  • V DQ is applied to the gate of transistor 58, it turns on and load capacitance 28 discharges through transistor 58 and package inductance 22 to ground.
  • the output drivers of all the channels on the integrated chip are similarly connected to Vr j r j and V ss .
  • FIG. 2 represents the voltage, 0 TJT .
  • a output terminal 23 during the high to low transition of V IN t 0 represents the beginning of the transition interval of the change of state at output terminal 23 at which V 0 TJT starts to leave a binary "i" voltage level and t ⁇ represents the end of the transition interval of the change of state at output terminal 23 at which V TJT reaches a valid binary "0" voltage level.
  • the voltage at output terminal 23 drops approximately linearly until the transition is completed at a time t ] _.
  • the period between t ⁇ and t 0 which varies from chip to chip depending on manufacturing process variations, is referred to hereafter as the transition interval.
  • the high threshold at output terminal 23 for low to high transitions could be about 2.0 volts
  • the low threshold at output terminal 23 for high to low transitions could be about 0.8 volts
  • the typical transition interval could be between about 10 and 40 nanoseconds (ns) .
  • ns nanoseconds
  • FIG. 3 represents the discharge current, I G , passing through transistor 58 and package inductance 22 during a high to low transition. As illustrated, this current rises sharply when transistor 58 turns on at t 0 and stays approximately constant at a short circuit sink current value, l s , during the remainder of the transition interval.
  • V D Q When the voltage V IN applied to input terminal 60 goes from low to high, V D Q is applied through a charging circuit 24 to load capacitance 28 to charge load capacitance 28.
  • Charging circuit 24 could comprise a simple charging resistor or a switch comprising a transistor that turns on during the charging interval and turns off outside the charging interval.
  • the broader aspects of the invention are illustrated by an output driver 20 in FIG. 5.
  • a capacitor 38 is connected between the gate of transistor 58 and V ss .
  • Capacitor 38 represents the combined capacitance of discrete capacitor and/or the usually smaller internal gate-source capacitance of transistor 58.
  • a source 34 of fixed bias current I F is connected between V DD and the drain of transistor 104.
  • the bias current I F is fixed in the sense that it is constant throughout the transition interval irrespective of changes in the input impedance of transistor 58.
  • the fixed current I F is applied by transistor 104 to capacitor 38.
  • an approximately linear ramp-shaped control voltage is generated across capacitor 38 during the entire transition interval.
  • the transistor in current source 34 eventually saturates and the current produced thereby " drops.
  • transistor 58 serves not only as a switch but also as a transconductance device to control the rate of the current change passing through transistor 58 and package inductance 22.
  • the discharge current I G has an approximately linear ramp-shape over the entire transition interval, i.e.
  • discharge current I G at the end of the transition interval t l r which is designated I M in FIG. 7 is equal to twice the minimum value of the current I s in FIG. 3 necessary to discharge the load capacitance from high to low during the transition interval, t ⁇ - t 0 .
  • this produces a voltage V 0T JT across output terminal 23 that changes slowly during the first half of the transition interval and changes rapidly during the second half of the transition interval.
  • the invention establishes the same required output voltage at t]_ as the prior art. Yet the discharge current, I G , passing through package inductance 22 is increased in ramp shaped fashion., preferably at a constant rate, over the entire transition interval so as to minimize the rate of current change through package inductance 22, and thus the transient noise voltage induced thereby.
  • FIG. 9 illustrates a more detailed embodiment of the output driver of the present invention in which charging circuit 24 in FIG. 5 takes the form of a P-channel MOS transistor 54.
  • the input voltage signal V IN - at the input terminal 60 of the output driver is high.
  • the P-channel transistor 54 located between the power supply voltage V DQ and the load capacitance 28 is therefore conductive, allowing the power supply voltage V DD to charge the load capacitance 28.
  • the N-channel transistor 56 is also conductive.
  • the opposed terminals of the capacitor 38 are therefore shorted and the capacitor 38 remains uncharged.
  • the high input voltage signal V IN is applied to the gate of the N-channel transistor 56, making transistor 56 electrically conductive, and transistor 58 nonconductive.
  • the package inductance 22 is thus electrically isolated from the load capacitance 28.
  • the output driver becomes activated to discharge load capacitance 28 when the input voltage signal V IN changes from high to low.
  • the N-channel transistor 56 becomes nonconductive, allowing the fixed, i.e. constant, current from the current source 34 to begin charging the capacitor 38. A linearly increasing voltage is therefore applied to the gate of the N-channel transistor 58.
  • P-channel transistor 54 turns off, electrically isolating the load capacitance 28 from the power supply voltage V DD . The charge stored on the load capacitance 28 therefore commences to discharge through transistor 58.
  • transistor 58 behaves as a transconductance element to control its output current to assume the same function over time as the voltage applied to its gate, i.e. to follow the gate vpltage, and the voltage applied , to the gate of the transistor 58 is linearly increasing in form, the N-channel transistor 58 begins to conduct a linearly increasing level of current I to the package inductance 22.
  • the rate of change of current I G passing through the package inductance 22 is therefore constant for a period of time, minimizing the level of switching noise generated during activation of the output driver.
  • an inverter 62 and a capacitor 64 may be provided between input terminal 60 and the gate of the transconductance transistor 58.
  • the capacitor 64 functions to transfer a charge to bias the gate of the transistor 58 to have a gate voltage just below its threshold immediately prior to activation of the output driver by the input voltage V IN . Only a minimal level of additional voltage therefore must be applied to the gate of the transistor 58 to make it conductive, speeding up the response time of the output driver 20 upon the change of state of the input voltage
  • a second refinement may be made to the output driver by replacing the fixed bias current source 34 illustrated in Figure 5 with a modified, compensating bias current source, as illustrated in Figure 9.
  • a MOS transistor 58 as the transconductance element, its transconductance decreases as temperature increases, lowering the slope of the ramp current passing through transistor 58 and increasing the delay in switching.
  • the current source illustrated in Figure 9 overcomes this problem and compensates for process variations in N-channel transconductance by providing a first P-channel MOS transistor 66, a second P-channel MOS transistor 68 having its gate and drain shorted in a diode configuration, a third P-channel MOS transistor 69, a first N-channel MOS transistor 70 having its gate and drain shorted in a diode configuration, a second N-channel MOS transistor 72 having its gate connected to the gate of transistor 70, and a degenerative resistor 74 placed between the sources of transistors 70 and 72.
  • the drain of the first P-channel MOS transistor 66 is connected to the drain of the first N-channel MOS transistor 70, and the drain of the second P-channel MOS transistor 68 is connected to the drain of the second N-channel MOS transistor 72.
  • the gates of the first P-channel MOS transistor 66 , the second P-channel MOS transistor 68 are also connected, as are the gates of the first N-channel MOS transistor 70 and the second N-channel MOS transistor 72.
  • the third P- channel MOS transistor 69 has a gate connected to the gate and drain of the second P-channel MOS transistor 68, a source connected to DQ and a drain connected to the gate of transistor 58.
  • the sources of the first, second and third P-channel MOS transistors are connected to the power supply voltage.
  • the degenerative resistor 74 consists of a heavily doped semiconductor material which exhibits a very low temperature coefficient. Although this circuit is in itself known, its use to generate the bias current in the practice of the invention is an important aspect of the invention, because it reduces the sensitivity to process variables, i.e. N-channel transconductance, and to operating conditions, i.e. power supply and temperature variations. Because transistors 70 and 72 are formed on the same chip as transistor 58, their variations in parameters due to processing are substantially the same. Transistors 66 and 68 are balanced current mirrors in that they have the same channel width to length ratio ( W /L) . .
  • Transistors 70 and 72 are imbalanced current mirrors in that the channel width to length ratio of the latter is larger than that of the former, e.g. by a multiple of four.
  • the unity gain feedback provided by this circuit constrains the current through transistors 66 and 70 to be the same as the current through transistors 68 and 72.
  • Transistor 69 is also designed to function as a current mirror with transistor 68, i.e. transistor 69 may have the same channel width to length ratio as transistors 66 and 68. As a result, the same current passing through transistors 68 and 72 also passed through transistor 69 and charges capacitor 38 when the output driver is activated.
  • the bias current supplied by transistor 69 to capacitor 38 is inversely related to the resistance of resistor 74, the transconductance of N-channel transistor 70 and 72, and the channel width to ' length ratio of N-channel transistors 70 and 72.
  • the bias current supplied by transistor 69 is relatively small, and visa versa, with the result that the slope of the discharge current I G remains approximately constant over the prescribed range of N-channel transconductance of the chip.
  • the bias circuit senses variations in N-channel transconductance of the chip and changes the bias current to compensate for such variations.
  • Selection of the described bias circuit also desensitizes the slope of the discharge circuit I to variations in power supply voltage V Dj .
  • the transconductance of imbalanced transistors 70 and 72 also tracks changes in the transconductance of the transistor 58 over temperature.
  • the imbalanced transistors produce an imbalanced voltage across the degenerative resistor 74.
  • the transconductance of the imbalanced transistors 70 and 72 decrease? with increasing temperature, causing an increasingly imBalanced voltage across the degenerative resistor 74 as temperature increases. As the voltage across the degenerative .
  • resistor 74 increases at higher temperatures, more current is supplied to the capacitor 38 to increase the slope of the ramp voltage per unit time applied to the gate of transistor 58, .thereby increasing the slope of the ramp of the current at terminal 23 and decreasing the delay caused by the temperature increase.
  • a positive temperature coefficient is created in the current used to charge the capacitance 38.
  • the capacitance 38 thus charges at an increasing rate as temperature increases, biasing the transistor 58 to conduct a ramp current which is stabilized over a large temperature range.
  • degenerative resistor 74 and transistors 66, 68, 70 and 72 compensate for the effect of temperature changes upon the transconductance of transistor 58 so that the ramp current passing through transistor 58 during discharge of load capacitance 28 maintains an approximately constant slope over a large temperature range.
  • a faster or slower switching speed may be attained for the same value of package inductance and/or load capacitance. For example, if a fast switching speed is required, a large W/L may be chosen, increasing the slope to the ramp illustrated in Figure 6, thereby increasing the speed of switching. Alternately, a small W/L may be chosen, decreasing the slope of the ramp illustrated in Figure 6, and thereby correspondingly decreasing the speed of switching.
  • the same switching speed may be attained for either larger or smaller values of load capacitance being driven by respectively increasing or decreasing the value of W/L for the third P-channel MOS transistor 69. Therefore, the geometric design of the output driver need not change for different applications. If in the specific application a smaller package inductance occurs, but the switching delay must be reduced, then the design only requires increasing the W/L of the third P-channel MOS transistor 69.
  • a third refinement of the present invention provides clamping circuitry for further maintaining a low level of noise during activation of the output driver.
  • the output voltage V QUT illustrated in Figure 5 of the output driver 20 decreases during discharge of the load capacitance 28, the transconductance of the transistor 58 decreases, regardless of the particular device utilized as the transconductance device. If the transconductance decreases, the discharge current level I G decreases dramatically.
  • the rate of change of the current which is the derivative of the current, may form a negative spike 82, as illustrated in Figure 6.
  • the negative spike 82 corresponds to a low voltage signal and is therefore a valid binary logic zero.
  • the negative spike 82 may forward bias the isolation diffusion or substrate of the inputs of the other electronic devices being driven by the output driver, leading to latch-up and/or improper operation of the other electronic devices being driven.
  • the clamp 83 illustrated in Figure 9 is designed to change the net current through the package inductance 22 when the rate of change of the current I G changes, thereby suppressing, i.e. reducing the magnitude of the voltage spike across package inductance 22.
  • the clamp 83 is composed of a current source having a P-channel transistor 84, and an N-channel transistor 90 with its gate and drain connected in a diode configuration and an N-channel MOS transistor 92.
  • a first P-channel transistor 86 with gate and drain connected in a diode con iguration, a second P-channel transistor 88, a resistor 94 and a capacitor 96 are also provided.
  • the current source, the transistor 86, the resistor 94 and the capacitor 96 cooperate to generate a precision gate voltage for the transistor 88 and a steady state charge across the capacitor 96.
  • capacitor 96 charges.
  • the resulting current temporarily increases the source to gate voltage of the transistor 88 and the current I 3 passing through transistor 88 and package inductance 22 to ground.
  • the increase in the current I 3 offsets at least in part decrease in the current I G and therefore suppresses the negative voltage spike 82.
  • clamp 83 thus suppresses negative voltage spike 82 and thereby reduces the likelihood of latch-up of the other electronic devices being driven by the output driver.
  • clamp 83 also serves to partially offset increases in the current I G . Specifically, when the rate of change of current I G starts to go positive, capacitor 96 discharges and the source to gate voltage of transistor 88 temporarily decreases. As a result, the current I 3 decreases to offset the increase" in the current I G . Thus, clamp 83 suppresses positive voltage spikes as well, until such time as transistor 88 turns off.
  • the transistor 88 functions as part of a negative feedback loop that supplies to package inductance 22 a bias current I 3 .
  • a bias current I 3 When the current I G changes, a voltage is generated across the package inductance 22 which changes the charge on and voltage across capacitor 96. This in turn changes the bias current I 3 .
  • the bias current I 3 thus varies according to the rate of change of the discharge current I G so as to suppress voltage spikes that would otherwise be induced across package inductance 22.
  • a P-channel transistor 104 has been connected to the N-channel transistor 56 to form a full CMOS ratioed complementary inverter pair 100.
  • the source of the transistor 104 is connected to the drain of transistor 69, and the input voltage V N of the output driver applied to the gates of both transistors 56 and 104.
  • the drains of the transistors 56 and 104 are connected to one another and to the gate of transistor 58.
  • the input voltage V jN Prior to activation of the output driver of Figure 10, the input voltage V jN is high.
  • the N-channel transistor 56 is therefore conductive, the opposing terminals of the capacitor 38 are shorted, and the .gate of the transistor 58 is low, as desired.
  • the P-channel transistor 104 is nonconductive , however. As a result, the current path through the transistor 56 is eliminated. Power consumption is thereby reduced during the time period prior to activation of the output driver.
  • a charge Q may develop at a node 106 when the transistor 104 is nonconductive.
  • the charge Q transfers towards the gate of transistor 58.
  • a nonlinear increase in voltage may therefore occur at the gate of the transistor 58. Since it is an object of the present invention to provide a linearly increasing voltage V G to transistor 58 such that it may operate as a transconductance element to output a linearly increasing output current to the inductance 22, it is desirable to minimize the effect of the momentary increase in gate voltage V G on the transistor 58.
  • the gate voltage V G of transistor 58 may increase in one of a number of manners as a result of the transfer of the charge Q.
  • the transistor 104 becomes conductive very quickly and the transistor 56 becomes nonconductive very quickly when the input voltage V IN goes low.
  • the gate voltage V G of the transistor 58 rises as a step function to the point A as much of the charge Q is transferred to the gate of the transistor 58, as illustrated by portion 11A of curve 11. The maximum value of.
  • the voltage attained at the point A is determined by the relative ratio of the parasitic capacitance C 106 of the node 1° 6 to tne sum °f tne parasitic capacitance C5 8 of the gate of transistor 58 and the integrated capacitance C 3 s of capacitor 38.
  • the gate voltage V G of transistor 58 transforms into a linearly .increasing function represented by the portion of the curve 11B as the capacitor 38 becomes charged by the current from the current source.
  • transistor 58 Since transistor 58 operates as a transconductance element only when its gate voltage is higher than its threshold voltage V T , the effect of the step-like increase in the gate voltage V G will be minimal if the gate voltage V G at point A does not rise above the level of the threshold voltage V ⁇ of the transistor 58.
  • Maintaining the gate voltage V G of transistor 58 below the threshold voltage V ⁇ as the inverter is switched may be accomplished by selecting an appropriate value for the ratio C 106 /( c 38 + c 5 ⁇ ) ' which, as previously stated, controls the highest voltage level at point A of the step function. Moreover, the delay due to the time required to make transistor 58 conductive may also be minimized by selecting an appropriate value of this ratio since a maximum gate voltage at point A may be selected which is only slightly below the threshold voltage V ⁇ of transistor 58. When current from the current source then begins to charge the capacitor 38, the transistor 58 will almost instantaneously commence to operate as a transconductive element.
  • the output driver of Figure 10 may also incorporate the clamp 83 illustrated in Figure 9.
  • the output driver of Figure 10 also provides additional refinements to the current source of Figure 9 which may also be used in the output driver of Figure 9.
  • the current source illustrated in Figure 9 utilizes a degenerative resistor 74 consisting of a heavily doped semiconductor material which exhibits a very low temperature coefficient.
  • the resistance value of the degenerative resistor 74 may vary significantly with manufacturing process variations. If the resistance value of the degenerative resistor 74 falls too far below its designed value, an excess voltage may be developed across the degenerative resistor 74, resulting in the supply of excess bias current to the transistor 58, fast switching of the output driver and the possible generation of excessive switching noise.
  • the modifications shown in Figure 10 provide a means for accommodating these process variations by sensing the actual voltage across the degenerative resistor 74. If the actual voltage level is significantly higher than the designed voltage value, the accommodating means tends to reduce the bias current supplied to the capacitor 38 to the originally designed bias current value.
  • the modified current source illustrated in Figure 10 includes a P-channel MOS transistor 108 having its gate connected to the gates of transistors 66 and 68 and its source connected to the power supply.
  • a resistance element 110 is provided having one terminal grounded and its other terminal connected to the drain of transistor 108.
  • a second P-channel MOS transistor 112 is provided having its source connected to the power supply and its drain connected to both its gate and the drain of an N-channel MOS transistor 114.
  • the gate of transistor 114 is connected to the drain of the transistor 108 and its source is connected to a grounded resistor 116.
  • a third P-channel MOS transistor 118 has its source connected to the power supply and its gate connected to the gate of the transistor 112 in a current mirror configuration.
  • a fourth P-channel MOS transistor 1.20 has its gate and drain connected in a diode configuration.
  • the drains of transistor 118 and 120 are connected to the gate of transistor 69.
  • a second N-channel MOS transistor 122 is also provided, with its source grounded, its drain connected to the drain of transistors 118 and 120, and its gate connected to the gates of transistors 70 and 72.
  • the current level I ⁇ lAS passing through the transistor 122 corresponds to the current I ⁇ iAS passing through the transistor 69 in Figure 9. Assuming that the value of the resistance of the degenerative resistor 74 may be lower than a specified value due to process variations, the actual value of the current I ⁇ lAS passing through the transistor 122 may have a level which is much higher than the desired.
  • the modified current source of Figure 10 tends to reduce the current I ⁇ lA S -°Y an amount I ⁇ to produce a new current level IJJIAS' formed by the P-channel transistors 69 and 120 of Figure 10 which is supplied to the current mirror structure of transistor 69 of Figure 10.
  • the amount of the current reduction I ⁇ is determined in the following manner.
  • the current passing through the degenerative resistor 74 is duplicated by the transistor 108 by the current mirror configuration formed with the transistor 68.
  • the duplicated current passes through the resistor 110 to develop an amplified voltage across the resistor 110 which is equal to the voltage produced across the degenerative resistor 74 multiplied by the ratio of the resistance of resistor 110 over the resistance of the resistor 74.
  • the amplified voltage across the resistor 110 is applied to the gate of the transistor 114.
  • the ratio of the channel width over the channel length (W/L) of the transistor 114, in conjunction with the resistance of the resistor 116, operate to develop a current I x of a selected value which is passed through transistor 112 in response to the amplified voltage across the resistor 110.
  • the current mirror transistors 112 and 118 subtract the selected current level I ⁇ from the current level I ⁇ i S passing through the transistor 122, resulting in I'BIAS passing through device 120.
  • the transistors 118, 112 and 122 thus form a current mirror subtractor with transistor 120 such that a new current- level I'siA S ⁇ S developed which has a value equal to the value of the current I ⁇ iAS m i nus tne value of the current I x . Since the P-channel transistors 69 and 120 are in a current mirror configuration, the transistor 69 produces a duplicate of the current I'BIAS which passes through the diode connected transistor 120.
  • charging circuit 24 takes the form of a full complement of the described discharging circuit in order to reduce transient noise voltage on the low to high voltage transitions due to a package inductance 22C.
  • Inductance 22C represents the parasitic inductance of the conductive path, including package lead, from the integrated circuit chip on which the output driver is formed to the system positive power supply voltage bus on the printed circuit board. Additional circuitry may also be provided for generating the complement of the bias current I'BIAS ° r ⁇ ° the complementary output driver.
  • the additional circuitry may include, for example, a P-channel MOS transistor 124 having its gate connected to the gate of transistor 120 and its source connected to the power supply.
  • An N-channel MOS transistor 126 is provided having a grounded source and its gate connected to both its drain and the drain of transistor 124.
  • a complementary bias line 134 may also be provided through which the complementary bias current I'BIAS C ⁇ na pass to the complementary transistor 69C.
  • Capacitors 128 and 130 are optionally provided to make bias line 132 and complementary bias line 134 immune to minor levels of noise.
  • Fig. 12 is a simplified version of Fig. 10.
  • the clamping circuitry is not shown. Neither is charging circuit 24.
  • the bias source is represented by a P-channel MOS transistor 150 and a current source 152. Power supply voltage V DD is applied to the source of transistor 150.
  • Current source 152 is connected from the gate and drain of transistor 150 to V ss .
  • the gate of transistor 150 is connected to the gate of transistor 69.
  • the gate and drain of transistor 150 are connected in a diode configuration.
  • An important refinement of this embodiment is a negative feedback connection from V ss through a capacitor 154 to a bias node 156 at the junction of the drain and gate of transistor 150 and the gate of transistor 69.
  • Transistor 150 and current source 152 provide a voltage bias at node 156.
  • this voltage bias is changed by the negative feedback applied to node 156 through capacitor 154.
  • Transient noise voltage induced across package inductance 22 by the current discharged from load capacitance 28 during a high to low input voltage transition gives rise to a current that is fed back through capacitor 154 which in turn reduces the net bias current of transistor 150.
  • the connection through capacitor 154 provides negative feedback that suppresses the transient noise voltage across package inductance 22.
  • a common i.e.
  • Capacitor 154 serves to block direct current coupling between Vss and node 156, and to provide a predetermined displacement current to the bias node.
  • a bias coupling resistor 158 individual to the illustrated output driver channel is connected in series with a capacitor 154 individual to the illustrated output driver channel.
  • Bias node 156 which is formed at the junction of capacitor 154 and resistor 158, is connected to the gate of transistor 69.
  • P-channel MOS transistors 160, 162, 164 and 166, and a current source 168 serve as a common voltage bias source for all the local driver channels.
  • Power supply voltage V D D is applied to the sources of transistors 160 and 164.
  • the drains of transistors 160 and 164 are connected to the drains of transistors 162 and 166, respectively.
  • Current source 168 is connected from the drain of transistor 166 to V S s-
  • the drain of transistor 162 is connected to V S s- Tne gates of transistors 160 and 164 are connected together; the gates of transistors 162 and 166 are connected together; the drain* and gate of transistor 166 are connected together; and the drain and gate of transistor 164 are connected together in a diode configuration.
  • a capacitor 170 connected between power supply V DD and the gates of transistors 160 and 164 serves to reduce the effect of switching noise in disturbing the bias current of 160 and 162.
  • the circuitry within a broken line box 20 comprises one of a number of output driver channels in the integrated circuit chip connected to the same package ground lead.
  • the circuitry within a broken line box 174 serves as the common voltage bias source in the integrated circuit chip connected to the same package power supply and ground leads.
  • a plurality of bias coupling resistors such as that designated 158 connect the common bias source 174 to the respective output driver channels. This is depicted in Fig. 14.
  • a common bias source 174 which is the same as bias source 174 in Fig. 13, is connected by resistors 158a, 158b, . . . 158n to output driver channels 20a, 20b, . . . 2On, respectively, which are the same as driver channel 20 in Fig. 13.
  • the binary signals to be carried “off chip” are applied to input terminals 60a, 60b, . . . 60n of driver channels 20a, 20b, . . . 2On, and output terminals 23a, 23b, . . . 23n are connected to the output data package leads of the integrated circuit.
  • Power supply voltage V DD is applied to common bias source 174 and each of output driver channels 20a, 20b, . . . 20n.
  • the local ground on the chip V S s is connected to common bias source 174 and each of output driver channels 20a, 20b, . . . 2On.
  • the connection of V ss to the system ground (GND) through the ground package lead of the integrated circuit gives rise to parasitic package inductance 22.
  • Common bias source 174 establishes at bias nodes 156a, 156b, . . . 156n for output driver channels 20a, 20b, . . . * 2On, respectively, bias voltages that establish the level of the constant charging current passing through transistors 69 and 104 when the driver circuit is actuated.
  • Feedback capacitors 154a, 154b, . . . 154n are connected from package inductance 22 to nodes 156a, 156b, . . . 156n, respectively. Because of the low impedance at the junction of transistors 160 and 162, resistance must be provided between such junction and node 156 in order to generate the gate voltage for transistor 69 due to the feedback current.
  • the size of the capacitance and resistance required to feedback the gate voltage for transistor 69 shall depend -on the number of output drivers, because the parasitic capacitance from V D £> to node 156 increases as the number of drivers increases.
  • This time constant is somewhat critical. An excessive time constant produces a prolonged disturbance in the bias voltage, possibly causing excessive delay in the output drivers. An insufficient time constant may produce inadequate displacement current to produce the required momentary shift in bias voltage.
  • the time constant of the feedback connection in essence occurs within the various output driver channels, and no longer shall R 158 and C 154 depend on the number of output drivers used.
  • the feedback factor is selected so the transition interval of the faster and slower transistors is about the same for worst case noise conditions, e.g. about 40 ns. This gives the most noise suppression without requiring a longer transition interval.
  • an output driver stage could comprise an operational amplifier formed by a difference amplifier 180, a bipolar transistor 182, an emitter follower resistor 184, and a feedback connection 186.
  • the junction of the drain of transistor 56 and source of transistor 104 comprises the input to the driver stage, which is connected to the positive input of amplifier 180.
  • the output of amplifier 180 is connected to the base of transistor 182.
  • the collector of transistor 182 is connected to output terminal 23.
  • the emitter of transistor 182 is coupled by feedback connection 186 to the negative input of amplifier 180 and by resistor 184 to V ss .
  • the described operational amplifier performs the same switching and transconductance functions of MOS transistor 58 in the previously described embodiments.
  • the bias circuit in this case would comprise a fixed current source over temperature since the transconductance is inversely proportional to the resistance of resistor 184.
  • an MOS transistor could, be employed in the operational amplifier instead of bipolar transistor 182, or the output driver stage could comprise an emitter follower bipolar transistor stage or a grounded emitter bipolar transistor stage.
  • inverters 188 and 190 are connected between input terminal 60 and the gate of transistor 104 and the gate of transistor 56, respectively.
  • the W/L ratio of inverter 188 is designed to be sufficiently small relative to that of inverter 190 to insure that transistor 56 turns off before transistor 104 turns on.
  • another inverter 192 e.g.
  • N-channel MOS transistor connects input terminal 60 to the inputs of inverters 188 and 190.
  • a parasitic capacitance is formed at the node 106.
  • the charge stored at node 106 is redistributed vis-a-vis capacitor 38, which gives rise to a change in current through the package inductance 22C.
  • This dl/dt results from the saturated current source 69 becoming active when the voltage at node 106 decreases as the charge of node 106 is redistributed.
  • P-channel transistor 194 and an N-channel transistor 196 comprise a CMOS logic element, i.e. an inverter, that feeds the output driver 20. Instead of an inverter, other CMOS logic elements such as a NOR circuit or a NAND circuit could be employed.
  • An additional N-channel transistor 198 provides a bias current to control the discharge of the capacitance of node 106. V D r j is applied to the source of transistor 194.
  • the drain of transistor 194 and the drain of transistor 196 are connected to the input terminal 60.
  • the source of transistor 196 is connected to the drain of transistor 198.
  • V ss is applied to the source of transistor 198.
  • Bias line 134 (see FIG. 10) is con ⁇ nected to the gate of transistor 198.
  • a data line 200 at one of two binary voltage levels is connected to the gates of transistors 194 and 196. When line 200 transitions from low to high, the voltage at input terminal 60, V N , transitions from high to low thereby discharging capacitance 28 in the manner described above.
  • transistor 198 controls the discharge current from node 106 through transistor 104 to be an increasing ramp-like current so as to provide an increase in the gate voltage on transistor 58 to point A (see FIG'. 11) in a curved path rather than as a step function.
  • transistor 198 would slow the rise time of the current discharge from node 106 from less than 1 ns to approximately 3 ns.
  • transistor 198 serves to control the discharge current from node 106 in a fashion analogous to the control of the discharge current from output capacitance 28 by transistor 58, although only during a small portion of the transition interval.
  • the voltage coupled to the gate of transistor 198 by bias line 134 serves to compensate for changes in P-channel transconductance analogous to the N-channel compensation provided to the gate of transistor 69 by bias line 132.
  • the output driver is designed so the controlling effect of transistor 198 occurs prior to the time that transistor 58 reaches its threshold V ⁇ so it does not influence the slope of the discharge current flowing through package inductance 22.

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  • Engineering & Computer Science (AREA)
  • Computer Hardware Design (AREA)
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  • General Engineering & Computer Science (AREA)
  • Mathematical Physics (AREA)
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Abstract

Le bruit occasionnel qui se produit au niveau des circuits d'attaque de sortie d'une puce est attenué lorsqu'on fait passer en permanence un courant d'intensité progressive dans chacun de ces circuits pendant tout le temps qui s'écoule entre des états binaires d'une charge capacitative. Un condensateur alimenté par une source de courant fixe est raccordé au niveau de l'entrée de chaque circuit d'attaque de sortie. La source de courant fixe et le condensateur sont choisis de manière à produire au niveau de l'entrée de chaque circuit d'attaque de sortie une tension de commande à profil de montée linéaire, qui régule le courant de charge/décharge passant dans le circuit d'attaque de sortie et l'inductance du module de la manière décrite. Un circuit de polarisation de conception particulière réduit la sensibilité du bruit occasionnel qui en résulte aux variations de traitement et aux conditions de fonctionnement. Un système de réaction reliant l'inductance du module au circuit de commande de polarisation de la source de courant fixe règle le courant fixe en fonction inverse du bruit occasionnel. Un dispositif de verrouillage dynamique supprime les pointes de tension en dehors de la plage de fonctionnement de l'alimentation en tension. Un dispositif du circuit de polarisation compense la résistivité de couche de la puce. Si la résistance de la couche tombe au-dessous d'une valeur déterminée, le courant fixe est limité afin de ne pas être supérieur à la valeur prévue à l'origine.
PCT/US1988/001051 1987-04-07 1988-04-01 Procede et appareil pour attenuer les bruits occasionnels dans les circuits integres WO1988008228A2 (fr)

Applications Claiming Priority (8)

Application Number Priority Date Filing Date Title
US3560287A 1987-04-07 1987-04-07
US035,602 1987-04-07
US6746787A 1987-06-26 1987-06-26
US067,467 1987-06-26
US10749687A 1987-10-09 1987-10-09
US107,496 1987-10-09
US161,469 1988-02-26
US07/161,469 US4947063A (en) 1987-10-09 1988-02-26 Method and apparatus for reducing transient noise in integrated circuits

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WO1988008228A2 true WO1988008228A2 (fr) 1988-10-20
WO1988008228A3 WO1988008228A3 (fr) 1989-01-26

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EP0493185A1 (fr) * 1990-12-27 1992-07-01 Automobiles Peugeot Circuit de commande d'un transistor de puissance utilisé en commutation forcée
EP0492506A2 (fr) * 1990-12-28 1992-07-01 STMicroelectronics S.r.l. Circuit de commande rapide à charge capacitive destiné à des circuits intégrés en particulier de type mémoire
EP0610064A2 (fr) * 1993-02-01 1994-08-10 STMicroelectronics Limited Commutation à transistor
EP0653841A1 (fr) * 1993-11-15 1995-05-17 International Business Machines Corporation Commutateur à haute puissance comportant commande de courant de démarrage
WO1999035743A1 (fr) * 1997-12-30 1999-07-15 Siemens Aktiengesellschaft Procede et dispositif de coupure en cas de court-circuit et de surcharge au moyen d'un composant a semiconducteur
US6424187B1 (en) 1996-12-18 2002-07-23 Hitachi, Ltd. Driving apparatus for power MOS transistor
FR2878065A1 (fr) * 2004-11-18 2006-05-19 St Microelectronics Sa Circuit de decharge d'une charge electrique, et etage de sortie de puissance comprenant un tel circuit de decharge pour la commande de cellules d'ecran plasma

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GB2140996A (en) * 1983-05-31 1984-12-05 Gen Electric Controlled switching of non-regenerative power semiconductors
EP0164615A2 (fr) * 1984-06-13 1985-12-18 International Business Machines Corporation Circuit de commande pour le contrôle de la montée et de la décroissance d'un signal dans des circuits à transistor à effet de champ

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GB2140996A (en) * 1983-05-31 1984-12-05 Gen Electric Controlled switching of non-regenerative power semiconductors
EP0164615A2 (fr) * 1984-06-13 1985-12-18 International Business Machines Corporation Circuit de commande pour le contrôle de la montée et de la décroissance d'un signal dans des circuits à transistor à effet de champ

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Cited By (16)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0493185A1 (fr) * 1990-12-27 1992-07-01 Automobiles Peugeot Circuit de commande d'un transistor de puissance utilisé en commutation forcée
FR2671241A1 (fr) * 1990-12-27 1992-07-03 Peugeot Circuit de commande d'un transistor de puissance utilise en commutation forcee.
EP0492506A2 (fr) * 1990-12-28 1992-07-01 STMicroelectronics S.r.l. Circuit de commande rapide à charge capacitive destiné à des circuits intégrés en particulier de type mémoire
EP0492506A3 (en) * 1990-12-28 1992-10-28 Sgs-Thomson Microelectronics S.R.L. Fast capacitive-load driving circuit for integrated circuits particularly memories
US5283478A (en) * 1990-12-28 1994-02-01 Sgs-Thomson Microelectronics, S.R.L. Fast capacitive-load driving circuit particularly memories
US5430388A (en) * 1993-02-01 1995-07-04 Inmos Limited Output noise reduction circuit
EP0610064A3 (fr) * 1993-02-01 1994-11-23 Inmos Ltd Commutation à transistor.
EP0610064A2 (fr) * 1993-02-01 1994-08-10 STMicroelectronics Limited Commutation à transistor
EP0653841A1 (fr) * 1993-11-15 1995-05-17 International Business Machines Corporation Commutateur à haute puissance comportant commande de courant de démarrage
US6424187B1 (en) 1996-12-18 2002-07-23 Hitachi, Ltd. Driving apparatus for power MOS transistor
DE19755653C2 (de) * 1996-12-18 2003-06-18 Hitachi Ltd Schaltung eines Kraftfahrzeug-Generators zum Aufladen einer Batterie
WO1999035743A1 (fr) * 1997-12-30 1999-07-15 Siemens Aktiengesellschaft Procede et dispositif de coupure en cas de court-circuit et de surcharge au moyen d'un composant a semiconducteur
FR2878065A1 (fr) * 2004-11-18 2006-05-19 St Microelectronics Sa Circuit de decharge d'une charge electrique, et etage de sortie de puissance comprenant un tel circuit de decharge pour la commande de cellules d'ecran plasma
WO2006053966A1 (fr) * 2004-11-18 2006-05-26 Stmicroelectronics Sa Circuit de decharge d’une charge electrique, et etage de sortie de puissance comprenant un tel circuit de decharge pour la commande de cellules d’ecran plasma
US7843238B2 (en) 2004-11-18 2010-11-30 Stmicroelectronics, Sa Circuit for discharging an electrical load, power output stage comprising such a discharge circuit for the control of plasma display cells, and related system and method
US8174295B2 (en) 2004-11-18 2012-05-08 Stmicroelectronics, Sa Circuit for discharging an electrical load, power output stage comprising such a discharge circuit for the control of plasma display cells; and related system and method

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AU2080088A (en) 1988-11-04
WO1988008228A3 (fr) 1989-01-26

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