USRE39076E1 - Apparatus and method for controlling brushless electric motors and position encoders and indicating position thereof - Google Patents

Apparatus and method for controlling brushless electric motors and position encoders and indicating position thereof Download PDF

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USRE39076E1
USRE39076E1 US08/588,301 US58830191A USRE39076E US RE39076 E1 USRE39076 E1 US RE39076E1 US 58830191 A US58830191 A US 58830191A US RE39076 E USRE39076 E US RE39076E
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Prior art keywords
motor
rotor
stator
values
control means
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Johann von der Heide
Uwe Muller
Micheal Hermann
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Papst Licensing GmbH and Co KG
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/20—Arrangements for starting
    • H02P6/22—Arrangements for starting in a selected direction of rotation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/08—Reluctance motors
    • H02P25/086—Commutation
    • H02P25/089—Sensorless control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/006—Controlling linear motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/14—Electronic commutators
    • H02P6/16—Circuit arrangements for detecting position
    • H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
    • H02P6/185—Circuit arrangements for detecting position without separate position detecting elements using inductance sensing, e.g. pulse excitation

Definitions

  • the invention relates to a motor or a position indicator (position sensor) comprising an apparatus or method for controlling a brushless electric motor.
  • the term motor is here understood to mean also e.g. a linear motor, whose position is to be detected. Both a rotary motor and a linear motor has as its movable part a rotor.
  • position sensors are known and are e.g. used for determining the position of motor elements. In most cases there is an associated linear acting principle for a rotary acting principle. The most important classes of such position sensors or indicators are based on the following:
  • the electrical angular position is measured in electrical degrees, e.g. 20° el.
  • electrical degrees e.g. 20° el.
  • the electrical degrees of the rotor position correspond to the mechanical degrees.
  • the described measuring methods agree within a range of 360° el, i.e. they determine the electrical angular position of the rotor.
  • the measured electrical angular position 90° el represents a rotor position of 90° el or (90+360)° el, which is unimportant for the commutation control.
  • the invention is suitable for internal rotor-type motors, external rotor-type motors, motors with a planar air gap, motors in which the permanent magnetic rotor rotates and the motor winding is stationary or conversely motors in which the motor winding rotates and is e.g. supplied across slip rings and the permanent magnetic part is stationary and other motor construction types, in the manner illustrated by examples hereinafter.
  • the position determination methods are based on the fact that the motors in question are not ideal, i.e. despite constant, distinct electrical quantities, there are normally variable positional and angular dependencies of the mechanical quantities.
  • interest is attached to those physical effects, which are based on the position-dependent change of electrical or electromagnetic parameters of the motor and which can be established by checking with electrical means only.
  • Such a check or measurement of interesting parameters can e.g. be performed in that one or more suitable additional measuring coils are fitted at one or more points within the motor.
  • the coils have complex resistors, which are characterized by ohmic resistances and inductance values and said complex resistors are normally also frequency-dependent.
  • one or more of the indicated quantities are dependent on the position taken up by the rotor in the case of linear and rotary motors.
  • EP-A-251785 and EP-A-171635 Examples of such arrangements are disclosed by EP-A-251785 and EP-A-171635.
  • EP-A-251785 essentially only the additional power stages are required for controlling the motor phases, as well as special components as electronics for controlling the motor.
  • the circuit suffers from the disadvantage that a range of 360° el to be measured is split up into two undistinguishable ranges of in each case 180° el. This is due to the fact that symmetrically to the angle of 0° el or 180° el, in the following range of 180° to 360° el there are equally large function values, i.e. it is impossible to know whether a basic value of 0° or 180° el must be added to the calculated rotor position (FIG. 22 A). Moreover the precision of the position determination is limited to approximately ⁇ 60° el.
  • This circuit gives a rotor position determination within a range of 360° el without ambiguity.
  • Another difficulty in the known methods is to exclude type-typical phase errors of ⁇ 180° el in the case of a rotor position determination. Such errors mean that if the rotor is further rotated by 180°, 360° el, etc., the typical position information remains largely unchanged compared with the original position.
  • a problem of the invention is to provide accurate, inexpensive and insensitive sensors for rotary and linear movements. Another problem of the invention is to ensure reliable motor operation through reliable commutation processes. Another problem of the invention is to bring about a certain starting direction of such motors.
  • the invention makes it possible to increase the usable speed range compared with the prior art. According to the invention using microelectronics means and the relevant software, it is possible to obtain an improved accuracy and reduce the relative errors during position determination.
  • FIG. 1 The basic structure of the control electronics for a motor.
  • FIG. 2 The basic structure of a commutating processor.
  • FIG. 3 The non-linear behaviour for ferromagnetic core materials for coils and relative permeabilities/inductances.
  • FIG. 4 The field pattern for a d.c. motor in no-load operation.
  • FIG. 5 The field pattern of a d.c. motor when current is applied.
  • FIG. 6 The diagrammatic construction of a reluctance motor with six coils and a rotor with quad symmetry.
  • FIG. 7 A 3-T-armature with three coils and two magnetic poles.
  • FIG. 8 A stator of an external rotor-type motor with stator plates and stator phases.
  • FIG. 9 The field pattern of an internal rotor-type motor with four poles and six stator phases.
  • FIG. 10 The motor according to FIG. 9 for illustrating the magnetic flux in different stator elements.
  • FIG. 11A A further flux pattern for a non-symmetrical rotor position of the motor of FIG. 9 .
  • FIG. 11B The construction of the rotor for a flat motor with a different construction of the individual rotor poles.
  • FIG. 11C The construction of a rotor as in FIGS. 8 to 11 A with different magnetization of individual rotor poles.
  • FIG. 11D A motor construction with a different construction of the air gap under individual stator poles.
  • FIG. 11E Diagrammatically variants for producing saturation in the stator iron.
  • FIG. 12 The current application pattern for measured value obtention.
  • FIG. 13 The derivation of a mean value from several measurements.
  • FIG. 14 The linear arrangement of several position-dependent mean values.
  • FIG. 15 The position comparison between two phase-displaced curves.
  • FIG. 16a Different methods for carrying out the position comparison of the 16 C two phase-displaced curves.
  • FIGS 17 A The spatial representation of a measurement result comprising and 17 B three individual measurements and its evaluation.
  • FIG. 18 The bidimensionally oriented representation of a measurement result with three individual measurements and the evaluation thereof.
  • FIG. 19A A diagram for the current application to three coils by means of six switches.
  • FIG. 19B The resulting current flow in the coils of FIG. 19A for two closed switches.
  • FIG. 19C A simple current application sequence and six associated current paths.
  • FIG. 19D A diagram for determining the fundamental wave of a function and in fact graphically with six values m 1 to m 6 (corresponding to a motor with three or six phases L 1 to L 6 , as shown e.g. in FIG. 9 ).
  • FIG. 20 The representation of a vector sum, calculated according to the above calculating specification of FIG. 19 D.
  • FIG. 21 A flow diagram for phase and amplitude determination of the first harmonic of a function determined by six interpolation nodes.
  • FIG. 21A The approximate and periodically continued curve pattern by six interpolation nodes m 1 and m 6 and the phase position of the associated first harmonic.
  • FIG. 22A The representation for inductance-dependent values relative 22 B & 22 C to the electrical angle.
  • FIG. 23 The path of a correction function for the association of an actual value with a characteristic value of the rotor position.
  • FIG. 24 A flow diagram for the current application to a motor or position sensor.
  • FIG. 25 A diagram of the different possibilities for controlling the start.
  • FIG. 26 A flow diagram showing the sequence of the measurement of the rotor position alternating with the current application for a motor.
  • the inventive idea of solving the set problem comprises the following parts.
  • FIG. 1 The basic structure of such an inventive apparatus is shown in FIG. 1 . Apart from the actual motor 10 , it comprises an associated current application and output stage unit 26 , a commutating processor 100 and the controlling or regulating microprocessor 23 . Hereinafter the term processor is used instead of microprocessor. As shown in FIG. 1 , the motor 10 can e.g. be a three-phase, commutatorless, d.c. motor.
  • the invention is not restricted to such motors and, as shown in subsequent drawings, can also be advantageously used with other motor constructions. If necessary, the invention also covers completely renouncing motor operation and the use of the motor exclusively in sensor operation for determining positions. That is, the motor is operated at essentially no load as a position indicator.
  • the output stage unit 26 essentially comprises a switching unit for supplying current to the individual motor phases.
  • the output stage unit 26 has several control inputs ( 261 to 266 ) for operating the not shown, electronic switches contained therein. These switches supply the motor with current 10 via the individual phases in accordance with the signal combination of the switch inputs 261 to 266 .
  • the resulting total motor current is measured by means of a random current sensor, e.g. the position resistor 1011 in FIG. 2 and via the current sensor output 267 is supplied as a measurement signal to the input 101 of the commutating processor 100 .
  • a random current sensor e.g. the position resistor 1011 in FIG. 2 and via the current sensor output 267 is supplied as a measurement signal to the input 101 of the commutating processor 100 .
  • regulating or control unit which ensures that a predetermined current intensity of a predetermined level is applied to the individual phases of the motor 10 , e.g. by using conventional, linearly acting controller stages.
  • switching controllers it is also possible to use conventional switching controllers for this purpose.
  • the function of the commutating processor 100 is to accept current sensor signals 267 and voltage signals 111 , 112 and 113 of the three motor phases L 1 , L 2 , L 3 ( FIG. 2 ) and to calculate commutating signals therefrom.
  • the commutating processor 100 has analog and digital control and communication connections 102 to 110 to the controlling processor 23 .
  • the commutating processor 100 can alone ensure a correct operation of the motor 10 , i.e. without the microprocessor 23 , but in interplay with the output stage unit 26 .
  • processor 23 It is also possible in a processor-controlled operation to carry out planned movement sequences of the motor or, whilst renouncing motor operation, to only carry out a position detection.
  • the interplay of processor 23 , commutating processor 100 and output stage unit 26 is necessary.
  • one function of the processor 23 is to establish times at which current sensor signals 267 are to be determined, said signals being supplied to the processor 23 across the current line 102 .
  • the commutating processor 100 can also require the microprocessor 23 across a line 104 to accept a measured current value or perform other processor activities.
  • the essential function of the processor 23 is to calculate the correct position of the rotor of the motor 10 with the aid of current sensor data obtained in this way.
  • the thus calculated data can be displayed e.g. on a display, but can also be supplied for further processing to a controlling computer system.
  • This positional information can be supplied across the position lines 108 to 110 to the commutating processor 100 , which in turn can carry out a correct commutation of the motor 10 on the basis of this information.
  • the commutating processor 100 essentially comprises the following unit.
  • the central component of the commutating processor 100 is the commutating logic 1002 , which calculates from external, predetermined position information fed in via the line 108 - 110 , or from coil voltages, which are to be measured via lines 111 to 113 , firstly the commutating time, i.e. the first time at which a commutation of the motor currents is to take place.
  • a commutating switching signal or control signal is generated, which is supplied via the signal line 261 to 266 to the output stage unit 26 .
  • such signals can naturally also be transmitted in coded form.
  • the commutating logic 1002 is also influenced by a commutating control unit 1005 , which can react to different external signals. Such signals can e.g. be the start mode signal 107 or the microprocessor presence signal 106 . Exception signals C 1 to C 4 can also influence the control unit 1005 , which is also influenced by signal voltages, which can be derived from the coil voltages of the motor 10 applied to inputs 111 to 113 .
  • an attenuator 1008 is provided, together with a voltage comparison unit 1007 and a signal delay unit 1006 .
  • the signals from phases L 1 to L 3 firstly reach the attenuator 108 , pass from there to the voltage comparison unit 1007 and finally from there to the signal delay unit 1006 , where they are passed on to the commutating logic 1002 and the commutating control unit 1005 .
  • emf induced voltage
  • Another device of the commutating processor 100 is the current detection unit 1009 , whose function is to detect, amplify and prepare the current sensor signals 267 . This takes place in that at given times the current sensor signals 267 are amplified and stored. In this way rapidly varying current sequences can, so-to-speak, be frozen and supplied to the microprocessor 23 across the current output and the current signal line 102 for further processing. The processor 23 then has sufficient time to evaluate these current signals, because the stored signals do not initially change further.
  • the commutating processor 100 normally has external switching means, such as resistors and capacitors, which are not shown in FIGS. 1 and 2 .
  • commutating processor 100 Further details of the commutating processor 100 are a starting oscillator 1004 and a counter 1003 for the random production of preset times or preset commutating patterns.
  • the actual function of the commutating processor 100 is, as stated, the evaluation of the induced voltages (emf) at the motor phases L 1 to L 3 , such as are generated e.g. under no-load conditions.
  • a position information obtention by appropriate current application to stator phases is brought about in that through a current application of differing levels the iron components of a magnetic circuit belonging to the coils of this phase are brought close to or at saturation.
  • L the overall inductance of a participating coil
  • the non-linear connection between the field strength H and magnetic flux density (induction) B can be used for establishing current-dependent, effective inductances of the participating stator coils.
  • stator coils are not only influenced by the coil current level, but also by the polarity of the permanent magnet of a rotor, said permanent magnets facing the stator coils. This is due to the fact that through the permanent magnets, there is a premagnetization of the coil cores, so that as a function of the polarity of the coil current there is a saturation of the coil core either faster or slower than in the neutral case, i.e. without permanent magnetic premagnetization. In the case of a conventional design of magnetic circuits of motors of a general type an attempt is generally made to keep the magnetic flux density in the iron parts well below saturation, so as to keep the stray flux components low.
  • FIG. 22A shows the path of a coil inductance over the rotor rotation angle.
  • this function is periodic with a period of 180° el.
  • FIG. 22C shows the same conditions as in FIG. 22B , i.e. the inductance plotted in polar coordinates and a function of the motor rotation angle.
  • this phase angle is surprisingly also a measure of the rotor position. It is clear that the dependence of the inductance of a phase on the rotor position should be as great as possible, so as to be able to perform a clear and unambiguous measurement.
  • An additional inventive possibility for bringing about this dependence comprises designing rotor permanent magnets in such a way that there is a non-uniformly high effective field strength of these permanent magnets. This can e.g. take place in that on magnetizing the rotor a non-uniformly high magnetization is carried out or, after magnetization, there is a planned demagnetization of individual magnetic poles.
  • Another possibility consists of following one combination of equally strong north and south pole rotor magnetic poles by another identical combination, but whose effective field strength is reduced by a constant amount.
  • FIGS. 11B and 11C e.g. the rotor magnetic poles 501 , 502 are more strongly magnetized than the rotor magnetic poles 503 and 504 .
  • FIG. 11E represents a linear motor, which has as the rotor element a permanent magnet with north poles 1101 and south poles 1102 . Opposite the magnetic poles are provided the stator unit 1103 with stator pole shoes 1104 . By means of specially shaped stator webs in elongated form ( 1105 ) or short form ( 1106 ), the stator pole shoes are connected to the return plates 1108 .
  • the stator webs are wound with corresponding coils 1107 .
  • the return plates 1103 can have a reduced cross-section 1109 or an increased cross-section 1110 . It is also appropriate to insert an air gap 1111 .
  • the advantage of these arrangements is that through the varyingly high magnetization within the magnetic circuit, an additional, position-dependent information (encoding) is introduced into the motor or position sensor. This permits a more precise and geometrically more extensive position determination, which allows an absolute position determination within the full rotation angle of 360° el.
  • the current I 1 is connected in at t o and measured at t 1 ).
  • this process is repeated in a further stator coil, without the first stator coil being disconnected.
  • FIG. 12 shows this time connection for the current application to a selected number of three coils, whose current is designated I 1 , I 2 and I 3 . If there are further coils, it is possible to use for a more extensive information obtention, one or more additional processes of the same type on a fourth or further stator coils. After current has been supplied to all the stator coils in question, disconnection either takes place completely and simultaneously or, as shown in FIG. 12 for the pulse sequences 1 , 2 , and 3 , there is a clearly defined disconnection procedure. Also in these two cases it is possible to perform time and/or amplitude measurements, e.g. the amplitudes M 4 , M 5 and M 6 at the times t 4 , t 5 and t 6 .
  • time and/or amplitude measurements e.g. the amplitudes M 4 , M 5 and M 6 at the times t 4 , t 5 and t 6 .
  • a first pulse sequence 1 is obtained a first set of measured values, which in the simplest case consists of a single value. If two measured values are obtained per coil, as shown in FIG. 12 , then for a motor with three coils there are six measured values M 1 to M 6 . It is also naturally possible to obtain additional measured points from rising and falling slopes and to allow them to influence the further calculation. Normally attempts are made to select a few representative measured values from the plurality thereof.
  • current application can e.g. commence with coil 2 and shortly thereafter current application to coil 3 and then coil 1 takes place.
  • the current application to the individual coils is broken off at given times.
  • the procedure is repeated again, but beginning with coil 3 , followed by coil 1 and then coil 2 . If there are more than three coils, then more complicated application sequences can be performed. Therefore the specification for the quasi-simultaneous current application to further coils will correspondingly relate to coils having comparable spacings with respect to the next coil to which current is to be applied.
  • current can be applied to a directly adjacent coil.
  • the second current application process can also relate to a coil which is at some distance from the coil to which current was first applied.
  • a fixing of the most useful coil current application sequence must take place individually as a function of the motor type. In order to obtain complete data the process is repeated several times.
  • Current application begins in each case with a different stator coil.
  • Current obtention is complete if there is a cyclically produced measured value set on the basis of the same diagram for all the motor stator coils.
  • evaluation is carried out (as a function of the efficiency of the evaluation unit evaluation can naturally also take place during the obtention of the measured value sets).
  • a special microprocessor or controller 23 whose program is tailor-made to the position detection problem to be solved, but which can also be used for regular motor operation.
  • a standard processor 23 with an associated, externally positioned expander circuit or commutating processor 100 .
  • Another possibility consists of combining units 23 and 100 , so that a standard processor unit is present on the same silicon surface, which is supplemented by customer-specific additional functional units.
  • the measured values determined in a pulse train e.g. 1 in FIG. 12 are appropriately combined to form a single, representative measured value.
  • the measurement accuracy of the individual measured values M 1 to M 6 can be of varying levels, it is appropriate to carry out a weighting in the case of the necessary mean value formation.
  • a similar mean value formation with the same coefficients can be used for the pulse train 2 on the measured values M 21 to M 26 , which gives a mean value m 2 .
  • the mean values m 1 to m 3 implicitly contain the sought position information, which must still be calculated therefrom in explicit form.
  • pulse trains 4 , 5 , 6 , etc. If there are further pulse trains, e.g. not shown pulse trains 4 , 5 , 6 , etc., then the associated mean values m 4 , m 5 , m 6 , etc. are to be determined by comparable procedures.
  • each measured value multiplet obtained in this way it is possible to place a substantially identical periodic curve K, as shown e.g. in FIG. 14 or by reference 161 or 163 in FIG. 15 , but whose phase position must be varied until the measured values, here e.g. m 1 to m 3 , are located on the curve (FIG. 15 and FIG. 16 A).
  • the measure of the periodicity for this curve is designated 144 in FIG. 14 .
  • the measure of said phase displacement constitutes a characteristic value for the actual motor rotor position.
  • FIGS. 19A and 19B One form of determining six values m 1 to m 6 is shown in FIGS. 19A and 19B .
  • Three motor coils L 1 , L 2 , L 3 with terminals R, S and T are supplied with current via a switch set SW 1 to SW 6 (reference numerals 178 to 183 ).
  • SW 1 to SW 6 In each case one of the switches SW 1 to SW 6 and one of the switches SW 4 to SW 6 is simultaneously closed.
  • the resulting current flows are defined in FIG. 19 B.
  • the current flow is given for in each case two switches and in this way six switching possibilities are provided.
  • a current application for determining the rotor position then takes place e.g. in accordance with FIG. 19 C.
  • SW 1 and SW 5 are used for producing the current flow i 1 which, rises in accordance with a rising slope 190 , whose intensity is measured after the time interval 193 and gives a value m 1 .
  • the current is interrupted by opening the switches, which leads to a disconnection slope 191 .
  • This procedure with relatively fast connection and disconnection times is repeated according to the other switch combinations shown in FIG. 19B for five further current application i 2 to i 6 and in each case following the time interval 193 the associated value m 2 , m 3 , . . . , m 6 is determined and buffer stored for further evaluation.
  • the aim is to place a known curve through the measured values obtained in this way and the phase is to be displaced in such a way that the measured values are located on or as close as possible to the curve.
  • This phase displacement can be mathematically determined in several different ways.
  • One possibility is e.g. to initially determine a curve K, which interconnects the measured values in an optimum manner.
  • This procedure has the advantage that on comparing a curve found with a reference curve not only a few, but a random number of points can be compared with one another (FIG. 14 ). In this way the individual points of a curve found can be displaced in fine steps until both curves to be compared have an almost congruent position.
  • spline function which represents an optimized continuous connection between all the points of the measured values set.
  • the generation of such functions differs compared with the first-mentioned process in that the spectrum obtained is modified in a planned manner, or the functions are approximated e.g. by third degree polynominals. In this way a limited number of points n 1 . . . n m of a mean value set is transferred into a random number of points of matching spline functions.
  • the difference between in each case two associated function values is determined and squared (curve 164 ).
  • the result is added to a sum total, i.e. the curve 164 is integrated between 0 and ⁇ 1.
  • the similarity is detected by means of the sum total. If the sum found, i.e. the integral of the curve 164 , is zero, then identical functions are present and the phase difference is zero. Therefore small sum values represent a large similarity of the functions, whereas large sum values represent small similarities of the functions.
  • summation or integrated summation must take place over a standard interval 0 to ⁇ 1, which is at least as long as that defined e.g. by the reference function.
  • the position-characteristic determination is now carried out in such a way that the first summation process is followed by others.
  • one of the two functions including their supplements or additions on both sides, are successively further displaced by a phase angle xi 114 (FIG. 16 C).
  • the correlation is at a maximum, so that a numerical value for the phase displacement found in this way can be used as the rotor position characteristic.
  • Another process consists of reciprocally displacing the reference and comparison functions until the sum of the absolute quantities of the differences of a number of function values of the reference and comparison function is as small as possible, this process being illustrated in FIG. 16 A.
  • the two functions to be compared have in each case a maximum 165 or 166 and the phase displacement is designated d.
  • d the phase displacement
  • a minimum comparison amount namely the integral of the function 168 between 0° and 360° el. amounts to a good coincidence of the curves. With a complete coincidence the difference surface amount is 0.
  • the numerical values of the rotor position obtained in this way are clearly more accurate than the determination of a maximum value of different interpolation nodes of the function, as described in EU-A-251785.
  • the calculating expenditure and effort for obtaining this improved precision is greater, because firstly it is necessary to determine the spline function values for interpolation nodes and then the optimum curve position must be determined by iterative curve comparison operations.
  • m 1 , m 2 , etc. are represented in graphic form, so that a surface is produced in the form of an irregular or regular polygon.
  • a position characteristic is to be calculated from three values only, i.e. the mean values m 1 , m 2 and m 3 .
  • the associated inventive rules for calculating a position characteristic in this case comprise the three measured values m 1 , m 2 , m 3 being plotted as axis portions in a three-dimensional cartesian coordinate system.
  • a coordinate system which is spanned by the three unit vectors i ( 801 ), j ( 802 ) and k ( 803 ), is shown in FIG. 17 A.
  • the latter also shows the space diagonal 804 passing through the origin and the point with the coordinates i, j, k, i.e. the space diagonal is defined by the origin of the coordinate system with the associated unit vectors i, j and k ( 801 , 802 , 803 ) and the vector sum i+j+k.
  • a first, second and third measured value m 1 , m 2 , m 3 of a trio of values is in each case plotted in the direction i, j or k.
  • a triangle 81 is defined, whose (surface) centroid S, 82 , as a function of the trio of measured values to be used as a basis, is of varying form, because in the case of a different position of the rotor position to be determined different trios of measured values have to be evaluated.
  • This surface centroid S can be determined by drawing or also by calculation/mathematically and is located on a curve 810 (FIG. 17 B), which essentially passes in cycloid manner around the space diagonal 804 and which is shown on a larger scale in FIG. 17 B.
  • the space diagonal 804 passes through the triangle 81 in a centre Z, reference 83 .
  • the centre 83 defines a straight line 804 located on the triangle 801 , based on a straight line 806 between the center 83 and an angle point or corner, e.g. m 1 of the triangle 81 , a clearly defined angle 807 in the plane of the triangle. This angle 807 is used as the position characteristic for the rotor position and further processed.
  • said three measured values can also be evaluated in a bidimensionally oriented representation (FIG. 18 ).
  • the three axes of space are projected onto three lines, which emanate from the origin of an x-y coordinate system and in each case have an angle difference of 120°. Starting from the origin, distances are plotted in the direction of these axes and are proportional to the mean values m 1 , m 2 , m 3 .
  • a triangle is spanned, whose centroid S does not necessarily coincide with the origin 1802 of the three lines. In known manner, this centroid can be constructed by drawing or mathematically.
  • a line between the centroid S and the origin of the three axes will assume a different angle 1801 with respect to one of these three axes.
  • the position of the centroid can be determined in known manner by construction of the median or alternatively using the centroid formula, which must here be applied e.g. to three equal-weight elements m 1 , m 2 and m 3 . According to this calculating rule the portions in the X and Y direction ( FIG. 18 ) can be separately calculated.
  • centroid analysis process can be looked upon as a special case for all processes based on Fourier analyses and transformations. Also in the case of such analyses and transformations a corresponding vector summation is carried out for the values for the amplitude and phase of the first harmonic of a function defined by several interpolation nodes.
  • FIG. 19D shows a corresponding example for the evaluation of e.g. three or six stator coils, for which the six values m 1 to m 6 are to be determined in accordance with the procedures represented in FIGS. 12 and 13 .
  • m 1 M 1
  • m 2 M 21
  • m 3 M 31
  • FIG. 19D shows the “complex plane”, in which the x axis corresponds to the real part and the y axis to the imaginary part.
  • FIG. 20 can also be looked upon as a complex plane.
  • the first harmonic i.e. the sinusoidal fundamental wave, which is associated with a predetermined number of measured values of a periodic function, is calculated in amplitude and phase by the addition of associated complex numbers.
  • This addition is carried out in much the same way as the addition of vectors and is e.g. represented in FIG. 20 for a so-called 6-point DFT (discreet Fourier transformation). If there are fewer or more measured values (measurement points), correspondingly modified procedures apply, as are known from the theory of Fourier transformations.
  • the measured values which are generally present as real quantities, e.g. as measured current values, are to be converted into the correct complex numbers, as is illustrated in FIG. 19 D.
  • a linear stretching of the unit cubes takes place in a successive order and in each case by the particular factor m 1 to m 6 , so that the points m 1 to m 6 of FIG. 19D are obtained in the complex plane.
  • the vectors emanating from the coordinate origin up to the said points m 1 to m 6 are shown in FIG. 20 , but in a different form.
  • the expression “stretching the unit cubes” is understood to mean that they are multiplied by the value m 1 to m 6 .
  • the value m 1 is greater than 1 and therefore the particular vector m 1 is greater than 1 (from the amount standpoint). Therefore m 3 is smaller than 1 and the particular vector m 3 becomes smaller than 1.
  • the vectors m 1 , m 2 , m 3 , m 4 , m 5 , m 6 are given the reference numerals 201 to 206 in FIG. 20 . To illustrate their vector sum they are also strung together in chain-like manner.
  • the x component of the resulting vector is designated x 7 and is formed from the x parts shown, whilst the y component is formed from the y parts and is designated y 7 .
  • the resulting vector has the end point Z 0 , which can once again be looked upon as a complex number with the real part x 7 and the imaginary part y 7 .
  • the amount R and the phase phi of said complex number can be determined according to known calculation rules from the real part and the imaginary part, e.g. the amount of R is equal to the square root of the sum of the squares of the real part and the imaginary part, i.e. ( x 7 2 + y 7 2 ) .
  • An associated calculating rule for the processing of the values m 1 to m 6 by a digital computer e.g. a microprocessor, is given as a flow diagram in FIG. 21 .
  • a digital computer e.g. a microprocessor
  • a first step S 40 are known values m 1 to m 6 determined by another program part are filed in a memory, e.g. a stack memory.
  • a memory e.g. a stack memory.
  • steps S 41 and S 42 different quantities are given initial values and in particular the storage locations with the allocation addresses SUM_R and SUM_I are set to zero and form the memories for the real and imaginary parts.
  • steps S 43 to S 58 values are added or subtracted in these memories, being derived from the values m 1 to m 6 .
  • step S 43 and S 44 initially the value m 1 is retrieved from the stack memory and then added to the content of the memory SUM_R, which then assumes the value m 1 .
  • step S 45 the value m 2 is obtained from the stack memory and is divided by two. This result is added to the content of the memory SUM_R and multiplied by the factor ⁇ 3 and then added to the content of the memory SUM_I.
  • This process is carried out in similar form in the following steps and in steps S 51 and S 52 the division by the factor two is omitted and the procedure is much as in steps S 43 and S 44 , but m 4 must be subtracted from the content of the memory SUM_R.
  • the further calculating steps of FIG. 21 are intended to determine the angle of this vector relative to the x axis and also its amount.
  • steps S 59 to S 64 of FIG. 21 for determining the amount initially the contents of the memories SUM_R and SUM_I are individually squared and then from the squares is formed the sum. From this sum is drawn the square root and this value represents the sought amount of the vector and is therefore stored in the storage location designated BETRAG or amount.
  • steps S 65 to S 68 firstly the contents of the memories SUM_R and SUM_I are obtained and the sign of these values is noted.
  • the sought main angle value phi is calculated from the quotient of SUM_I/SUM_R and the arc tan function used thereon.
  • the final angular value is then determined in accordance with known transformations on the basis of the stored sign values within a range 0° to 360°.
  • the last-described process can be compared with calculation procedures and processes of discreet Fourier transformation (DFT).
  • DFT discreet Fourier transformation
  • FIG. 21A This is illustrated by FIG. 21A , where at A) are shown the values m 1 to m 6 in the form of 6 interpolation nodes of a periodic function 230 , which are subsequently repeated. Comparison shows that the last-described process can be looked upon as DFT for the determination of the amount R and phase phi o of the first harmonic of this function 230 , which is also defined by equidistant interpolation nodes and which are repeated outside the interpolation nodes in periodic manner, i.e. continued in the same form.
  • FIG. 21A shows a B) the determined first harmonic with its amplitude R and its phase angle phi o, which essentially gives the rotor position and which is e.g. calculated according to the flow diagram of FIG. 21 .
  • the angle or phase phi of this first harmonic graphically represented in FIGS. 20 and 21A and calculatable according to FIG. 21 is of interest and represents the sought position characteristic for the motor rotor position.
  • the latter shows the cartesian coordinates “position actual value” and “position characteristic value”.
  • the functional association of the actual and characteristic values is given by the assignment curve 72 , as shown by the example of the abscissa and ordinate values of the function value 71 .
  • the assignment curve 72 generally does not have a linear path and consequently differs from a straight line 73 with optimum matching to the function values.
  • the deviations of individual interpolation nodes of the function 72 represented on the example of the paths 71 to 76 , can be kept in a correction table. According to a known mathematical process, one or more matching lines can be implied through a number of interpolation nodes.
  • a matching line 73 which need not necessarily be characterized by a best matching to the interpolation nodes, as is known it is sufficient to have the coordinate intersections 74 and 75 .
  • Additional correction information can be determined from the previously not considered higher harmonics of the fundamental measured value sequences, provided that correspondingly large numbers of measurement points are available.
  • a further correction function of this type can be derived from the amplitude of the first harmonic.
  • An additional correction function or table is required for this procedure and the sought rotor position is then calculated from the angle phi ( FIG. 20 ) and an additive or multiplicative correction value, which is directly dependent on the amount (amplitude) of the first harmonic. Correction functions or tables are to be fixed by an experiment carried out beforehand.
  • step S 1 it is firstly necessary to establish whether or not the stepping motor can be started, because the entry into a basic position can take place by an inadmissible rotation direction ( FIG. 24 , S 1 ). If stepping motor operation is permitted, then e.g. a stepping pulse is supplied ( FIG. 24 , S 2 ), which increments the counter 1003 (FIG. 2 ). In step S 3 the commutation logic 1002 produces a new commutation pattern, which is supplied to the output stage unit 26 , which in a step S 4 ensures the energizing of the motor. In the following step S 5 , it is established by emf observation whether the rotation direction is correct.
  • step S 6 the rotation direction can be maintained, otherwise commutation sequences must be ensured which bring about a rotation direction change. This sequence is initially repeated until a first speed limit is reached. If alternatively the motor start in the correct rotation direction is immediately necessary, steps S 7 to S 9 are performed. Thus, by the rotor position determined in step S 7 , as is also the case in the regular operation of a commutatorless d.c. motor, it is possible in one step S 8 to generate and emit the necessary correction pattern, which brings the motor into the correct rotation during the step S 9 . A decision is made in the step S 10 as to whether this process must be repeated or whether a speed limit has been reached.
  • step S 11 to S 14 a further motor acceleration is ensured. This is achieved in that after detecting the zero passages of the motor counter-emf, at an appropriate time and in accordance with a new-calculated commutation pattern, power is supplied to the motor. In this way the motor is run to the maximum attainable speed.
  • the motor speed or position is regulated or controlled according to the inventive process in the following way.
  • the precise determination of the position of a rotor using the inventive process can be performed with a sufficiently high repetition frequency, e.g. 100 times per second, in order to record deviations from a desired motor rotor position and to keep same as small as possible by an active counteraction (control principle).
  • control may only be a scanning control. Numerous publications are known in this field.
  • the attainable control quantity is influenced by the scanning time and also the different system time constants, as well as by the control algorithm used.
  • Another inventive process for controlling the speed comprises only carrying out one rotor position measurement and then carrying out the current application steps.
  • the duration of the latter will generally depend on when the next motor commutation is to be carried out and will constantly decrease on accelerating the motor.
  • an advantageous process when calculating the necessary pulse shortening is to take account of the number of already performed current application cycles. Generally, with increasing rotor speed, the pulses are shortened.
  • FIG. 25 another possibility consists of making the current application period duration linearly dependent on the obtained overall angle or position, e.g. (starting with a relatively large value for the current application time), in accordance with the already attained position, the current application time is continuously and linearly ( 131 ) reduced, up to e.g. a minimum value, below which no further drop must take place.
  • step S 21 a time counter is set to zero and then periodically increments its count automatically and in quasi-continuous manner, i.e. with a predetermined short cycle time.
  • step S 22 With the aid of the previously described processes, in the following step S 22 and without the aid of external sensors, at least the electrical position is determined.
  • the following steps relate to the running-up phase, in which the kinetic energy of the motor constantly increases.
  • step S 23 a suitable commutation signal is generated and is used in step S 24 for motor current application, in such a way that the rotor acceleration takes place in the desired direction.
  • the current is initially not interrupted.
  • step S 25 the actual time t is determined, by means of which in the next step S 26 the provided current application time tau is determined according to a predetermined calculation rule or according to a predetermined table stored in the processor. Waiting takes place in the next step S 27 until the current application time determined in step S 26 has elapsed, so that in step S 28 the motor current can initially be switched off.
  • step S 29 As the acceleration phase only lasts for a certain time, a check is carried out in step S 29 to establish whether the provided acceleration time t LIM has been exceeded or not. If this is not the case, a continuation occurs with step S 23 and so on. However, if the time t LIM has been exceeded, then the program is branched into the constant speed phase and a further commutation signal is supplied in step S 30 . This step is followed by the step S 31 which, as a result of the generated commutation signal, ensures the associated correct current application to the motor.
  • step S 32 waiting takes place in the same way as in step S 27 to establish if the motor current is to be temporarily switched out again, which is possible after a time corresponding to tau o time units, the constant quantity tau o being known to the processor.
  • step S 34 A check is made in the following step S 34 to establish whether any condition has led to the generation of a stop signal. If this is not the case, there is again branching to step S 30 , in which a new commutation signal is supplied, etc. If this is not the case, i.e. when a stop signal is present, branching takes place to the end of the program, i.e. step S 35 . No further motor current is supplied and the motor stops.
  • the use of the invention is not restricted to standard commutatorless d.c. motors, such as are preferably used with 4 or 8 magnetic poles and e.g. 6 or 12 coils ( FIGS. 8 , 9 and 10 ). It is also possible to extend the embodiment to all motors, whose inductance varies in position-dependent manner, e.g. simple motors with so-called 3-T-armatures ( FIGS. 4 , 5 and 7 ) and in particular all types of reluctance motors with or without permanent magnets (FIG. 6 ).
  • FIG. 4 firstly shows a conventional 3-T-armature 44 , which is located in the magnetic field of a permanent magnetic pole pair with north pole 40 and south pole 41 and which is rotatable about the centre 46 .
  • the leg which is nearer to the north pole 40 than the legs 43 and 44 to the south pole 41 , is exposed to a higher magnetic flux density (10 flux lines) than the two other legs (approximately 5 flux lines).
  • a coil 47 increases the magnetization of leg 42 when current flows through and an iron part of said leg can be saturated with less current than if no premagnetization existed or only existed to a slight extent through the permanent magnets 40 , 41 , as is the case with leg 43 in FIG. 5 .
  • a higher current application to such a coil 47 is required, in order to achieve a magnetic saturation of the leg 42 in the opposite direction.
  • current application processes of this type are subject to a first time constant due to the initial inductance present, the time required for magnetization up to saturation with a desired polarity will differ.
  • FIG. 7 shows a simple spatial arrangement of such a 3-T-armature motor.
  • the contacting of the coils 47 and 48 can take place directly with the armature 44 stationary and the magnets 40 , 41 rotating.
  • the coils are appropriately supplied with current via slip rings (not shown in FIG. 7 ).
  • a similar action can be obtained with the arrangement according to FIG. 6 (reluctance motor).
  • a dynamo iron stator 60 with six distinct stator poles 62 acts on a movable rotor 61 , which can also be made entirely from iron.
  • stator pole 62 there are coils L 1 to L 6 , which can be supplied with d.c. voltage of random polarity.
  • the rotor 61 is firstly non-magnetic, comparable conditions to those of FIG. 4 can be produced, in that the coils L 1 and L 4 are connected in such a way that the indicated south poles of the rotor 61 are obtained.
  • the simultaneously obtained north poles of the rotor with the south poles produced by the same magnetization process then face those stator poles on which the coils L 2 and L 3 or L 5 and L 6 are located.
  • the iron cores of these coils are consequently also premagnetized and consequently influence the current path through these coils, if they are supplied with short test pulses for position measurement purposes.
  • test pulses can naturally also be of different electrical polarity and it is also possible to use coil combinations as premagnetization devices, particularly in series and parallel connection.
  • FIG. 8 A similar configuration to FIG. 7 is shown in FIG. 8 , but in which a fixed armature 44 ′ has twice as many stator poles, which are provided with associated coils L 1 to L 6 .
  • the number of permanent magnetic poles of the rotor 55 is also doubled in FIG. 8 and the said poles of the magnet 50 are interconnected in interruption-free manner. This magnet is held by an iron yoke 51 , which simultaneously leads to an improvement of the magnetic circuit.
  • FIGS. 9 and 10 A comparable arrangement with rotating, but internal magnets (internal rotors) is shown in FIGS. 9 and 10 , which additionally illustrate the magnetic conditions under no-load conditions of the motor.
  • the stator 915 also has six distinct poles with in each case associated coils L 1 to L 6 .
  • the magnets 912 of the rotor are located on an iron yoke 913 and can rotate about the centre 914 .
  • the stator poles with the coils L 1 , L 2 , L 4 and L 5 detect the full magnetic flux of the permanent magnets 912 . They can consequently be easily saturated by additional current application to the said coils than is possible with current application to coil L 3 or L 6 in the represented rotor position. This is illustrated by the section lines 910 and 911 in FIG. 10 . As can be seen the magnetic flux Phi 1 is represented by a larger number of magnetic flux lines, which cross the section line 910 , whereas there is no significant magnetic flux (Phi 2 ) at the section line 911 .
  • FIG. 11 C A corresponding embodiment of a rotor with four magnetic poles 501 to 504 is shown in FIG. 11 C.
  • Poles 501 and 502 are more strongly magnetized than poles 503 and 504 .
  • Those stator poles which face the magnetic poles 501 and 502 consequently receive a higher pre-magnetization and can consequently be brought into magnetic saturation in an easier manner, i.e. with less current.
  • a surface structure of the rotating permanent magnet as shown in FIG. 11B , has a set back surface 116 on one or more poles, e.g. the north pole 117 .
  • FIG. 11 D Similar facts are illustrated by FIG. 11 D.
  • the permanent magnet with the north pole 1123 and south pole 1124 is rotatable about the centre 1127 and faces the stator poles 1126 of the stator 1125 .
  • the air gap 1122 is larger than the air gap 1121 and other, not shown air gaps, as in the aforementioned case, in this way a mechanical/magnetic asymmetry is produced, which allows a clear determination of the rotor position over the full mechanical angle.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)
  • Control Of Electric Motors In General (AREA)
  • Transmission And Conversion Of Sensor Element Output (AREA)
  • Ultra Sonic Daignosis Equipment (AREA)
  • Surgical Instruments (AREA)
  • Vehicle Body Suspensions (AREA)
US08/588,301 1989-06-01 1990-05-31 Apparatus and method for controlling brushless electric motors and position encoders and indicating position thereof Expired - Lifetime USRE39076E1 (en)

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DE3942755 1989-12-23
PCT/DE1990/000411 WO1990015473A1 (de) 1989-06-01 1990-05-31 Motor oder lagemelder
US07/777,283 US5280222A (en) 1989-06-01 1990-05-31 Apparatus and method for controlling brushless electric motors and position encoders and indicating position thereof

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US7973502B2 (en) * 2008-03-21 2011-07-05 Aisin Aw Co., Ltd. Drive unit and manufacturing method thereof
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US20140326579A1 (en) * 2011-11-18 2014-11-06 Novomatic Ag Conveyor device for coins
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ATE121876T1 (de) 1995-05-15
US5280222A (en) 1994-01-18
HK79497A (en) 1997-06-20

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