US2231997A - Frequency discriminator - Google Patents
Frequency discriminator Download PDFInfo
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- US2231997A US2231997A US213589A US21358938A US2231997A US 2231997 A US2231997 A US 2231997A US 213589 A US213589 A US 213589A US 21358938 A US21358938 A US 21358938A US 2231997 A US2231997 A US 2231997A
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- 230000001939 inductive effect Effects 0.000 description 10
- 230000004048 modification Effects 0.000 description 9
- 238000012986 modification Methods 0.000 description 9
- 238000006243 chemical reaction Methods 0.000 description 5
- 238000000034 method Methods 0.000 description 5
- 238000009499 grossing Methods 0.000 description 4
- 230000000903 blocking effect Effects 0.000 description 3
- 238000010586 diagram Methods 0.000 description 3
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- 230000010355 oscillation Effects 0.000 description 3
- 230000003321 amplification Effects 0.000 description 2
- 230000005540 biological transmission Effects 0.000 description 2
- 238000013016 damping Methods 0.000 description 2
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- 230000002452 interceptive effect Effects 0.000 description 1
- 230000010363 phase shift Effects 0.000 description 1
- 229920000136 polysorbate Polymers 0.000 description 1
- 230000001105 regulatory effect Effects 0.000 description 1
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Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
- H03D3/02—Demodulation of angle-, frequency- or phase- modulated oscillations by detecting phase difference between two signals obtained from input signal
- H03D3/22—Demodulation of angle-, frequency- or phase- modulated oscillations by detecting phase difference between two signals obtained from input signal by means of active elements with more than two electrodes to which two signals are applied derived from the signal to be demodulated and having a phase difference related to the frequency deviation, e.g. phase detector
Definitions
- This invention relates to new and' improved means for and a method of receiving a high frequency carrier wave modulated in phase or in frequency by'intelligence to be transmitted.
- intelligence is to include all kinds of signals such as audio, video and other signals, or impulses and the like to be transmitted from one point to another by the aid of a carrier wave.
- a frequency or phase modulated high frequency carrier wave may be rearded as a sine wave of substantially constant amplitude and continuously variable instantaneous frequency.
- demodulating a wave of this 15, type that is in reproducing the low frequency signals or intelligence transmitted by the aid. of such a wave, it is necessary to generate a potential which varies in dependence upon the instantaneous deviation of the frequency of the wave from a predetermined average (carrier) frequency.
- demodulation of a phase or frequency modulated high frequency wave is effected by comparing the phases of the high frequency potentials or ourrents at different points such as in the input and output of a resonant circuit or network.
- a resonant circuit or network may comprise two mutually coupled tuned circuits one of which is excited by the signal waves to be demodulated.
- the arrangement is preferably such that the input and output phases differ exactly by 90 for waves having a frequency corresponding to the resonant (carrier) frequency of the circuit, and deviate in either sense from a 90 mutual phase angle according to the sense and in proportion to the amount of detuning of the impressed frequency relative to the resonant frequency of the circuit.
- the circuit is exactly tuned to the carrier frequency of the waves being received, the input and output phases
- the input and output potentials of varying mutual phase differences are then applied to a device or system adapted to generate an output potential varying in dependence upon the mutual phase angle between the applied potentials, that is according tov the low frequency signal wave, or intelligence impressed upon the carrier wave by frequency modulation.
- Figures 1 and 2 illustrate resonant circuits suited for effecting phase rotation for the purpose of the invention
- Figure 3 shows a modified resonant circuit with a multi-grid vacuum tube as a demodulating device for carrying out the invention
- Figure 4 is a modification of Figure 3 embodying a rectifier bridge as a demodulating device
- Figure 5 is a modification of Figure 4,
- FIG. 6 is a further modification employing vacuum tube rectifiers
- FIGS '7, 8 and 9 illustrate further modifications of the invention embodying a modulator or rectifier bridge circuit as a demodulating de vice
- Figure 10 shows a further resonant circuit for effecting phase rotation suited for the purpose of the invention
- Figure 11 shows a combination of a circuit according to Figure 10 embodying a rectifier or modulator bridge as a demodulating device
- Figure 12 illustrates a further resonant circuit in combination with a vacuum valve demodulator
- Figure 13 is a modification of Figure 12 employing a rectifier bridge as a demodulating arrangement
- FIGS 14 and 15 illustrate in block diagram form improved systems for receiving frequency or phase modulated high frequency waves in accordance with the invention.
- f0 is the average (carrier) frequency of a high frequency wave being received from which the instantaneous frequency ,f deviates by an amount f1.
- This frequency deviation ii in case of frequency modulation is representative of the instantaneous value u of the low frequency or modulating signal wave (intelligence) being transmitted and is related therewith according to a definite, preferably linear law as follows:
- phase shift 12 efiected by the phase rotating resonant circuit may then be expressed as follows:
- band-pass filters suited as phase rotating networks for the purpose of the invention and comprising a pair of tuned or resonant circuits mutually coupled with each other inductively, capacitatively or in any other suitable way.
- the network shown in Figure 1 comprises a first resonant circuit constituted by an induction coil I0 shunted by a condenser H which may be fixed or variable as in the example shown.
- a high frequency input potential e5 which may be a phase or frequency modulated carrier potential.
- the resonant network further comprises a second resonant circuit constituted by an induction coil I2 shunted by a condenser I3 and having its lower side connected directly and having its upper side connected through a suitable coupling reactance such as an induction coil M in the example illustrated with the corresponding sides or terminals of the resonant circuit Ill-l I.
- the output potential es is derived from the resonant circuit l2--l 3 through terminals 0-42.
- a common induction coil I4 in series with the induction coils l0 and I2 of the resonant circuits forming a common coupling link therebetween with or without additional inductive coupling between the coils l0 and I2.
- This circuit may be further modified by omitting the induction coil l4 and providing an inductive coupling only between the coils l0 and I2 thereby obtaining a band-pass filter of known design as shown in the embodiment according to Figure 3.
- the band-pass filters may also be constructed symmetrically as shown in Figures 7 and 8.
- phase of a high frequency output potential (e6) relative to the phase of the impressed input potential (e5) varies in dependence upon the frequency within the range from 0 to If both resonant circuits I0l
- FIG. 3 A, complete demodulating arrangement accordingto the inventioniutiliz'ing an electronic converter or mixer valve is shownin Figure 3, wherein the mixer valve I5 has a cathode IS, a first control grid l'l, a second control grid l8, the latter being shielded from the control grid I! by a positively biased screen I9 in a manner well known, and an anode or plate 20.
- the input potential e5 developed by a frequency modulated wave impressed upon the resonant circuit Ill-ll isapplied to the cathode and control grid l8 of the valve, while the output potential es of variable phase depending on the instantaneous values of the low frequency or demodulating signal and developed across the tuned circuit l2--l3 is applied to the cathode and control grid ll of the valve I5. There is thus produced by the action of the two grids I"!
- valvea current which is a function of the product of the potentials at and 66 impressed upon the grids i8 and II, respectively, which current contains a component 22 varying according to the relative phase difference between the impressed potentials or in turn according to the low frequency or demodulating signal variations
- the latter may be applied to a further utilization or translation device connected to the terminals e,f through a coupling condenser 2
- the sum and difference potentials e5+ee and esee, respectively may be rectified separately by means of suitable rectifiers and the rectified potentials combined differentially in a balanced demodulator arrangement such as of the type shown in Figure 4. In this manner too an output current containing'the low frequency or signal component is obtained.
- a pair of rectifiers l6 and I1 serially connected with their conductive directions opposed to each other across the secondarytuned circuit l2--! 3 through a pair of blocking condensers l4 and 15'.
- the opposite sides of the primary circuit Ill-4 I are connected to the electrical center point of the secondary circuit i2-I3 such as to the mid-point of the induction coil l 2 as shown in the example and to l the common terminalof the rectifiers I6 and H.
- This circuit is also suited for producingan amplitude responsive (AVG) potential T which latter in the example illustrated is derived from the common terminal (g) of "the rectifiers l6 and II on the one hand and from the common terminal of a pair of .additionalhigh ohmic resistances 22' and 23 serially connected across the rectifiers.
- the condensers I4 and I5 serve for blocking the demodulated potential 12 from the resonant circuits of the filter network; In order to afford a constant discharge of these condensers there are provided a pairiof'leak resistances l8 and I 9 connected between the right-hand terminals of the condensers I4.
- the l'eakresistances l8 and I9niay be connected in parallel to the condensers l4? and I 5', respectively, aswill be readily understood.
- the rectifiers l4 and i5 may be of any known type such-as dry rectifiers or. valve rectifiers having a sharply pronounced blocking potential.
- rectifiers with a gradually curved characteristic such asthose with a square law characteristicmay be provided vas is customary in push pull modulators.
- grid leak or audion rectification may be employed as shown in Figure 6.
- the induction coils l and I2 may or maynot be inductively coupled to provide additional coupling between the primary and secondary resonant circuits in addition to the coupling afforded by coil 25 arranged symmetrically betweenthe coils l0. and I 2.
- a capacitative. coupling reactance or condenser may be provided to form a common coupling impedancebetween the primary and secondary resonant circuits. The balanced demodulation of.
- the primary and secondary potentials 65 and ea is carried out bygrid leak detection by means of a twin triode valve 2Q having a pair of control grids*2'l,28 cooperatively arranged with a common cathode 29.
- the potentials developed at the opposite sides of the output resonant circuit are applied to the grids 21 and 28 through coupling condensers l4 and I, the grids 21 and 28 being further provided with grid leak resistances l8 and. 19 in series with a common grid biasing source 24.
- the cathode 29 is further connected to one end (terminal b) of the input resonant circuit.
- the anodes of the two triode units of tube 26 are connected through a pair of resistances 32 and 33 the common terminal of which is connected to'the cathode 29 through ahigh tension source SI and a further. resistance 30.
- the latter serves to supply an amplitude responsive (AVC) potential 1' derived from terminals g and h.
- AVC amplitude responsive
- FIG. 7 A further demodulating arrangement according to the invention is shown in Figure 7.
- the latter also comprises a band-passxfilter constituted by a .pair of resonant circuits l0
- the rectifier bridge comprises four rectifiers 35, 36, 31, 38 connected inseries in like sense as regards their conductive directions to form a closed circuit.
- the opposite sides of the input resonant circuit 10-1 1 are connected to one pair of diagonal points of the rectifier bridge and the opposite ends of the secondary resonant circuit 12-13 of the hand-pass filter are connected to the remaining pair of diagonal points of the bridge, whilethe demodulated or frequency responsive output potential 2) is derived from the electrical centers of the circuits llll I and
- . are preferably dry rectifiers whose impedance may be increased by providing high ohmic resistances in series therewith to reduce the damping imposed upon the associate filter circuits.
- the received instantaneous frequency is higher than the average or carrier frequency in, the potentials es and es will be in counter-phase and the point i'will become negative relative to the .point 7'.
- the average or carrier frequency fo the mutual phase angle between the potentials is 90 whereby the output potential 0' between i and 7' becomes zero.
- a demodulated or output potential '0 varying according to the frequency deviations ii of the received high frequency carrier wave from a predetermined average or carrier frequency (f0).
- the damping of the circuits should be chosen in such a manner that the phase rotation and in turn the output potential 1; is linearly related to the frequencyvariations to be dealt with.
- the circuit according to Figure 7 may be derived'from the principal circuit diagram according to Figure 2 by reducing the inductance l4 to a small value. If a band filter coupling is employed with large coupling impedances according to Figure 1, an arrangement is obtained'as shown in Figure 8 wherein 39 and 40 represent 's'ufiiciently high inductive or capacitative coupling impedances. The latter should be designed in such a manner to prevent a short circuit or circulatory current path for the output or demodulated potential o occurring between terminals e).
- the operation of the rectifier or demodulator bridge 35, 36, 31, 38 in Figure 8 is substantially similar to the embodiment of Figure '7.
- the input impedance of the band-pass filter is determined essentially by the impedance of the primary resonant circuit I0
- a frequency responsive or output potential 1 may be obtained also in the case that the condenser II is omitted and replaced by a sufficiently high inductive or ohmic impedance.
- An arrangement of this type is shown in Figure 9 wherein a tuned primary circuit is dispensed with.
- a sufiiciently high impedance such as an ohmic resistance 41 connected across the input terminals ab and having a center tap leading to the output terminal c.
- This impedance serves merely for restoring the symmetry of the terminal e relative to the terminals ab which symmetry is required for producing the demodulated potential 2; in view of the described function and operation of the rectifier bridge 35, 36, 31, 38.
- FIG. 10 there is shown a different type of resonant network for effecting a phase rotation suited for the purpose of the invention.
- the circuit shown comprises a series combination connected across the input terminals a-b including a parallel tuned circuit constituted. by an induction coil 44 shunted by a condenser 33 and an induction coil 45 in series with the tuned circuit.
- a potential derived from the circulatory current is within the tuned circuit 43, 44 is compared with a potential derived from the total current is flowing into the tuned circuit.
- the currents is and id will be in 1 phase in case of resonance of the circuit 43, 44 with the impressed input signal as (frequency in).
- phase relation between the currents will assume values differing from zero. If the impedance 45 is inductive or capacitative for all the frequencies to be dealt with the potential e7 between the terminals l and n and the potential ea between terminals n and m will be phase shifted by 90 relative to each other in case of resonance while the phase rotation in case of detuning deviates in either sense and in proportion to the amount of detuning from this value.
- a frequency responsive or demodulated output potential may be obtained by differential demodulation of the two potentials in a manner similar to the embodiments described hereinbefore.
- FIG 11 there is shown a complete demodulating circuit arrangement employing a phase rotating resonant network of the type according to Figure 10 and a rectifier or modulator bridge of the type described previously.
- the impedance 45 in the example chosen an inductive impedance is split into two halves 45 arranged serially at opposite sides of the tuned circuits 43-44.
- Both inductances 45' are in symmetrical coupling relation with a further inductance coil 46 serving to supply the potential or to be combined with the potential 66 developed by the circuit 43-44 by means of the rectifier bridge 45, 46, 41, 48 thereby to produce a frequency responsive or demodulated potential 12 at the output terminals e-f connected to the center 'tapcpoints of the indu'ctanc'es 4-4 and 46, respectively.
- a simple series tuned circuit is employed as a-phase' rotating circuit for carrying out the method according to the invention.
- Anarrangement of this type is illustrated in Figure 12.
- a series tuned circuit comprising an induction coil '41 and a condenser 48 is connected across the input terminals aP-b whereby the total potential as impressed upon the circuit may be compared with the potential as developed by the inductance 41 by impressing bothpotentials upon the grids l8 and II, respectively, ofan electronic modulator or mixer valve in substantially a similar manner as described in connectionwith Figure 3.
- the resonant circuit 41, 48 offers a pure ohmic impedance to the impressed potential whereby the potential drop e6 across the inductance 4? is in phase quadrature to thetotal input potential as.
- the relative phase between the potentials'esv and c6 varies in either direction in dependence upon the sense and in proportion to the amount of detuning whereby afrequency responsive or demodulated out-put potential 1.1 is obtained at terminals e-f in amanner well understood from the above;
- the same result is obtained by comparing and combining the potential drop e7 developed across the condenser 48 with the total input potential 5.
- the series tuned circuit is constituted by an induction coil 51! in series with condensers 49 and 5
- an impedance in the example shown a high ohmic resistance connected across the input terminals a-b and having its center tap connected to one of the output terminals (e) the other output terminal (f) being connected to the center tap of the inductance 50 of the series tuned circuit.
- the potentials es and c6 are impressed upon different pairs of diagonal points of the rectifier bridge 35, 36, 31, 38 in substantially the same manner as described hereinbefore to produce a demodulated or frequency responsive output potential 1).
- the frequency demodulating or conversion systems as described may be energized by the high frequency signals received directly or the received signals may be converted to signals of a fixed intermediate frequency by combination with local signalsof auxiliary frequency in which case the conversion or demodulation of the signals of varying frequency is carried out at intermediate frequency whereby the demodulating circuit may be fixedly designed and adjusted for this frequency.
- the amplitude of the received frequencymodulated signals should have a substantially constant value. In practice rapid amplitude variations may occur for one reason or another liable to interfere with the demodulation or conversion of the signals. In most cases a certain amplitude modulation in addition to the frequency modulation of the transmitted carrier wave cannot be avoided. In other cases the amplitude of the amplified high frequency carrier varies in accordance with fluctuations of the operating potentials if the supply potentials'are insufiiciently filtered or smoothened. Moreover, an apparent amplitude modulation-of the received carrier wave may be caused by interference with oscillations received from disturbing transmitters or other interfering I sources.
- the regulation to a constant amplitude may also be effected by means of a control system of the type of the anti-fading or automatic volume control arrangements by regulating the amplification of a control valve in accordance with an amplitude responsive control potential.
- the time constants of this arrangement should be sufficiently low to ensure equalization of the very rapid amplitude fluctuations corresponding to the low frequency modulation.
- the arrangements for receiving frequency modulated carrier waves may furthermore be combined with a system for automatic frequency control (AFC) to the average or carrier frequency of the received oscillations.
- AFC automatic frequency control
- the resonant circuits are controlled by a tuning responsive potential which may be generated in a simple manner by proper smoothing or filtering of the output potential '0.
- FIG. 14 A principal arrangement of the aforementioned type is shown in block diagram form in Figure 14.
- the frequency modulated high fre-- quency potential 61 is amplified in an amplifier 53 to a value e5 sufficient for impression upon the frequency converter or demodulator 54 which may be of any of the previously described types and which serves to produce a frequency responsive or output potential '11 in the manner described. Since this potential 11- is proportional tothe de-' viations of the received frequency from a normal (carrier) frequency, it corresponds to the low frequency or modulating signal except for an additional direct current component.
- This direct current component is caused by small deviations of the tuning of the receiver relative to the average (carrier) frequency of the received signals and may be suppressed by means of a highpass filter 55 to obtain a true demodulated or low frequency output signal 121 from the output of the filter 55.
- this direct current component is applied through a low-pass smoothing filter 56 which latter is impermeable to all speech and modulating signal frequencies, to the variable tuning elements of the selective or resonant circuits of the amplifier 52 and the demodulator 54 (potential '02) in such a manner that the amount of detuning of the latter relative to the average (carrier) frequency ft is automatically maintained at a predetermined small value.
- the degree of amplification or gain of the amplifier 53 may be controlled in accordance with an amplitude responsive control potential 1 obtained for instance in a manner described in connection with Figures 4 to 6 to substantially suppress any disturbing and undesired amplitude modulation of the received signals.
- An additional automatic volume control to compensate slow variations of the carrier amplitude due to fading and similar causes may be employed by impressing a normal AVC potential r1 obtained from the potential 1' by suitable filtering by means of a filter 51.
- the amplitude control potential 1' is obtained by ordinary rectification of the high frequency potential and under circumstances may be derived from the demodulator 54 as shown in Figures 4 to 6.
- FIG 15 there is shown a similar receiving arrangement to Figure 14 as applied to a superheterodyne receiver.
- the frequency modulated input potential e1 is applied to a radio frequency preamplifier 58 and the amplified potential e2 impressed upon a frequency converter or mixer 59 to be combined therein with a locally generated auxiliary potential 63 of different frequency (oscillator 60) resulting in a beat or intermediate frequency potential er in a manner well known.
- the latter is amplified by the aid of an intermedite frequency amplifier 6i and the amplified intermediate frequency potential (25 applied to a demodulator 54 in substantially the same manner as in the case of Figure 14.
- the demodulated or output potential '0 is again impressed upon a high-pass filter 55 to obtain a r final output potential in free from any direct current component while the unfiltered potential 11 is passed through a low-pass filter 56 and the filtered output 02 thereof applied to a frequency controlling element of the local oscillator 60 in such a manner that the intermediate average frequency never deviates to an undesirable degree from the predetermined value.
- the amplitude control potential in the example shown is derived from the demodulator 54 and applied to a gain control element of the intermediate frequency amplifier 6! to suppress undesirable amplitude modulation of the potential as.
- the slow variations of potential 1' derived by means of a filter 62 are applied to a gain control element of the preamplifier 53 to effect a normal automatic volume control (AVC) in a manner similar to the arrangement in Figure 14.
- AVC normal automatic volume control
- the arrangements according to the invention are also suited for the reception of a phase modulated carrier wave as it is well known that a phase modulation is equivalent to a frequency modulation with a linearly distorted low frequency or modulating wave.
- a phase modulation is equivalent to a frequency modulation with a linearly distorted low frequency or modulating wave.
- a network comprising a capacitative and an inductive reactance element in series, said network being resonant to a predetermined frequency, means for impressing a high frequency signal potential upon said network, the relative frequency of said signal potential with respect to the resonant frequency of said network being varied in either direction from said predetermined frequency, an electron discharge tube having a cathode and an anode for producing an electron space current, a pair of control grids located in the path of said space current, a further grid located between said control grids, means for maintaining said further grid at a positive potential with respect to said cathode, an output circuit connected to said anode, means for impressing the total signal potential developed by said network upon one of said control grids, further means for impressing signal potential developed by one of said reactance elements upon said other control grid, and impedance means in said output circuit adapted to develop output potential having an amplitude varying in sign and magnitude proportionately to the frequency departure of said signal potential from the frequency to which said network is reson
- a network comprising a capacitative and an inductive reactance element in series, said network being resonant to a predetermined frequency, means for impressing a high frequency signal potential upon said network, the relative frequency of said signal potential with respect to the resonant frequency of said network being varied in either direction from said predetermined frequency, an electron discharge tube having a cathode and an anode for producing an electron space current, a pair of control grids located in the path of said space current, a further grid located between said control grids, means for maintaining said further grid at a positive potential with respect to said cathode, an output circuit connected to said anode, means for impressing the total signal potential developed by said network upon one of said control grids, further means for impressing signal potential developed by said inductive reactance element upon said other control grid, and impedance means in said output circuit adapted to develop output potential having an amplitude varying in signand magnitude proportionately to the frequency departure of said signal potential from the frequency to Which said network is reson
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Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
CH2231997X | 1937-07-11 |
Publications (1)
Publication Number | Publication Date |
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US2231997A true US2231997A (en) | 1941-02-18 |
Family
ID=4568128
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
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US213589A Expired - Lifetime US2231997A (en) | 1937-07-11 | 1938-06-14 | Frequency discriminator |
Country Status (2)
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US (1) | US2231997A (de) |
CH (1) | CH206044A (de) |
Cited By (11)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US2423594A (en) * | 1942-07-23 | 1947-07-08 | Liebel Flarsheim Co | Resonance indicating and controlling device |
US2426187A (en) * | 1941-12-19 | 1947-08-26 | Standard Telephones Cables Ltd | Pulsed carrier frequency demodulator |
US2525780A (en) * | 1947-01-14 | 1950-10-17 | Charles E Dennis | Electrical frequency discriminator circuit |
US2550524A (en) * | 1945-08-20 | 1951-04-24 | Rca Corp | Balanced microwave detector |
US2667576A (en) * | 1950-05-26 | 1954-01-26 | Int Standard Electric Corp | Frequency discriminator circuit |
US2700763A (en) * | 1949-08-19 | 1955-01-25 | Jr Owen F Foin | Angle detector circuit for radar use |
US2751495A (en) * | 1952-05-07 | 1956-06-19 | W L Maxson Corp | Frequency error sensing means |
US2930892A (en) * | 1954-03-26 | 1960-03-29 | Sperry Rand Corp | Demodulator for a phase or frequency modulated signal |
DE1094312B (de) * | 1956-01-28 | 1960-12-08 | Philips Nv | Frequenzdemodulator |
DE1121135B (de) * | 1959-10-17 | 1962-01-04 | Siemens Ag | Demodulator fuer frequenzmodulierte Hochfrequenzschwingungen |
US3516000A (en) * | 1965-07-29 | 1970-06-02 | Us Army | Regenerative frequency modulation detector with voltage-controlled reactance controlled by output |
Families Citing this family (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
DE741721C (de) * | 1940-10-22 | 1953-05-04 | Siemens & Halske A G Berlin Un | Einrichtung zur Demodulation frequenzmodulierter Schwingungen |
DE938386C (de) * | 1941-04-01 | 1956-01-26 | Heinz Fleck Dr | Gleichrichter mit Gegenkopplung auf einen vorgeschalteten Verstaerker |
DE865008C (de) * | 1949-07-21 | 1953-03-05 | Siemens Ag | Demodulator fuer frequenzmodulierte Hochfrequenzschwingungen |
DE969436C (de) * | 1951-04-26 | 1958-06-04 | Lorenz C Ag | Anordnung zur Demodulation frequenz- oder phasenmodulierter Schwingungen |
DE1194010B (de) * | 1958-10-06 | 1965-06-03 | Tno | Schaltung zur Demodulation von frequenz-modulierten elektrischen Schwingungen |
US3667060A (en) * | 1970-08-26 | 1972-05-30 | Rca Corp | Balanced angle modulation detector |
-
1937
- 1937-07-11 CH CH206044D patent/CH206044A/de unknown
-
1938
- 1938-06-14 US US213589A patent/US2231997A/en not_active Expired - Lifetime
Cited By (11)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US2426187A (en) * | 1941-12-19 | 1947-08-26 | Standard Telephones Cables Ltd | Pulsed carrier frequency demodulator |
US2423594A (en) * | 1942-07-23 | 1947-07-08 | Liebel Flarsheim Co | Resonance indicating and controlling device |
US2550524A (en) * | 1945-08-20 | 1951-04-24 | Rca Corp | Balanced microwave detector |
US2525780A (en) * | 1947-01-14 | 1950-10-17 | Charles E Dennis | Electrical frequency discriminator circuit |
US2700763A (en) * | 1949-08-19 | 1955-01-25 | Jr Owen F Foin | Angle detector circuit for radar use |
US2667576A (en) * | 1950-05-26 | 1954-01-26 | Int Standard Electric Corp | Frequency discriminator circuit |
US2751495A (en) * | 1952-05-07 | 1956-06-19 | W L Maxson Corp | Frequency error sensing means |
US2930892A (en) * | 1954-03-26 | 1960-03-29 | Sperry Rand Corp | Demodulator for a phase or frequency modulated signal |
DE1094312B (de) * | 1956-01-28 | 1960-12-08 | Philips Nv | Frequenzdemodulator |
DE1121135B (de) * | 1959-10-17 | 1962-01-04 | Siemens Ag | Demodulator fuer frequenzmodulierte Hochfrequenzschwingungen |
US3516000A (en) * | 1965-07-29 | 1970-06-02 | Us Army | Regenerative frequency modulation detector with voltage-controlled reactance controlled by output |
Also Published As
Publication number | Publication date |
---|---|
CH206044A (de) | 1939-07-15 |
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