US2296100A

US2296100A - Frequency modulated wave receiver

Info

Publication number: US2296100A
Application number: US353322A
Authority: US
Inventors: Dudley E Foster; John A Rankin
Original assignee: RCA Corp
Current assignee: RCA Corp
Priority date: 1940-02-20
Filing date: 1940-08-20
Publication date: 1942-09-15
Anticipated expiration: 1959-09-15
Also published as: BE474981A; FR950385A

Description

Sp't .15," 1942.

13.5. FOSTER ETAL FREQUEHCY- uommmgn WAVE EcEivER Original Filed Feb. 20, 1940 2 Sheets -Sheet 1 INVENTORS DUDLEY 5. FOSTER AND gun A. Q

ATTORNEY Patented Sept. 15, 1942 FREQUENCY MODULATED WAVE RECEIVER Dudley E. Foster, South Orange, N. J and John A. Rankin, Port Washington, N. Y., assignors to Radio Corporation of America, a corporation of Delaware Original application February 20, 1940, Serial No. 319,830. Divided and this application August 20, 1940, Serial No. 353,322

3 Claims.

Our present invention relates to radio receivers of the ultra-high frequency type, and more particularly to a system adapted to receive frequency modulated waves. This application is a division of our application Serial No. 319,830, filed February 20, 1940,

In co-pending application Serial No. 320,103, filed February 21, 1940, by D. E. Foster there is disclosed a frequency modulated Wave (hereinafter referred to briefly as FM) receiver adapted to receive signal waves located in the ultra-high frequency band of 42.6 to 43.4 megacycles (me). The receiver is of the superheterodyne type, and utilizes push-button tuning for a plurality of channels within the aforesaid operating band, or range. Automatic frequency control (AFC) is there employed to minimize the effect of frequency drift.

Another important object of our invention is to provide, in an ultra-high frequency receiver, an AFC circuit utilizing an improved frequency control tube circuit which is adapted to provide a simulated inductive effect across a local oscillator tank circuit without damping effect on the latter.

Other objects of our invention are to improve generally the simplicity and efiiciency of ultrahigh frequency signal receivers, and more particularly to provide an FM receiver capable of being economically manufactured and easily as sembled.

The novel features which we believe to be characteristic of our invention are set forth in particularity in the appended claims; the invention itself, however, as to both its organization and method of operation will best be understood by reference to the following description taken in connection with the drawings in which we have indicated diagrammatically a circuit organization whereby our invention may be carried into effect.

In the drawings:

Fig. 1 shows a circuit diagram of an FM receiver embodying our invention,

Fig. 2 illustrates the FM detector characteristic,

Fig. 3 graphically shows the frequency control characteristic of the AFC tube.

Referring now to the accompanying drawings, and specifically to Fig. 1, the networks of the FM receiver up to the limiter tube may be similar to those described and claimed in the aforesaid Foster application. Hence, such networks will be only generally described in this application. The tube I, of the 6SA7 type, is a pentagrid con-' Yerter tube provided with a cathode 2, oscillator grid electrode 3, signal input electrode 4, a grounded suppressor grid 5, an output electrode 6 and positive screen grids l surrounding the signal grid 4. a

The collected signal energy is impressed upon a radio frequency amplifier which is tuned to the mid-band frequency. Since the complete frequency band is 42.6 to 43.4 mc., then the midband frequency of amplifier 8 is 43 me. The signal collector can be a dipole, and at the output terminals of amplifier 8 there will be present in amplified form the various modulated carrier Waves in the desired frequency band. By having the input circuit of the amplifier 8 of the bandpass type and covering the entire spectrum of frequencies desired, there will be developed across the signal input circuit coil 9 voltages of the various frequencies in the desired signal frequency band. Channel selection is accomplished by providing in shunt with coil 9 five independent channel selection coils. Each of the coils I0 is shown with an arrow passing therethrough, and it is to be understood that the arrow denotes the fact that the inductance value of each, of the coils is adjusted by the utilization of an adjustable brass cylinder, as disclosed in the aforesaid Foster application.

The grounded terminals of each of coils I0 is connected to the low potential end of coil 9 througha direct current blocking condenser II while the opposite ends of each of coils ID are connected to the adjustable contact arms of each of a plurality of push-button switches 12. The contact points of each of switches I2 are connected to the signal grid lead, and the direct current blocking condenser I3 is inserted between the high potential terminal of coil 9 and the contact points of switches l2. It will be understood that the value of each of coils 10 is adjusted so that upon closing of a particular one of pushbuttons I2 there will be selected from the band of impressed signal energy solely signal voltage of the channel frequency representative of the particular push-button depressed.

The tunable oscillator tank circuit is constructed in a manner similar to the signal input circuit just described. That is to say, the tank circuit coil M has connected in shunt therewith a plurality of independent channel selection coils I5, and the latter coils are constructed in the same manner as explained in connection with coils ID. The push-button switches it are connected so that closing of a particular push-button connects the corresponding coil [5 in shunt with the tank coil M; The oscillator tank circuit frequency range is higher than the signal input frequency range, and is constantly different therefrom by the value of the operating I. R, which is chosen to be 2.2 mc. The dotted line ll is to be understood as symbolically representing the mechanical couplings between each corresponding pair of switches l2 and 96. That is to say, each of switches !2 has a corresponding tank circuit switch it, and they are closed in pairs to efiect channel selection. The channels themselves are arranged at intervals of 0.2 mc.

The oscillator grid 3 is connected to the higlr potential end of coil I4 through a condenser l8, while the resistor l3 connects the grid end of condenser [8 to ground. The low potential end of coil [4 "is established at ground potential, and the cathode 2 of tube l is connected to an intermediate tap 23 on the oscillator tank coil [4. There is developed across the I. F.-tuned output circuit 2! the FM wave energy which has been selected at the input circuit, but the midchannel frequency has the I. F. value of 2.2 me. It is not believed necessary to describe the man-v ner in which I. F. voltages are developed by the converter tube, since those skilled in the art are fully aware of the manner of functioning of an electron-coupled converter of the type shown herein. The tube l and its associated circuits provide the Well known form of electroncoupled combined local oscillator and first detector network.

The I. F, signal energy is transmitted through the bandpass network which comprises circuits 2| and 22 each fixedly tuned to the I. F. of 2.2 mc., it being pointed out that a resistor 23 is shunted across secondary circuit 22 in order to provide a wide band transmission network which is readily able to handle a wide band of I. F. energy. While the I. F. value may be generally chosen in the range of 2 to 4 mc., it is preferred to select an I. F. value of 2.2 mc., the band width being of the order of 200 kilocycles (kc.). In general, the coupling between each of the I. F. transformer circuits is so adjusted that the desired band Width is attained with the secondary damping resistor 23. The numeral 24 denotes an I. F. amplifier stage, and the tube employed may be of the same type as employed in stage 8, and an additional I. F. amplifier stage may follow tube 24 if desired. Each of these amplifier tubes may be, for example of the SAC? type.

The amplified I. F. energy is then transmitted through the I. F. transformer 25 which feeds the input electrodes of the limiter tube. The transformer 25 has its primary and secondary circuits each fixedly resonated to the operating I. F. value, and the secondary damping resistor 26 performs a function similar to that described in connection with resistor 23. The numeral 2'! denotes the SAC? type limiter tube. By virtue of its associated circuits this tube functions as a combined amplitude modulation (AM) limiter and automatic volume control (AVC) stage. The function of the limiter tube is essentially substantially to eliminate any amplitude modulation eifect in the output thereof. That is to say, since the energy applied to the following FM detector should be a pure FM wave, it is necessary to subject the output of the I. F. amplifier to a limiting device in order to suppress any amplitude modulation which may have appeared on the envelope of the FM carrier wave. In brief, then, a limiter stage is a device whose output is constant for wide variations in the amplitude of the applied signal.

However, this is necessary only because of the inherent limitations of the detector, and thus the limiter might properly be considered as an integral part of the detection circuit.

Considering, first the electrical connections to the electrodes of the limiter tube, the cathode is established at ground potential, while the low potential end of the input circuit 25 is connected to the grounded cathode through a resistor 3| shunted by condenser 32. The signal input electrode 28 of the limiter tube is connected to the high potential end of the input circuit. Across resistor 3| is developed a direct current voltage which is derived from the grid current flow of the limiter tube. The direct current voltage is utilized as a bias for grid 28, and for a function to be later described. The voltage is, furthermore, employed for securing AVC action, and for this purpose there is shunted across resistor 3| a second resistor 33, an intermediate point thereof being connected to the lead 34 designated as AVC.

Those skilled in the art are fully aware of the manner of utilizing an AVC circuit, and it is only necessary to point out that lead 34 applies the AVG bias to the signal grids of each of amplifiers 8 and Y24 through filter resistors 35. The applied AVC bias acts to increase the efiective signal grid bias of each of the controlled amplifiers so as to reduce the gain of each of the controlled amplifiers. In this Way it is possible to maintain the selected mid-channel frequency amplitude at the input circuit 25 at a desired amplitude. An mentioned in the aforesaid Foster application each of the controlled amplifiers preferably includes a degenerating resistor in its cathode circuit in order to prevent detuning effects on the controlled amplifier input circuits due to the variable signal grid biasing action of the AVG circuit.

The screen grid 29 of tube 2'! is connected to the source of positive voltage which also supplies the plate 30 through a resistor 36, the screen grid, furthermore, being connected to ground through a second resistor 31, the screen grid end of the resistor 31 being by-passed by a condenser 38,. The plate 30 is comiected to the positive potential source through a resistor 39 whose plate end is, also, by-passed to ground through a condenser 40. Across the I. F-tuned output circuit 4| there is developed I. F. energy practically free of amplitude modulation on its modulation envelope. A damping resistor may be shunted across the circuit 4| in order to impart the desired wide band characteristic thereto. The AM limiter operates with grid current-developed bias and with reduced screen potential to secure limiting action on both the positive and negative halves of the wave envelope. The limiting on the positive half of the wave envelope is due to the developed grid bias fluctuations, and on the negative half of the wave the limiting action is secured by virtue of plate current cut-off.

- Resistor 3i and condenser 32 determine the time constant on discharge. Resistor 28 is the secondary circuit damping resistor. Resistors 31 and 38 comprise the bleeder for fixing the screen potential, and hence the plate current characteristic. The time constant for charging condenser 32 is determined by the value of the condenser and the value of the charging resistance. The equivalent resistance on the charge portion of the cycle is due to resistor 26, the I. F. transformer impedance, and the limiter tube. Further discussion of the limiter operation will be found in the aforesaid parent application.

The purely FM signal energy at circuit 4! is applied to the input circuits of the double diode tube 50, which may be of the 6H6 type. The tube has two pairs of independent diode electrodes therein. Anode 5| and cathode 52 comprise one diode, while anode 53 and cathode 54 provide the second diode. The

cathodes

52 and 54 are con nected by a pair of resistors 58' and 58. The junction of the resistor 58 and cathode 54 is at ground potential, and independent I. F. carrier by-

pass condensers

55 and 56 shunt each of the resistors. The diode 5I-52 has an input circuit comprising coil 59 and shunt condenser 60, and the input circuit is fixedly tuned to a frequency of the order of 2.1 me. The auxiliary coil 6! is arranged in series with the coil 59 and condenser 60, and coil 59 is magnetically coupled to circuit 4|. The lead 62 connects the resistor section in series with the space current path of diode 51-52.

The input circuit of diode 53-54 comprises coil'63, auxiliary coil 65 and condenser 64. Lead 62 connects the junction of resistors 58 and 58 to the junction of condensers 60 and 62. The input coil 63 is magnetically coupled to circuit 4|. Each of circuits 59-60 and 63-64 are tuned by equal frequency amounts to opposite sides of the I. F. value. For example, the circuit 59-60 may be fixedly resonated to 2.1 mc., while the other input circuit is tuned to 2.3 mc. Hence, there will be developed across each of resistors 58 and 58 rectified signal voltage. The rectifiers being in opposed relation, there will be tapped off at the cathode end of resistor 58 the difference of the voltages across the two load resistors.

Switch 10, when closed on the lower contact provides a path for tapping off the audio voltage developed across the resistive load of the rectifiers. The resistor-condenser path I 1-12, shunt ed across the detector output, provides a deemphasizing network for the high frequency components of the modulation voltage output. Since FM transmitters usually employ some high audio frequency component emphasis, the path ll-12 will provide a compensation therefor; this action arises by virtue of the attenuation of the high audio components. The audio voltage across condenser 12 is then utilized by any desired form of audio utilization network, and the usual tonec-ompensated audio volume control device 13 may be connected across condenser 12.

Since the detector stage comprises a pair of oppositely mistuned, relative to I. F. value, rectifiers having output load resistors connected in phase opposition, or back-to-back, the detector characteristic will be a slope between the offresonance frequencies of the rectifiers. The output of the detector will depend on frequency variation, and not on amplitude variation. In an FM wave the instantaneous frequency is varied at an audio rate, so that if such a wave is ap-- plied to the detector shown the output will vary at the audio rate, and, hence, reconstitute the original modulation imposed on the transmitter. The magnitude of the audio frequency output is proportional to the slope of the detector characteristic and to the amount of frequency deviation. The slope of the characteristic is dependent on circuit constants. Hence, in a given detector, the output is proportional to frequency deviation; that is, to the amplitude of the audio frequency wave modulating the transmitter.

Despite the fact that off-resonance circuits are used, alignment of the two detector input circuits may be performed at the center I. F. value. The auxiliary coils 6i and 65 function as the aligning coils. During alignment connection is made to the upper contact by switch arm 66; that is, coil 6| is short circuited. In this position of switch 66 the circuits 4|, 59-68, 63-64 are each tuned to the I. F. value of 2.2 me. The switch arm 66 is then shifted to the lower contact to short circuit coil 65. This results in a decrease of the resonant frequency of circuit 59-68, and a concurrent increase of the resonant frequency of circuit 63-64 by a like frequency amount. The coils 59 and 63 may have the same magnitudes, but coils BI and 65 are of different values.

In Fig. 2 there is shown the FM detector characteristic. The separation between the peaks of the curve depends upon the ratio of the inductance of coils 6| and 65 to that of coils 59 and 63. Inductances 6| and 65 may conveniently be outside the I. F. shield can so that its value may be readily varied to obtain the desired separation of the resonance peaks during the aligning process. While a single coil could be used in place of the two auxiliary coils 6l-65 by connecting the single coil in series between the coils 59 and 63, yet with two coils the desirable symmetrical characteristic of Fig. 2 is obtained. This curve shows clearly how the frequency shifting carrier is converted into amplitude modulation, and rectified. If an FM Wave is one with unvarying amplitude whose frequency is cyclically altered above and below its mean unmodulated value (I. F.), then Fig. 2 shows how each rectifier supplies the variable unidirectional voltage from its load resistor. The algebraic sum of the two voltages, varying at an audio rate, is supplied to the audio circuit when switch Ill is closed on the lower contact. If, however, switch 10 is closed on the upper contact, the grid circuit of the limiter tube functions as a diode rectifier, and provides audio voltage over lead to the audio circuit. The latter is the A. M. reception position of switch 10. The detector now acts solely as an AFC discriminator in the usual manner.

As explained in the aforesaid Foster application, it is highly necessary to apply AFC to the local oscillator because of the use of push-button tuning, and the fact that at the ultra-high frequencies employed slight departures from correct oscillator frequencies will result in considerable mistuning and consequent severe distortion. The tolerance in frequency drift is materially less in these high frequencies. Further, since it is essential to have the applied I. F. energy have a mid-channel frequency locatedat the center between the two peaks of the characteristic of Fig. 2, AFC becomes practically essential for ease of tuning operation.

The FM detector obviously is inherently an AFC discriminator; Fig. 2 illustrates this fact. Those skilled in the art are fully aware of the manner of employing AFC in a superheterodyne receiver. Reference is made to application Serial No. 130,630 filed March 13, 1937, granted December 23, 1941, as U. S. Patent No. 2,267,453 to D. E. Foster to show such an AFC system. In the present application, a highly desirable and improved form of frequency control tube circuit is utilized across the oscillator tank coil M. The control tube 81 is a high transconductance tube of the 6AC7 type, and its plate 82 is connected to the high potential side of coil I4 by the direct current blocking condenser 83. The plate is connected to a source of positive potential through a choke coil 84. The cathode of tube 8| is connected to ground through self-bias resistor 85 shunted by bypass condenser 86. There is applied to grid 8'! an alternating voltage which is in quadrature with the plate alternating poten tial. The grid 87 is connected to point 20 on coil M for this purpose through a path including direct current blocking condenser 83 and resistor 89.

The oscillator tank circuit current flowing through the path including resistor 89, condenser 88 and the grid to cathode capacity 9|] (shown dotted) develops across capacity 90 the quadrature voltage. As is known, and as explained in the aforesaid Foster AFC patent, between the plate 82 and ground is simulated an inductive effect which appears in shunt to coil M. The dotted coil 9! is the simulated inductive effect due to tube 8 I. The magnitude of the inductance 9| depends on the gain of tube 8|. Hence, the signal grid 81 is connected to ground through a source of frequency-dependent direct current voltage. The latter source is the direct current voltage developed across resistive load 5858. The audio pulsations of the voltage across the latter are filtered out by filter resistor 92 and condenser 93. The lead 94, designated as the AFC lead, connects resistor 92 to grid 81 through filter resistors 95 and 96, the junction of the latter being by-passed to ground by condenser 91.

The characteristic of the frequency control tube circuit is shown in Fig. 3. There are plotted Control tube bias volts against Oscillator frequency change (KC). It wil be noted that a satisfactorily wide range of correction can be secured with the circuit. This characteristic together with that of the detector (Fig. 2) determine the mistuning correction of the AFC circuit for signals above the limiter stage threshold. This control action is seen to be approximately 10 to 1 so that mistuning of the oscillator tank circuit of 100 kc, would cause a mistuning from the I. F. value of only 10 kc. It will be clear to those skilled in the art that if the I. F. energy has a center frequency which departs from the assigned 2.2 mc. value, that rectifier whose input circuit is closer to the shifted value will dominate in direct current voltage production. The resulting bias applied over AFC lead 94 will vary the gain of tube 8| in that sense, and to an extent, such that the effect 9| will vary to cause the oscillation frequency to be shifted so as to compensate for the I. F. center frequency shift.

It is pointed out that the inductance 9| increase in value as the gain of the control tube is decreased by AFC bias, and that an increase in shunt inductance results in a decreased tank frequency.

With respect to the simulated inductive effect 9|, it can be shown that its value is:

In the aforesaid formula, C is the capacity 90; R is the series resistor; Gm is the mutual conductance of the control tube; and K is the portion of the tank voltage appearing between tap and ground. Since K is less than unity this would appear to give a higher L; therefore, less shift is produced than if the voltage across the entire coil is used. However, one may reduce R in proportion to K and obtain the same voltage across C as if the entire coil voltage were used. Suppose K were 0.3 then if R is reduced by some factor from the value that would be used across the entire coil, the same radio frequency voltage is applied to the control tube grid. But if K=0.3 the inductance across which it is tapped becomes 0.09 of total inductance (inductance being approximately proportional to the square of the number of turns) so that the shunting effect of R and of input resistance of tube is decreased by a factor of 0.3 thereby giving more nearly pure quadrature voltage, and better control action.

The grid 81 of tube 8| is tapped down to 20 for the following reason. All amplifier tubes that might be used as tube 8| have high input conductance (low resistance) at the ultra-high frequencies hence they seriously load any tuned circuit if connected across the entire tuned circuit. By tapping the grid down this loading effect is reduced. The grid circuit of grid 81 is tapped down to the oscillator cathode tap. However, this is not necessarily the only place that it could be tapped. It could be tapped either above or below the cathode tap, and, further, the oscillator circuit might not use a hot cathode but might obtain feedback coupling by mutual inductance coupling. In the case of an oscillator using mutual inductance coupling the advantage of tapping the grid circuit of grid 81 across part of the coil would still prevail at ultra-high frequencies.

The advantages of the present frequency control tube circuit will be seen to involve the following. Firstly, the inherent capacity can be employed as a quadrature condenser. Second, the grid tap 20 on coil l4 prevents the low input resistance of tube 8| from damping the tank circuit. Thirdly, the high transconductance tube 8| gives sufficient control with the small inductances l5 required for the ultra-high frequency range.

The receiver shown herein will readily receive FM transmissions using deviations of about 75 kc. and transmitting modulation frequencies up to 15,000 cycles. If the deviation is less a proportionate reduction in I. F. band width should be made, and if less than 15,000 cycle modulation is employed then the audio pass band should be likewise reduced to maintain noise reducing capabilities. The receiver will deliver 0.5 watt output with 12 kc. deviation for any input over 3 microvolts, and it will be sensitive. The noise threshold for fluctuation noise is at about 3 microvolts so that noise reduction is obtained above that input. Since the limiter operates fully at 6 microvolts above the latter input the receiver will be completely quiet as far as fluctuation noise is concerned.

It is pointed out that the present system is adapted for receiving phase modulated carrier waves. Since the electrical relations between phase and frequency modulation are similar, it is obvious that a common receiving system can be used for receiving both types of waves. The term timing modulation is used herein to describe a common characteristic of phase and frequency modulated waves, since in both cases the modulation varies the timing of the carrier wave frequency with respect to a hypothetical point.

Since in phase modulation the frequency deviation is proportional to the modulating frequency, uniform output for all audio frequencies when the receiver herein described is supplied with phase modulated waves may be secured by appropriate values of resistor H and capacitor I2. For example, if resistor H be made high in comparison with the reactance of 12 for even low audio frequencies (say down to 30 cycles) the voltage across 12 will be inversely proportional to audio frequency, and uniform frequency response will be secured from phase-modulated waves.

The following circuit constants are given, it being clearly understood however that they are in no way restrictive, but are merely illustrative:

Rs9=2000 ohms R9s=50,000 ohms R95=50,000 ohms Ra=500 ohms 12:11:20,000 ohms R92=1 megohm R5a'=250,000 ohms R5s=250,000 ohms Ra9=100,000 ohms Rzs=10,000 ohms Rav=9,000 ohms R33=1 megohm+300,000 ohms R31=50,000 ohms R2s=25,000 ohms La4=40 microhenries L41=56 microhenries L59=56 microhenries L63=56 microhenries Le1=5.6 microhenries Ls5=5.0 microhenries C32=5O micromicrofarads (mmf.) Caa=0.1 microfarad (mf.) C4o=5000 mmf.

C55=50 mmf.

C5s=50 mmf.

C72=1000 mmf.

While we have indicated and described a system for carrying our invention into effect, it will be apparent to one skilled in the art that our invention is by no means limited to the particular organization shown and described, but that many modifications may be made without departing from the scope of our invention, as set forth in the appended claims.

What we claim is:

1. In a radio receiver of the superheterodyne type for amplitude, phase, or frequency modu lated Waves a local oscillator circuit comprising an inductance and capacitance, an automatic frequency control tube comprising at least a grid electrode, a cathode and an anode, a quadrature circuit comprising a resistor and capacitor in series, said quadrature circuit being connected across a portion of said oscillator inductance to prevent the input conductance of said control tube from loading said oscillator circuit, the grid electrode of said frequency control tube having radio frequency energy supplied to the grid electrode thereof from the junction of said resistor and condenser of said quadrature circuit, means for simultaneously applying direct current control potential to said grid electrode, and the anode of said control tube being connected across a greater portion of said oscillator inductance than is said quadrature circuit.

2. In a radio receiver of the superheterodyne type for amplitude, phase, or frequency modulated waves a local oscillator circuit comprising an inductance and capacitance, an automatic frequency control tube comprising at least a grid electrode, a cathode and an anode, a quadrature circuit comprising a resistor and capacitor in series, said quadrature circuit being connected across a portion ofsaid oscillator inductance to prevent the input conductance of said control tube from loading said oscillator circuit, the grid electrode of said frequency control tube having radio frequency energy supplied to the grid electrode thereof from the junction of said resistor and condenser of said quadrature circuit, means for simultaneously applying direct current control potential to said grid electrode, and the anode of said control tube being connected across the entire oscillator inductance.

3. In a radio receiver of the superheterodyne type for amplitude, phase, or frequency modulated waves a local oscillator circuit comprising an inductance and capacitance, an automatic frequency control tube comprising at least a grid electrode, a cathode and an anode, a quadrature circuit comprising a resistor and capacitor in series, said quadrature circuit being connected across a portion of said oscillator inductance to prevent the input conductance of said control tube from loading said oscillator circuit, the grid electrode of said frequency control tube having radio frequency energy supplied to the grid electrode thereof from the junction of said resistor and condenser of said quadrature circuit, means for simultaneously applying direct current control potential to said grid electrode, the anode of said control tube connected across a greater portion of said oscillator inductance than is said quadrature circuit, and said quadrature capacitor being provided by the inherent capacity between the grid and cathode of the control tube.

DUDLEY E. FOSTER. JOHN A. RANKIN.