TWI472140B - DC-DC converter - Google Patents

Description

DC-DC converter

The present invention relates to a power supply device that supplies power of a direct current power source to a load.

In recent years, as awareness of global environmental protection has increased, systems for developing DC power supplies such as batteries, solar cells, and fuel cells have been continuously developed. In these systems, a DC-DC converter that supplies power from a DC power source to a load or other DC power source is required.

Examples of the DC-DC converter include a DC-DC converter in which a voltage type switching circuit and a current type switching circuit having a smoothing inductance are connected by a transformer.

Further, Non-Patent Document 1 discloses a DC-DC converter technology in which a voltage type full bridge circuit and a current type full bridge circuit are connected by a transformer, whereby electric power can be supplied in both directions.

[Previous Technical Literature] [Non-patent literature]

[Non-Patent Document 1] K. Wang, C. Y. Lin, L. Zhu, D. Qu, F. C. Lee and J. S. Lai: "Bi-directional DC to DC Converters for Fuel Cell Systems”, IEEE power electronics in transportation, pp. 47-51, Dearborn, MI (1998)

However, a DC-DC converter in which a voltage type switching circuit and a current type switching circuit having a smoothing inductance are connected by a transformer, including the technique disclosed in Non-Patent Document 1, has the following problem, that is, when When the voltage type switching circuit side supplies power to the current type switching circuit side, if the input voltage is reduced to a certain level or less, the required output power cannot be obtained.

In addition, another problem is that in order to obtain the required output power under the condition that the input voltage is low, it is necessary to increase the turns ratio of the transformer (the turns ratio is defined as the number of turns of the current type switching element circuit side divided by the voltage type switch) The value of the number of turns on the circuit side).

In addition, if the turns ratio is increased, even if the input voltage is low, a high voltage can be obtained in the winding of the current type switching circuit, and a large output power can be easily obtained. However, if the turns ratio is increased like this, the input voltage is obtained. When it goes high, the winding on the side of the current mode switch circuit generates a higher voltage. Therefore, there is a problem that a switching element requiring a high withstand voltage is required.

Further, another problem is that if the withstand voltage of the switching element becomes high, the loss also becomes large, so that the efficiency of the DC-DC converter is lowered.

As described above, the conventional DC-DC converter including the technique disclosed in Non-Patent Document 1 has the following problems, that is, when the voltage type switch is used. When the circuit side supplies power to the current type switching circuit side, when the input voltage range is to be widened, the current type switching circuit must have a high withstand voltage switching element, which hinders the efficiency of the DC-DC converter.

Accordingly, the present invention solves such a problem and aims to provide an efficient DC-DC converter.

In order to solve the aforementioned problems, the object of the present invention is achieved as follows.

That is, the DC-DC converter according to the present invention includes: a transformer including a primary winding and a secondary winding, wherein the primary winding is magnetically coupled to the secondary winding; and the first switching circuit has a plurality of switches In the device, a first smoothing capacitor is connected between the DC terminals, and the primary winding is connected between the AC terminals to convert DC power into AC power; the second switching circuit has a plurality of switching elements, and the second terminal is connected between the AC terminals. a winding, wherein a second smoothing capacitor and a smoothing inductor are connected in series between the DC terminals to convert AC power into DC power; a resonant inductor is connected in series with the primary winding and/or the secondary winding; and an auxiliary resonant circuit is connected to the foregoing Between the AC terminals of the second switching circuit, or between the DC terminals of the second switching circuit, or between the AC terminal and the DC terminal of the second switching circuit, the auxiliary resonant capacitor and the auxiliary resonant switching element are connected in series. a series connection body; a first DC power source connected in parallel with the first smoothing capacitor, and a second smoothing capacitor The DC load connected in parallel supplies power, and when the auxiliary resonance switching element is OFF In the case of the state, charging of the aforementioned auxiliary resonance capacitor is prevented.

Further, other means will be described in the embodiments.

According to the present invention, it is possible to provide a highly efficient DC-DC converter.

1‧‧‧Switch circuit (1st switch circuit)

2, 21, 22‧‧‧ switch circuit (2nd switch circuit)

3‧‧‧Auxiliary resonant circuit

31‧‧‧Auxiliary resonant circuit (first auxiliary resonant circuit)

32‧‧‧Auxiliary resonance circuit (2nd auxiliary resonance circuit)

4, 41‧‧‧ voltage clamp circuit

5‧‧‧Control means

6‧‧‧DC load

10‧‧‧ Current Sensor

11, 12‧‧‧ voltage sensor

101, 102, 103‧‧‧DC-DC converter

C1‧‧‧Smoothing Capacitor (1st smoothing capacitor)

C2‧‧‧Smoothing capacitor (2nd smoothing capacitor)

Cb, Cb1, Cb2‧‧‧ auxiliary resonant capacitor

Cc, Cc1‧‧‧ clamp capacitor

Cr‧‧‧Resonance Capacitor

DH1~DH4, DS0~DS4, DS11, DS12, DS31, DS41, DSb, DSb1, DSb2‧‧‧ diode

H1‧‧‧Switching element (first switching element)

H2‧‧‧Switching element (2nd switching element)

H3‧‧‧Switching element (3rd switching element)

H4‧‧‧Switching element (4th switching element)

S0, S31, S41‧‧‧ switching elements (clamping switching elements)

S1‧‧‧ switching element (5th switching element)

S2, S21‧‧‧ switching element (6th switching element)

S3‧‧‧Switching element (7th switching element)

S4, S11‧‧‧ switching elements (8th switching element)

Sb, Sb1, Sb2‧‧‧ switching elements (auxiliary resonant switching elements)

L‧‧‧Smooth inductance

L1‧‧‧Smooth inductance (1st smoothing inductance)

L2‧‧‧Smooth inductance (2nd smoothing inductance)

Lr‧‧‧Resonance Inductance

N1‧‧‧ winding (primary winding)

N2, N23‧‧‧ winding (secondary winding)

N21‧‧‧ winding (first secondary winding)

N22‧‧‧ winding (second secondary winding)

Nd1~Nd4, Nd31, Nd32, Nd41, Nd42‧‧‧ nodes

T, T1, T2‧‧‧ transformer

V1‧‧‧DC power supply (1st DC power supply)

V2‧‧‧DC power supply (2nd DC power supply)

Fig. 1 is a block diagram showing a circuit configuration of a DC-DC converter according to a first embodiment of the present invention, and a connection between a DC-DC converter and a DC power supply V1, and a connection between a DC-DC converter and a DC power supply V2 and a DC load. schematic diagram.

2A is a schematic diagram showing the ON/OFF state of each switching element in the mode a of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

2B is a schematic diagram showing the ON/OFF state of each switching element in the mode b of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

2C is a schematic diagram showing the ON/OFF state of each switching element in the mode c of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

[Fig. 2D] In the mode d of the DC-DC converter according to the first embodiment of the present invention, the ON/OFF states of the respective switching elements are distributed in each circuit. A schematic diagram of the current direction.

2E is a schematic diagram showing the ON/OFF state of each switching element in the mode e of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

FIG. 2F is a schematic diagram showing the ON/OFF state of each switching element and the current direction flowing through each circuit in the mode f of the DC-DC converter according to the first embodiment of the present invention.

Fig. 3 is a waveform diagram showing the operation of the operation 1 of the DC-DC converter according to the first embodiment of the present invention.

4A is a schematic diagram showing the ON/OFF state of each switching element in the mode A of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4B is a schematic diagram showing the ON/OFF state of each switching element in the mode B of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4C is a schematic diagram showing the ON/OFF state of each switching element in the mode C of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4D is a schematic diagram showing the ON/OFF state of each switching element in the mode D of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4E is a schematic diagram showing the ON/OFF state of each switching element and the current direction flowing through each circuit in the mode E of the DC-DC converter according to the first embodiment of the present invention.

4F is a schematic diagram showing the ON/OFF state of each switching element and the current direction flowing through each circuit in the mode F of the DC-DC converter according to the first embodiment of the present invention.

[Fig. 4G] Fig. 4G is a schematic diagram showing the ON/OFF state of each switching element in the mode G of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4H is a schematic diagram showing the ON/OFF state of each switching element in the mode H of the DC-DC converter according to the first embodiment of the present invention, and the current direction flowing through each circuit.

Fig. 5 is a waveform diagram showing the operation of the operation 2 of the DC-DC converter according to the first embodiment of the present invention.

Fig. 6 is a block diagram showing a circuit configuration of a DC-DC converter according to a second embodiment of the present invention, and a connection between a DC-DC converter and a DC power supply V1, and a connection between a DC-DC converter and a DC power supply V2 and a DC load. schematic diagram.

Fig. 7 is a block diagram showing a circuit configuration of a DC-DC converter according to a third embodiment of the present invention, and a connection between a DC-DC converter and a DC power supply V1, and a connection between a DC-DC converter and a DC power supply V2 and a DC load. schematic diagram.

The present embodiment will be described below with reference to the drawings.

(First embodiment)

A first embodiment of the DC-DC converter of the present invention will be described.

1 is a circuit configuration diagram of a DC-DC converter 101 according to a first embodiment of the present invention, and a connection between a DC-DC converter 101 and a DC power supply V1 (first DC power supply), and a DC-DC converter 101 and A schematic diagram of the connection between the DC power source V2 (the second DC power source) and the DC load 6 is shown.

In FIG. 1, the DC-DC converter 101 is connected between a DC power source V1 and a DC power source V2 to which a DC load 6 is connected, and converts a DC voltage of DC power input from a DC power source V1 into a DC voltage of different DC voltages. The power source V2 and the DC load 6 supply power.

In addition, DC power is supplied to the DC power source V1 from the DC power source V2 as necessary.

The DC-DC converter 101 includes a switch circuit 1 (first switch circuit), a switch circuit 2 (second switch circuit), an auxiliary resonance circuit 3, a voltage clamp circuit 4, and a switching element for controlling the circuits ( Control means 5 for the ON/OFF state of H1 to H4, S1 to S4, Sb, and S0).

Further, the DC-DC converter 101 includes a smoothing capacitor C1 (first smoothing capacitor), a smoothing inductor L, a smoothing capacitor C2 (second smoothing capacitor), a resonant capacitor Cr and a resonant inductor Lr, and a winding N1 (primary winding). Transformer T of winding N2 (secondary winding), voltage sensors 11, 12, and current sensor 10.

The switching circuit 1 is a switch composed of an N-type MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) The elements H1 to H4 are fully bridged.

a first switching branch formed by connecting the switching element H1 (first switching element) and H2 (second switching element) in series by the node Nd1, and a switching element H3 (third switching element) connected in series by the node Nd2, The second switching branch formed by the switching element H4 (fourth switching element) is connected in parallel, and between the two ends of the first switching branch is used as a DC terminal, and between the node Nd1 and the node Nd2 as an alternating current terminal.

The control terminals (gate input terminals) of the switching elements H1 to H4 composed of MOSFETs are connected to the control means 5 and controlled.

In the switching circuit 1, the switching elements H1 to H4 are controlled to perform predetermined conversion, thereby converting DC power between the input DC terminals into AC power, and output AC power between the AC terminals (Nd1, Nd2).

Further, in the switching elements H1 to H4, the diodes DH1 to DH4 are connected in anti-parallel, respectively. However, in the case of the MOSFET, since the diode is parasitic in the inside of the MOSFET device, it is not necessary to add a diode. . That is to say, the addition of the diodes DH1 to DH4 may be omitted.

The switching circuit 2 is fully bridged by the switching elements S1 to S4 composed of N-type MOSFETs. The third switching branch formed by connecting the switching element S1 (the fifth switching element) and the switching element S2 (the sixth switching element) in series by the node Nd3, and the switching element S3 (the seventh switching element) are connected in series by the node Nd4 The fourth switching branch formed by the switching element S4 (the eighth switching element) is connected in parallel, and the two ends of the third switching branch are used as the DC terminals, and the node Nd3-the node Nd4 is used as the alternating current terminal.

The control terminals of the switching elements S1 to S4 composed of MOSFETs are connected to the control means 5 and controlled.

The switching circuit 2 rectifies the AC power between the input AC terminals (Nd3, Nd4) and converts it into DC power, and outputs DC power between the DC terminals.

Further, in the switching elements S1 to S4, the diodes DS1 to DS4 are connected in anti-parallel, respectively. However, in the case of the MOSFET, it is not necessary to add a diode for the same reason as the switching circuit 1.

A smoothing capacitor C1 is connected between the DC terminals of the switching circuit 1, and a smoothing inductance L and a smoothing capacitor C2 are connected in series between the DC terminals of the switching circuit 2.

Further, DC power supplies V1 and V2 are connected in parallel to the smoothing capacitors C1 and C2.

Further, a resonant capacitor Cr, a resonant inductor Lr, and a winding N1 of the transformer T are connected in series between the AC terminals (Nd1, Nd2) of the switching circuit 1. Further, the leakage inductance component of the transformer T may have the same function as the resonance inductance Lr.

A winding N2 of the transformer T is connected between the AC terminals (Nd3, Nd4) of the switching circuit 2.

Transformer T magnetically couples windings N1, N2.

The resonance capacitor Cr has a DC component that removes the current flowing through the winding N1, thereby reducing the effect of the transformer T bias.

The auxiliary resonance circuit 3 is a switching element (auxiliary resonance switching element) Sb composed of an N-type MOSFET and an auxiliary resonance capacitor Cb. They are connected in series (series connected).

A diode DSb is connected in anti-parallel to the switching element Sb. However, in the case of a MOSFET, for the same reason as described above, it is not necessary to additionally attach a diode.

This auxiliary resonance circuit 3 is connected between the DC terminals of the switching circuit 2. The operation of the auxiliary resonance circuit 3 will be described later in detail.

The voltage clamp circuit 4 is configured by connecting a switching element (clamping switching element) S0 composed of an N-type MOSFET and a clamp capacitor Cc in series.

A diode DS0 is connected in anti-parallel to the switching element S0. However, in the case of a MOSFET, for the same reason as described above, it is not necessary to additionally attach a diode.

The voltage clamping circuit 4 is connected between the DC terminals of the switching circuit 2 to suppress application of a surge voltage between the terminals.

In this way, the DC-DC converter 101 of the first embodiment shown in FIG. 1 is configured by a voltage-type full-bridge circuit composed of a smoothing capacitor C1 and switching elements H1 to H4 (switching circuit 1), and is smoothed. The current-type full-bridge circuit composed of the inductor L and the switching elements S1 to S4 (switching circuit 2) is connected by a transformer T.

Further, the DC terminal of the current-type full-bridge circuit is connected to an auxiliary resonance circuit 3 composed of the switching element Sb and the auxiliary resonance capacitor Cb, and a voltage composed of the switching element S0 and the clamp capacitor Cc. Clamp circuit 4.

That is to say, with the above configuration, the direct current of the direct current power source V1 The force is converted into AC power in a voltage-type full-bridge circuit, that is, a smoothing capacitor C1 and a switching circuit 1, and then converted into an AC voltage in the transformer T, and then in a current-type full-bridge circuit, that is, a smoothing inductor L and a switching circuit 2 Transform it into DC power again. Further, the characteristics of the DC-DC converter are optimized by the auxiliary resonance circuit 3 and the voltage clamp circuit 4.

Further, a voltage sensor 11 is connected to the smoothing capacitor C1 to detect a DC input-in voltage of the voltage type full bridge circuit.

Further, a voltage sensor 12 is connected to the smoothing capacitor C2 to detect a DC input-in voltage of the current-type full-bridge circuit.

In addition, a current sensor 10 is connected to the smoothing capacitor C2 to detect the current of the smoothing inductor L, that is, the DC output current of the current-type full-bridge circuit.

The voltage sensors 11, 12 and the current sensor 10 are connected to the control means 5.

The control means 5 refers to the circuit information obtained from the voltage sensors 11, 12 and the current sensor 10, and is reflected in the control of the switching elements H1 to H4, S1 to S4, and the control of the switching elements Sb and S0.

<About circuit action>

Next, the circuit operation of the DC-DC converter 101 of the first embodiment shown in Fig. 1 will be described.

The DC-DC converter 101 may be used when the auxiliary resonant circuit 3 switching element (auxiliary resonant switching element) Sb is turned on (ON), and may be used by turning it off (OFF).

When the switching element Sb is turned on and the auxiliary resonance circuit 3 is activated, a higher voltage is generated. Further, a large amount of electric power can be transmitted from the DC power source V1 to the DC power source V2.

However, in the case where the load of the DC load 6 is light, or a small voltage is required, or when the power consumption is small, it is not necessary to generate a high voltage or transmit a large amount of power. At this time, the switching element Sb is turned OFF to stop the auxiliary resonance circuit 3.

Therefore, it is preferable to turn the switching element Sb ON or OFF, and to use the auxiliary resonance circuit 3 in response to the situation.

Next, a circuit operation when the switching element Sb is turned off will be described (ACT 1).

In addition, the circuit operation when the switching element Sb is turned on thereafter (operation 2) will be described.

<action 1. (Switching element Sb: OFF)>

2A to 2F and FIG. 3, an operation 1 for supplying electric power to the DC power source V2 from the DC power source V1 when the switching element Sb is in the OFF state will be described.

2A to 2F are schematic diagrams showing circuit operations in modes a to f showing changes in the operating state of the DC-DC converter 101, respectively. First, the operation waveforms of FIG. 3 will be described first, and the circuit operations of FIGS. 2A to 2F will be described in order after FIG. 3 is described.

Action Wave of Action 1

FIG. 3 is a schematic diagram showing the action waveform of the action 1.

In Fig. 3, the horizontal axis represents time lapse, and the vertical direction reveals items such as gate signals, currents flowing through the circuit, and voltages of circuit elements, and their state changes over time.

In addition, in FIG. 3, the period a~f corresponds to the period of the mode a~f.

Further, VgH1 to VgH4 and VgS0 respectively indicate gate signals input to the control terminals of the switching elements H1 to H4 and S0.

Further, since the switching element Sb is kept OFF, the gate signal of the switching element Sb is omitted in FIG.

Further, the current ILr indicates the current flowing through the resonance inductance Lr, and the direction from the node Nd1 to the node Nd2 is taken as the forward direction.

The current IL indicates the current flowing through the smoothing inductance L, and is in the forward direction flowing in the direction of the direct current power source V2.

The current IN2 represents the current flowing in the winding N2, and is forward in the direction from the node Nd4 to the node Nd3.

The currents ICb and ICc respectively indicate the currents flowing through the auxiliary resonance capacitor Cb and the clamp capacitor Cc, and both of them take the direction of charging as the forward direction.

Further, the voltage VLr represents the voltage across the resonant inductor Lr.

Voltage VN2 represents the voltage across winding N2.

In the present specification, the absolute value of the voltage is a voltage equal to or lower than the voltage across the switching element in the ON state or the forward voltage drop of the diode, and is referred to as zero voltage.

Further, when the voltage across the switching element is zero voltage, the operation of turning the switching element on is referred to as "zero voltage switching". Zero voltage switch has suppression The effect of switching loss.

<Circuit action in mode a~f>

2A to 2F, the circuit operations in modes a to f will be described for each mode. In addition, modes a~f correspond to Figures 2A-2F, respectively.

Mode a

2A is a schematic diagram showing the ON/OFF state of each switching element in the mode a of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the mode a (period a in FIG. 3), the switching elements H2 and H4 are in an ON state (VgH2 and VgH4 are High), and the switching elements H1 and H3 are in an OFF state (VgH1 and VgH3 are Low).

At this time, the electric energy accumulated in the resonant inductor Lr (and the resonant capacitor Cr) becomes a current, and flows through the resonant capacitor Cr, the switching element H2, the diode DH4 (switching element H4), and the winding N1 (IN2 is a negative current).

At this time, the current ILr flowing out of the resonant inductor Lr flows substantially at a negative constant value.

The current flowing out at this time flows substantially at a constant value as described above, so that the voltage VLr across the resonant inductor Lr is close to 0 (VLr = Lr. dILr / dt).

Further, since the switching element H2 is turned ON, the DC power supply V1 is applied to the diode DH1 as a reverse voltage, so that no current flows in the diode DH1. In addition, the diode DH4 is turned on, and the DC power supply V1 becomes a reverse power. Since it is applied to the diode DH3, the current does not flow in the diode DH3.

Further, the switching element S0 is in an OFF state, and the current of the smoothing inductance L is supplied to the DC power source V2 through the winding N2 and the diodes DS1 to DS4.

In addition, the path through which the current flows has a first path, that is, a diode DS2 → a winding N2 → a diode DS3; and a second path, that is, a diode DS2 → a diode DS1; and a third path, that is, Diode DS4 → Diode DS3.

As described above, even if only the configuration of the diodes DS1 to DS4 is constituted by a full-wave rectifying circuit, the path through which the current flows is ensured.

However, when a MOSFET is used as the switching elements S1 to S4 at this time, when the switching elements S1 to S4 are turned on, the currents flowing through the diodes DS1 to DS4 are shunted to the switching elements S1 to S4. It is possible to reduce losses.

Therefore, it is useful to use the switching elements S1 to S4.

In this way, when a diode connected in anti-parallel with the MOSFET or a forward current of the diode flows through the body diode of the MOSFET, the MOSFET is turned ON to reduce the loss. , hereinafter referred to as "synchronous rectification".

Mode b

2B is an ON/OFF state of each switching element in mode b of the DC-DC converter 101 according to the first embodiment of the present invention, and is in each circuit flow. A schematic diagram of the current direction.

In the mode b (period b in Fig. 3), when the switching element H2 is turned off (VgH2 → Low in Fig. 3), the current flowing through the switching element H2, the previous current path is cut off. As a result, the resonant inductor Lr is subjected to a current change, resulting in a high back electromotive force (VLr = Lr.dILr / dt). Due to this high voltage, the current corresponding to the current flowing through the switching element H2 is commutated to the diode DH1 and flows into the DC power source V1.

When a current flows to the diode DH1, the voltage across the switching element H1 connected in parallel with the diode DH1 becomes small, and the switching element H1 (VgH1 → High, zero voltage switch during b in FIG. 3) is turned on. In addition, by performing a zero voltage switch, noise is suppressed or power loss is reduced.

The current of the resonant inductor Lr flows through the diode DH4 (switching element H4), the winding N1, the resonant inductor Lr, the resonant capacitor Cr, and the diode DH1 (switching element H1) to the DC power source V1.

The current flows in the above-described path, whereby the voltage of the DC power source V1 is applied to the resonant inductor Lr, and the absolute value of the current ILr of the resonant inductor Lr is gradually decreased (b period of FIG. 3).

Further, during this b period, the switching elements S2 and S3 which were originally turned ON for the synchronous rectification of the switching circuit 2 are first turned off before the end of the next mode c.

Further, the current path through which the winding N2 and the switching circuit 2 flow is substantially the same as the period a during the period b. However, the current value flowing in the winding N2 or the switching elements S2, S3 and the diodes DS2, DS3 The ratio of the current will vary.

Mode c

2C is a schematic diagram showing the ON/OFF state of each switching element in the mode c of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the aforementioned mode b, when the current ILr of the resonant inductor Lr decreases and reaches zero, it becomes the state of the mode c.

In the mode c (period c in FIG. 3), the switching elements H1 and H4 are turned on in the period of the b period (VgH1 and VgH4 are High), and the current ILr of the resonant inductor Lr gradually increases in the opposite direction.

Along with this, the direction of the current flowing through the windings N1, N2 is reversed (in FIG. 2C and FIG. 2B in the opposite direction) and gradually increases, and the currents of the diodes DS2 and DS3 gradually decrease.

Mode d

2D is a schematic diagram showing the ON/OFF state of each switching element in the mode d of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the last state of mode c, as described above, the currents of the diodes DS2, DS3 will gradually decrease to reach zero.

However, for a short period of time, the diodes DS2 and DS3 have a reverse recovery current. In addition, in the diode, it is possible to flow in the opposite direction in a short time after the current flows in the forward direction. The current flowing at this time in the opposite direction is called the reverse recovery current.

Moreover, once the diodes DS2 and DS3 are reversely recovered (the current does not flow in the opposite direction), current flows in the diodes DS2 and DS3, and the reverse recovery current is reversed to two. Polar body DS0. This state is the state of mode d.

When the timing of the current flowing out to the diode DS0 is calculated, the switching element S0 is turned on (VgS0 → High) (zero voltage switch).

Further, the voltage of the DC power source V1 is applied to the winding N1. Further, the voltage VN2 generated in the winding N2 is applied to the DC power source V2 through the diodes DS1, DS4 and the smoothing inductance L, and supplies energy to the DC power source V2. Further, the voltage VN2 generated in the winding N2 is applied to the clamp capacitor Cc through the diode DS0, and the clamp capacitor Cc is charged. Further, by the action of the clamp capacitor Cc, the surge voltage is suppressed from being applied to the switching elements S2 and S3.

Mode e

2E is a schematic diagram showing the ON/OFF state of each switching element in the mode e of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the last state of mode d, as described above, the clamp capacitance Cc is charged. Starting from mode d, the current ICc of the clamp capacitor Cc continues to decrease until it reaches zero, and becomes the initial state of mode e.

Since the switching element S0 is in the ON state, the current ICc of the clamp capacitor Cc is turned to discharge, and the discharge current (absolute value) is gradually increased. another, Since it is a discharge current, it is -ICc, and the current ICc in the period e in Fig. 3 is represented as a negative value.

Mode f

2F is a schematic diagram showing the ON/OFF state of each switching element and the direction of current flowing in each circuit in the mode f of the DC-DC converter 101 according to the first embodiment of the present invention.

In the mode f (f period in FIG. 3), when the switching element H4 is turned off (VgH4 → Low), the current originally flowing through the switching element H4 is switched to the diode DH3, and the mode f is in the initial state.

At this time, the switching element H3 is turned on (VgH3 → High) (zero voltage switching). Further, the operation of turning on the switching element H3 is slightly delayed by a period of time from the start of the mode f in the period f of Fig. 3, and VgH3 becomes High.

In FIG. 2F, since the switching element H4 is turned off as described above, the current ILr of the resonant inductor Lr flows through the paths of the winding N1, the diode DH3, the switching element H1, and the resonant capacitor Cr.

Further, the switching element S0 is turned off (VgS0 → Low). As a result, the current flowing through the switching element S0 is switched to the diodes DS2 and DS3. At this time, as long as the switching elements S2 and S3 are turned on, synchronous rectification is performed.

Further, the voltage between the node Nd3 and the node Nd4 becomes close to zero, and no voltage is applied to the winding N2 (VN2 during f in Fig. 3). At the same time, no voltage is applied to the winding N1.

Further, in the mode f, as in the mode a, the current IL generated by the electric energy accumulated in the smoothing inductance L is supplied to the DC power source V2 through the winding N2 and the diodes DS1 to DS4.

This mode f is the symmetric action of mode a. Then, after the symmetrical motion of the mode b~e, the mode a is returned.

In the case of the operation 1 in which the switching element Sb is turned off, the charge and discharge of the auxiliary resonance capacitor Cb are not performed. Further, in general, there is a parasitic capacitance component between both ends of the switching element, so that a slight current may actually flow.

<action 2. (Switching element Sb: ON)>

Next, an operation 2 of supplying power to the DC power source V2 from the DC power source V1 when the switching element Sb is in an ON state will be described with reference to FIGS. 4A to 4H and FIG. 5 for the DC-DC converter 101 shown in FIG. 1.

Further, the first embodiment of the present invention is characterized in that the auxiliary resonance circuit 3 is provided. Therefore, the operation 2 and its action and effect when the switching element (auxiliary resonance switching element) Sb is turned on have a great feature.

4A to 4H are schematic diagrams showing circuit operations in modes A to H showing changes in the operating state of the DC-DC converter 101, respectively.

In addition, FIG. 5 is a schematic diagram of the action waveform of the action 2. In FIG. 5, the periods A to H correspond to the periods in which the modes A to H are performed. However, the direction of voltage and current in Figure 5 is defined as in Figure 3.

In addition, with respect to the patterns a to f (period a to f, 6 periods) in Fig. 3, as shown in Fig. 5, the mode becomes A to H (period) A~H, 8 period). In other words, 2 modes are added (2 periods). This added mode (period) is mode D (period D) and mode G (period G) in FIG.

Therefore, the mode d~e (period d~e) of FIG. 3 corresponds to the mode E~F (period E~F) in FIG. 5, and the mode f (period f) of FIG. 3 corresponds to the mode H in FIG. (Period H).

Further, since the switching element Sb is kept ON, the gate signal of the switching element Sb is omitted in FIG.

Next, FIG. 4A to FIG. 4H will be sequentially described.

Mode A

4A is a schematic diagram showing the ON/OFF state of each switching element in the mode A of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4A is the same as FIG. 2A except that the switching element Sb is kept ON, and the mode A in the operation 2 operates as the mode a in the operation 1.

Therefore, the repeated explanation is omitted.

Mode B

4B is a schematic diagram showing the ON/OFF state of each switching element in the mode B of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In FIG. 4B, except that the switching element Sb is kept ON, it is the same as FIG. 2B. Similarly, the mode B action in action 2 is the same as the mode b action in action 1.

Therefore, the repeated explanation is omitted.

Mode C

4C is a schematic diagram showing the ON/OFF state of each switching element in the mode C of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

4C is the same as FIG. 2C except that the switching element Sb is kept ON, and the mode C in the operation 2 operates as the mode c in the operation 1.

Therefore, the repeated explanation is omitted.

Mode D

4D is a schematic diagram showing the ON/OFF state of each switching element in the mode D of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the last state of mode C, the currents of the diodes DS2, DS3 will gradually decrease and reach zero.

However, for a short period of time, the diodes DS2 and DS3 have reverse recovery currents.

Once the diodes DS2 and DS3 are restored in reverse, no current flows through the diodes DS2 and DS3. The reverse recovery current flows to the auxiliary resonant capacitor Cb (current ICb), and the auxiliary resonant capacitor Cb Start charging. This state is the state of mode D.

Further, when the electrostatic capacitance values of the auxiliary resonance capacitor Cb and the clamp capacitor Cc are compared, the relationship is Cb ≪ Cc. However, in mode D (period D), the voltage of the clamp capacitor Cc is higher than the voltage of the auxiliary resonant capacitor Cb, so that the current does not flow into the clamp capacitor Cc.

Further, the auxiliary resonant capacitor Cb is a resonant circuit formed by the resonant inductor Lr passing through the transformer T.

Further, the capacitance value of the resonance capacitor Cr and the auxiliary resonance capacitor Cb are compared, and the ratio of the turns of the winding N1 to the winding N2 is considered to be the relationship of Cb ≪ Cr. The relationship between the resonant capacitor Cr and the auxiliary resonant capacitor Cb and the resonant inductor Lr is in a series connection relationship. Therefore, the combined capacitance value of the resonant capacitor Cr and the auxiliary resonant capacitor Cb is substantially equal to the relationship between Cb and Cr. Cb. In addition, the resonance capacitor Cr is not so much a resonance, but it is more effective to cut off the DC component (preventing the transformer T from being biased).

Therefore, the resonance frequency is substantially determined by the auxiliary resonance capacitor Cb and the resonance inductance Lr. However, the transformer T is an ideal transformer.

The resonant current between the resonant inductor Lr and the auxiliary resonant capacitor Cb flows into the auxiliary resonant capacitor Cb.

In the period D of FIG. 5, the current ICb flowing through the auxiliary resonance capacitor Cb is a positive current (charging current) and flows largely in comparison with other periods.

Further, the voltages of the windings N1 and N2 gradually rise together with the voltage of the auxiliary resonance capacitor Cb (refer to VN2, period D in FIG. 5). Resonance The current ILr of the inductor Lr and the current IN2 of the winding N2 continue to increase gradually (refer to IN2, period D in FIG. 5).

Mode E

4E is a schematic diagram showing the ON/OFF state of each switching element in the mode E of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the last state of mode D, when the voltage of the auxiliary resonant capacitor Cb reaches the voltage of the clamp capacitor Cc, the current that originally charges the auxiliary resonant capacitor Cb is switched to the diode DS0 and the clamp capacitor Cc is charged. . This state is the state of mode E.

In the period D of the previous moment, the resonance inductance Lr accumulates a large amount of electric energy by the auxiliary resonance capacitor Cb, so that the clamp capacitor Cc has a large current (compared to the period d in FIG. 3) (refer to The positive current of ICc is charged in the period E) in Fig. 5.

Further, since the current starts to flow through the diode DS0, the switching element S0 is turned on at this time (VgS0 → High) (zero voltage switching). By turning on the switching element S0, the electric energy is efficiently charged at the clamp capacitor Cc.

Further, the voltage generated in the winding N2 is applied to the DC power source V2 through the diodes DS1, DS4 and the smoothing inductance L, and electric energy is supplied to the DC power source V2.

In addition, the voltage generated in the winding N2 is applied to the clamp capacitor Cc through the diode DS0 as described above, so the clamp capacitor Cc is charged. However, the current ICc for charging the pair of clamp capacitors Cc is gradually reduced, and accordingly, the current ILr of the resonant inductor Lr is gradually reduced.

Therefore, the resonant inductor Lr generates a voltage that is positive in the direction of the node Nd2. The voltage VLr generated by the resonant inductor Lr and the voltage of the DC power source V1 are added and applied to the winding N1.

As a result, a voltage higher than the period d in FIG. 3 is generated in the winding N2.

Mode F

4F is a schematic diagram showing the ON/OFF state of each switching element in the mode F of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In the last state of mode E, the current ICc of the clamp capacitor Cc is reduced to reach zero. This state is the initial state of mode F.

At this time, since the switching element S0 is in the ON state, the current ICc of the clamp capacitor Cc is turned to discharge (-ICc), and the absolute value of the current gradually increases.

Along with this, the current of the resonant inductor Lr continues to decrease gradually. Therefore, as in the mode E, the voltage VLr generated by the resonant inductor Lr and the voltage of the direct current power source V1 are added and applied to the winding N1.

Mode G

4G is an ON/OFF state of each switching element in the mode G of the DC-DC converter 101 according to the first embodiment of the present invention, and is in each circuit flow. A schematic diagram of the current direction.

In mode G, when the switching element H4 is turned off, the current originally flowing through the switching element H4 is commutated to the diode DH3. This is the initial state of mode G. Further, since the current starts to flow through the diode DS3, the switching element H3 is turned on (zero voltage switching).

Since the switching elements H2 and H4 are OFF and the switching elements H1 and H3 are turned on, the current of the resonant inductor Lr flows through the paths of the winding N1, the diode DH3, the switching element H1, and the resonant capacitor Cr.

Further, when the switching element S0 is turned off, the current flowing through the switching element S0 will flow from the auxiliary resonant capacitor Cb, and the auxiliary resonant capacitor Cb will start discharging (negative current) (refer to the current ICb, period G in FIG. 5) ).

In addition, the auxiliary resonance capacitor Cb charges electric energy during the period D, and discharges the accumulated electric energy during the period G, but the current ICb value (negative value) at this time is compared with the period e and the period f of FIG. In terms of the current ICb value of the junction, the absolute value is large.

This is because in the case of FIG. 3, even if the auxiliary resonance capacitor Cb has energy accumulation, the switching element Sb is turned off and does not discharge.

Further, a voltage of the auxiliary resonance capacitor Cb is applied to the winding N2.

Further, the voltage generated in the winding N1 is applied to the resonant inductor Lr, and the current ILr of the resonant inductor Lr is gradually decreased. The voltage VN2 of the winding N2 gradually decreases as the auxiliary resonant capacitor Cb is discharged (refer to ICb, ILr, VN2 of the period G in Fig. 5, respectively).

Mode H

4H is a schematic diagram showing the ON/OFF state of each switching element in the mode H of the DC-DC converter 101 according to the first embodiment of the present invention, and the current direction flowing through each circuit.

In mode H, the voltage of the auxiliary resonant capacitor Cb becomes zero voltage, and when the discharge of the auxiliary resonant capacitor Cb is completed (current ICb is 0, mode H in Fig. 5), the current corresponding to the discharge current is commutated to the diode. DS2, DS3. This is because the current IL flowing through the smoothing inductance L cannot be changed abruptly.

Further, at this time, as long as the switching elements S2 and S3 are turned on, synchronous rectification is performed.

In addition, no voltage is applied to the winding N2, so that no voltage is generated in the winding N1, and the current of the resonant inductor Lr remains unchanged.

Further, as in the mode A, the current of the smoothing inductance L is supplied to the direct current power source V2 through the winding N2 and the diodes DS1 to DS4.

This mode H is the symmetric action of mode A. Then, after the symmetrical motion of the mode B~G, the mode A is returned.

<Comparison of Action 2 and Action 1>

In the case of the operation 2 in which the switching element Sb is turned on, the charge/discharge current of the clamp capacitor Cc is increased as compared with the operation 1 in which the switching element Sb is turned off.

Along with this, in the modes E to F, the current slope of the resonant inductor Lr increases in the negative direction, and the resonant inductor Lr generates a voltage. This voltage will be straight Since the voltage of the current source V1 is added and applied to the winding N1, a higher voltage is applied to the winding N1 than in the case of the operation 1, and a voltage higher than the operation 1 can be generated in the winding N2.

"About the working cycle of switching components"

In the above-described operations 1 and 2, the time ratio (hereinafter referred to as the duty cycle) during which the switching element H1 (H2) and the switching element H4 (H3) are both turned ON is changed, thereby adjusting the supply to the DC power supply V2. Electricity, that is, output power.

The more the work cycle increases, the greater the output power. When the switching element H1 (H2) and the switching element H4 (H3) are turned ON/OFF at the same time, the duty cycle becomes maximum.

Further, when the voltage of the DC power source V1, that is, the input voltage is lowered, the output power can be suppressed from decreasing by increasing the duty cycle. However, when the input voltage is further lowered, even if the duty cycle is maximized, the required output power cannot be obtained.

Therefore, in order to reduce the input voltage as described above, even if the duty cycle is maximized, the required output power cannot be obtained, and the auxiliary resonance switching element Sb of the above operation 2 is turned ON, and the auxiliary resonance circuit 3 secures the The required output power.

"The ratio of the turns of the transformer"

In order to obtain the required output power even under low input voltage, just the turns ratio of the transformer (number of turns of winding N2 / winding N1) The number of turns can be increased. When the turns ratio is increased, even if the input voltage is low, a high voltage is generated in the winding N2, so that large output power can be easily obtained.

However, if the turns ratio is increased like this, a higher voltage is generated in the winding N2 when the input voltage becomes higher. Therefore, the voltage applied to the switching elements S1 to S4 also becomes high, and a switching element having a high withstand voltage is required as the switching elements S1 to S4. In general, if the withstand voltage of the switching element becomes high, the loss also becomes large, so that the efficiency of the DC-DC converter is easily lowered.

In order to cope with this problem, the auxiliary resonance circuit 3 is provided. A large output voltage can be obtained by the auxiliary resonance circuit 3, so that it is not necessary to specifically increase the turns ratio of the transformer. In other words, by the ON/OFF of the switching element Sb and the duty cycle selection of the switching elements H1 to H4, the voltage applied to the switching elements S1 to S4 can be suppressed regardless of whether the input voltage is low or high. Get the right output power.

In other words, when the operation 1 and the operation 2 are operated in the same duty cycle, as described above, the voltage generated in the winding N2 in the operation 2 is higher, so the output power in the operation 2 becomes larger.

In the DC-DC converter 101 of the first embodiment, when the input voltage is low, the switching element Sb is turned on, and the voltage drop generated in the winding N2 is suppressed to obtain the required output power.

On the other hand, when the input voltage is high, the switching element Sb is turned off, and the voltage rise in the winding N2 is suppressed to suppress the voltage applied to the switching elements S1 to S4 from rising.

Thus, the DC-DC converter 101 of the first embodiment of the present invention is In the case where a desired output power is to be obtained in a wide input voltage range, a switching element having a low withstand voltage can be used as the switching elements S1 to S4.

The switching element having a low withstand voltage has a small loss, so that the DC-DC converter 101 can achieve high efficiency.

"Times for Action 1 and Action 2"

In the above description, the switching element Sb is turned on when the input voltage is low, and the switching element Sb is turned off when the input voltage is high. However, the switching element Sb may be turned on when the output power is large. When the output power is small, the switching element Sb is turned off.

Alternatively, the switching element Sb may be turned on when the duty cycle is increased to a certain level, and the switching period Sb is turned off after the duty cycle is reduced to a certain level.

Further, when the ON/OFF state of the switching element Sb is switched, in order to suppress the instability of the output, between the threshold value for changing the switching element Sb to the ON state and the threshold for changing the switching element Sb to the OFF state, Set the adjustment sensitivity (hysteresis; hysteresis).

Further, if the load is reduced while the switching element Sb is changed to the ON state, and the load is increased while the switching element Sb is changed to the OFF state, the output power fluctuation accompanying the switching of the switching element Sb can be suppressed.

Further, by turning the switching element Sb into an ON state, in the mode D, when the diodes DS2 and DS3 are reversely restored, they are applied to the diode. The voltage between the two ends of DS2 and DS3 rises later, so that the loss associated with the reverse recovery of the diodes DS2 and DS3 can be reduced.

Further, in the mode G, the current of the resonant inductor Lr is reduced, so that the interrupting current when the switching element H1 (H2) is turned off is reduced, and the effect of reducing the loss is also obtained.

"Other Usages of DC-DC Converter 101"

In the DC-DC converter 101 of the first embodiment, the switching elements S1 to S4 are appropriately switched by the control means 5 in the switching circuit 2, whereby the DC power can be converted into AC power. Further, in the switching circuit 1, the switching elements H1 to H4 are appropriately switched by the control means 5, whereby the alternating current power can be converted into direct current power. Further, the transformer T can convert the AC power of the winding N2 into the AC power of the winding N1.

Therefore, if the DC-DC converter 101 is appropriately controlled, the DC power supply V1 can be supplied with power from the DC power supply V2.

At this time, when the input current from the DC power source V2 is large, the switching element Sb is turned on, whereby when the switching elements S1 to S4 are turned off, the voltage applied between the ends of the switching elements S1 to S4 rises. It is delayed, so it has the effect of reducing the loss.

(Second embodiment)

Next, a second embodiment of the DC-DC converter of the present invention will be described.

Figure 6 is a diagram showing a DC-DC converter 102 according to a second embodiment of the present invention. A schematic diagram of the circuit configuration, and a connection between the DC-DC converter 102 and the DC power source V1, and a connection between the DC-DC converter 102 and the DC power source V2 and the DC load 6 are shown.

In FIG. 6, the DC-DC converter 102 is connected between the DC power supply V1 and the DC power supply V2 to which the DC load 6 is connected, and converts the DC voltage of the DC power input from the DC power supply V1 into a different DC voltage to DC. The power source V2 and the DC load 6 supply power.

In addition, DC power is supplied to the DC power source V1 from the DC power source V2 as necessary.

The DC-DC converter 102 includes a switch circuit 1 (first switch circuit), a switch circuit 21 (second switch circuit), an auxiliary resonance circuit 3, and a voltage clamp circuit 41.

Further, the DC-DC converter 102 includes a smoothing capacitor C1, a smoothing inductance L and a smoothing capacitor C2, a resonant capacitor Cr and a resonant inductor Lr, and a winding N1 (primary winding) and a winding N21 (first secondary winding) and N22. Transformer T1 (second secondary winding). Further, one end of the winding N21 (first secondary winding) is connected to one end of the N22 (second secondary winding) to constitute a connector.

In addition, in FIG. 6, the control means 5 and the voltage sensor of the ON/OFF state of the control switching element (H1 - H4, S11, S21, S31, S41, Sb in FIG. 6) shown in FIG. 11, 12, the circuit element corresponding to the current sensor 10, but the illustration is omitted.

The circuit configuration of the switch circuit 1 of Fig. 6 is the same as that of the switch circuit 1 of Fig. 1, and the overlapping description will be omitted.

The switch circuit 21 includes a switching element S11 (eighth switching element) and a switching element S21 (sixth switching element).

One end of the switching element S11 and the other end of the winding N21 are connected by a node Nd41, and one end of the switching element S21 and the other end of the winding N22 are connected by a node Nd31.

Further, the other end of the switching element S11 is connected to the other end of the switching element S21. Further, a connection point between the windings N21 and N22 and a connection point between the switching elements S11 and S21 are used as a DC terminal, and a node Nd31-node Nd41 is used as an alternating current terminal.

A smoothing capacitor C1 is connected between the DC terminals of the switching circuit 1, and a smoothing inductance L and a smoothing capacitor C2 are connected in series between the DC terminals of the switching circuit 21.

DC power supplies V1 and V2 are connected in parallel to the smoothing capacitors C1 and C2, respectively.

Further, a resonant capacitor Cr, a resonant inductor Lr, and a winding N1 are connected in series between the AC terminals (Nd1, Nd2) of the switching circuit 1.

Windings N21 and N22 are connected between the AC terminals (Nd31 and Nd41) of the switching circuit 21.

Transformer T magnetically couples windings N1, N21, N22.

One end of the winding N21 and one end of the winding N22 are connected to each other to form a DC terminal.

The other end of the winding N21 is connected to the node Nd41, and the other end of the winding N22 is connected to the node Nd31.

a resonant capacitor Cr connected to one end of the winding N1 of the transformer T, There is a DC component that removes the current flowing through the winding N1, and the effect of biasing the transformer T is reduced.

The auxiliary resonance circuit 3 is configured to include a switching element (auxiliary resonance switching element) Sb and an auxiliary resonance capacitor Cb. This configuration is the same as that of the first embodiment, and the overlapping description will be omitted.

The voltage clamp circuit 41 includes a switching element (clamping switching element) S31, a switching element (clamping switching element) S41, and a clamp capacitor Cc1.

One end of the switching element S31, one end of the switching element S41, and one end of the clamp capacitance Cc1 are connected to each other.

Further, the other end of the switching element S31 is connected to the node Nd41.

Further, the other end of the switching element S41 is connected to the node Nd31.

Further, the other end of the clamp capacitor Cc1 is connected to one end of the smoothing capacitor C2.

The voltage clamp circuit 41 is connected between the DC terminal of the switch circuit 21 and the node Nd41 and the node Nd31 to suppress application of a surge voltage between the terminals.

As described above, in the DC-DC converter of the second embodiment, the switching circuit 21 can be configured by the two switching elements S11 and S21. Therefore, the number of switching elements can be reduced as compared with the switching circuit 2 of the first embodiment.

Further, the operation of the circuit and the effects of the invention are substantially the same as those of the first embodiment.

Further, the DC-DC converter 102 of the second embodiment is also like the first As in the embodiment, as long as the DC-DC converter 102 is appropriately controlled, electric power can be supplied from the DC power source V2 to the DC power source V1.

(Third embodiment)

Next, a third embodiment of the DC-DC converter of the present invention will be described.

7 is a circuit configuration diagram of a DC-DC converter 103 according to a third embodiment of the present invention, and a DC-DC converter 103 connected to a DC power source V1, and a DC-DC converter 103 and a DC power source V2 and a DC load. The connection of 6 constitutes a schematic diagram.

In FIG. 7, the DC-DC converter 103 is connected between the DC power source V1 and the DC power source V2 to which the DC load 6 is connected, and converts the DC voltage of the DC power input from the DC power source V1 into a different DC voltage to DC. The power source V2 and the DC load 6 supply power.

In addition, DC power is supplied to the DC power source V1 from the DC power source V2 as necessary.

The DC-DC converter 103 includes a switch circuit 1 (first switch circuit), a switch circuit 22 (second switch circuit), an auxiliary resonance circuit 31 (first auxiliary resonance circuit), and an auxiliary resonance circuit 32 (second auxiliary resonance circuit) ), voltage clamping circuit 41.

Further, the DC-DC converter 103 includes a smoothing capacitor C1, a smoothing inductance L1 (first smoothing inductance), a smoothing inductance L2 (second smoothing inductance), a smoothing capacitor C2, a resonant capacitor Cr and a resonant inductor Lr, and a winding N1. Transformer (primary winding) and winding N23 (secondary winding) T2.

In addition, in FIG. 7, the control means 5 and the voltage of the ON/OFF state of the control switching element (corresponding to H1 to H4, S11, S21, S31, S41, Sb1, and Sb2 in FIG. 7) are provided. The sensors 11 and 12 and the current sensor 10 correspond to circuit elements, but are not shown.

The circuit configuration of the switch circuit 1 of Fig. 7 is the same as that of the switch circuit 1 of Fig. 1, and the overlapping description will be omitted.

The switch circuit 22 includes a switching element S11 (eighth switching element) and a switching element S21 (sixth switching element).

One end of the switching element S11 and the other end of the winding N23 are connected by a node Nd42, and one end of the switching element S21 and the other end of the winding N23 are connected by a node Nd32.

Further, the node Nd42 is connected to one end of the smoothing inductance L1, and the node Nd32 is connected to one end of the smoothing inductance L2.

Further, the other end of the smoothing inductor L1 is connected to the other end of the smoothing inductor L2, and is configured as a connector.

Further, the other end of the switching element S11 is connected to the other end of the switching element S21. Both ends of the switching element S11 are between DC terminals. Further, both ends of the switching element S21 are between the DC terminals.

Further, the node Nd32-Nd42 is used as an alternating current terminal.

A smoothing capacitor C1 is connected between the DC terminals of the switching circuit 1.

Further, a smooth connection is formed between the connection point of the switching elements S11 and S21 and the connection point of the smoothing inductance L1 and the smoothing inductance L2. Capacitor C2.

DC power supplies V1 and V2 are connected in parallel to the smoothing capacitors C1 and C2, respectively.

Further, a resonant capacitor Cr, a resonant inductor Lr, and a winding N1 are connected in series between the AC terminals (Nd1, Nd2) of the switching circuit 1.

A winding N23 is connected between the AC terminals (Nd32, Nd42) of the switch circuit 22.

The auxiliary resonant circuit 31 is configured by connecting a switching element (auxiliary resonant switching element) Sb1 and a secondary resonant capacitor Cb1 in series.

The auxiliary resonant circuit 32 is configured by connecting a switching element (auxiliary resonant switching element) Sb2 and a secondary resonant capacitor Cb2 in series.

An auxiliary resonance circuit 31 is connected between both ends of the switching element S11, and an auxiliary resonance circuit 32 is connected between both ends of the switching element S21.

The voltage clamp circuit 41 includes a switching element (clamping switching element) S31, a switching element (clamping switching element) S41, and a clamp capacitor Cc1.

One end of the switching element S31 is connected to one end of the switching element S41 and one end of the clamp capacitor Cc1.

The other end of the switching element S31 is connected to the node Nd42, the other end of the switching element S41 is connected to the node Nd32, and the other end of the clamp capacitor Cc1 is connected to one end of the smoothing capacitor C2.

The voltage clamp circuit 41 suppresses application of a surge voltage to the terminals of the node Nd42 and the node Nd32.

As described above, the DC-DC converter of the third embodiment 103, the number of windings can be reduced as compared with the second embodiment.

The operation of the circuit and the effects of the invention are substantially the same as those of the first embodiment, and thus the overlapping description will be omitted.

Further, as in the first embodiment, the DC-DC converter 103 of the third embodiment can supply electric power to the DC power source V1 from the DC power source V2 as long as the DC-DC converter 103 is appropriately controlled.

(Other embodiments)

The embodiments of the present invention have been described in detail above with reference to the drawings. However, the present invention is not limited by the embodiments and the modifications thereof, and various modifications and the like may be made without departing from the scope of the invention.

In the first embodiment, the auxiliary resonance circuit 3 is connected between the DC terminals of the switching circuit 2, but the connection is not limited thereto. The first auxiliary resonant circuit corresponding to the auxiliary resonant circuit 3 may be provided, and the first auxiliary resonant circuit may be connected between both ends of the switching element S1, and the second auxiliary resonant circuit may be connected between both ends of the switching element S2.

Further, the auxiliary resonance circuit using the bidirectional switch as the switching element Sb is connected between the nodes Nd3-Nd4, and the above-described effects of the present invention can be obtained as well.

Further, in the second embodiment, the auxiliary resonance circuit 3 is connected between the DC terminals of the switch circuit 21, but the first and second auxiliary resonance circuits may be provided, and the first auxiliary device may be connected between both ends of the switching element S11. In the resonant circuit, a second auxiliary resonant circuit is connected between both ends of the switching element S21.

Further, the auxiliary resonance circuit which uses the bidirectional switch as the switching element Sb is connected between the nodes Nd31-Nd41, and the above-described effects of the present invention can also be obtained.

Further, in the third embodiment, the auxiliary resonance circuit 31 is connected between both ends of the switching element S11, and the auxiliary resonance circuit 32 is connected between both ends of the switching element S21. However, the first embodiment may be used. The bidirectional switch is connected as an auxiliary resonance circuit of the switching element Sb of the auxiliary resonance circuit 3 to the node Nd32-Nd42. The effects of the aforementioned invention can also be obtained at this time.

In the above, the switch circuit 1 in the first to third embodiments is configured by a full-bridge circuit composed of the switching element H1 to the switching element H4. However, the function of the switch circuit 1 is to convert DC power into AC power. Therefore, it is not limited to a full bridge type circuit configuration.

In addition, in FIG. 1, an example in which the switching elements (H1 to H4, S1 to S4, Sb, and S0) are N-type MOSFETs is disclosed. However, if it is controlled in consideration of polarity, it may be a P-type MOSFET. Further, as long as it has a switching function, it is not limited to a MOSFET.

In other words, IGBT (Insulated Gate Bipolar Transistor), Bipolar junction transistor, BiCMOS (Bipolar Complementary Metal Oxide Semiconductor) can be used. ) and other components and devices.

Further, the voltage clamp circuit described in the first to third embodiments or the auxiliary resonance power including the auxiliary resonance capacitor and the switching element The circuit and method for controlling the switching elements of the circuit 3, or the circuit and method for combining the same, can be widely applied to DC-DC conversion in which a voltage type switching circuit and a current type switching circuit having a smooth inductor are connected by a transformer. Device.

1‧‧‧Switch circuit (1st switch circuit)

2‧‧‧Switch circuit (2nd switch circuit)

3‧‧‧Auxiliary resonant circuit

4‧‧‧Voltage clamp circuit

5‧‧‧Control means

6‧‧‧DC load

10‧‧‧ Current Sensor

11, 12‧‧‧ voltage sensor

101‧‧‧DC-DC converter

C1‧‧‧Smoothing Capacitor (1st smoothing capacitor)

C2‧‧‧Smoothing capacitor (2nd smoothing capacitor)

Cb‧‧‧Auxiliary Resonance Capacitor

Cc‧‧‧Clamp Capacitor

Cr‧‧‧Resonance Capacitor

DH1~DH4, DS0~DS4, DSb‧‧‧ diode

H1‧‧‧Switching element (first switching element)

H2‧‧‧Switching element (2nd switching element)

H3‧‧‧Switching element (3rd switching element)

H4‧‧‧Switching element (4th switching element)

S0‧‧‧Switching element (clamping switch element)

S1‧‧‧ switching element (5th switching element)

S2‧‧‧ switching element (6th switching element)

S3‧‧‧Switching element (7th switching element)

S4‧‧‧Switching element (8th switching element)

Sb‧‧‧ switching element (auxiliary resonant switching element)

Lr‧‧‧Resonance Inductance

N1‧‧‧ winding (primary winding)

N2‧‧‧ winding (secondary winding)

Nd1~Nd4‧‧‧ nodes

T‧‧‧Transformer

V1‧‧‧DC power supply (1st DC power supply)

V2‧‧‧DC power supply (2nd DC power supply)

Claims (15)

A DC-DC converter comprising: a transformer having a primary winding and a secondary winding, wherein the primary winding is magnetically coupled to the secondary winding; and the first switching circuit has a plurality of switching elements between the DC terminals a first smoothing capacitor is connected, and the primary winding is connected between the AC terminals to convert DC power into AC power; the second switching circuit has a plurality of switching elements, and the secondary winding is connected between the AC terminals, between the DC terminals The second smoothing capacitor and the smoothing inductor are connected in series to convert the alternating current power into the direct current power; the resonant inductor is connected in series with the primary winding and/or the secondary winding; and the auxiliary resonant circuit is connected to the alternating current of the second switching circuit Between the terminals, or between the DC terminals of the second switch circuit, or between the AC terminal and the DC terminal of the second switch circuit, and a series connection body in which the auxiliary resonance capacitor and the auxiliary resonance switching element are connected in series; a first DC power source connected in parallel with the first smoothing capacitor, and a DC connected in parallel with the second smoothing capacitor The load is supplied with electric power, and when the auxiliary resonance switching element is in an OFF state, charging of the auxiliary resonance capacitor is prevented. A DC-DC converter according to claim 1, wherein the resonant inductor has a leakage inductance of the transformer. The DC-DC converter of claim 1, wherein when the voltage of the first DC power source is lowered, and/or when When the voltage of the DC load supply rises, and/or when the current supplied to the DC load increases, the auxiliary resonance switching element is changed from the OFF state to the ON state. A DC-DC converter according to claim 1, further comprising a voltage clamp circuit connected between the DC terminals of the second switch circuit or the AC terminal of the second switch circuit and the second smoothing Between one end of the capacitor. The DC-DC converter of claim 4, wherein the voltage clamping circuit has a series connection of a clamp switch element and a clamp capacitor. The DC-DC converter according to claim 1, wherein the first switching circuit including the plurality of switching elements includes a first switching branch, and the first switching element and the second switching element are connected in series And the second switching branch is formed by connecting the third and fourth switching elements in series and connected in parallel with the first switching branch; and the two ends of the first switching branch are used as DC terminals The connection point between the first and second switching elements and the connection point between the third and fourth switching elements is defined as an alternating current terminal. The DC-DC converter according to claim 1, wherein the second switching circuit including the plurality of switching elements includes a third switching branch, and the fifth and sixth switching elements are connected in series. and And the fourth switching branch is formed by connecting the seventh and eighth switching elements in series and connected in parallel with the third switching branch; and the two ends of the third switching branch are used as DC terminals. The connection point between the fifth and sixth switching elements and the connection point between the seventh and eighth switching elements is defined as an alternating current terminal. The DC-DC converter according to claim 1, wherein the secondary winding includes a connection body that connects one end of the first secondary winding and one end of the second secondary winding, and the second switching circuit a sixth and eighth switching elements, wherein one end of the eighth switching element is connected to the other end of the first secondary winding, and one end of the sixth switching element is connected to the other end of the second secondary winding, and the first The other end of the 8 switching element is connected to the other end of the sixth switching element, and a connection point between the connection point of the eighth and sixth switching elements and the first and second secondary windings is used as a DC terminal. One end of the eighth switching element and one end of the sixth switching element serve as an alternating current terminal. The DC-DC converter according to claim 1, wherein the smoothing inductor includes a connector that connects one end of the first smoothing inductor and one end of the second smoothing inductor, and the second switching circuit includes a sixth , the eighth switching element, The other end of the first smoothing inductor is connected to one end of the eighth switching element, and the other end of the second smoothing inductor is connected to one end of the sixth switching element, and the other end of the eighth switching element and the sixth switch are connected The other end of the element is connected, and the second smoothing capacitor is connected between a connection point of the first smoothing inductor and a connection point between the eighth and sixth switching elements, and between the two ends of the eighth switching element The two ends of the sixth switching element are used as a DC terminal, and one end of the eighth switching element and one end of the sixth switching element are used as an alternating current terminal. The DC-DC converter according to claim 9, further comprising: a first auxiliary resonance circuit connected in parallel with the eighth switching element; and a second auxiliary resonance circuit connected in parallel with the sixth switching element. A DC-DC converter according to claim 1, wherein the first DC power source is supplied with electric power from a second DC power source connected in parallel with the second smoothing capacitor. The DC-DC converter according to claim 11, wherein when the electric power is supplied to the first DC power source from the second DC power source, when the current supplied to the first DC power source increases, The auxiliary resonance switching element changes from an OFF state to an ON state. The DC-DC converter of claim 1, wherein the DC-DC converter further includes a resonant capacitor connected in series with the primary winding and/or the secondary winding. The DC-DC converter according to any one of claims 1 to 13, further comprising a diode, and each of a plurality of the switching elements constituting the first switching circuit and the second switching circuit Or any anti-parallel connection. A DC-DC converter according to claim 5, wherein a diode is provided, which is connected in anti-parallel with the clamp switch element.