TW201509111A - Motor speed control apparatus - Google Patents

Abstract

A speed proportional gain limit arithmetic unit calculates a speed proportional gain limit on the basis of a motor speed and load inertia ratio. A speed integration time constant torque command low-pass filter limit arithmetic unit calculates a speed integration time constant limit and a cutoff frequency limit on the basis of the speed proportional gain limit. A speed controller receives a speed command and outputs a torque command on the basis of a speed proportional gain and a speed integration time constant. A torque command low-pass filter allows a torque command of frequencies lower than a cutoff frequency to pass and thereby reduces harmonics contained in the torque command.

Description

Motor speed control device Field of invention

The present invention relates to a high-speed rotation control capable of simultaneously achieving stable high-speed rotation control to a medium-speed domain, and a wide-range output field in the case of having a core offset of an encoder and a core offset of a spindle. Motor speed control device.

Background of the invention

The spindle of the working machine for high-precision tapping is required to achieve high-precision tapping at high-speed milling and high-speed tapping, and heavy-duty cutting at low speed.

Therefore, the heavy-duty cutting and high-speed rotation are realized by the constant output control by the magnetic field weakening of the induction motor, and the base speed is set to be high, and the deterioration of the characteristic due to the weakening of the magnetic field is reduced to achieve high precision. Tapping processing.

3 is a block diagram of a conventional speed control device for a motor. The speed control device of the motor operates as follows.

First, the speed command is compared with the motor rotational speed ωm from the speed calculator 15, and the speed controller 20 is rotated by the speed command and the motor. The q-axis current command IqC is obtained by the deviation of the speed ωm. The motor rotation speed ωm output from the speed calculator 15 is calculated using the position feedback detected by the encoder 10.

The q-axis current command IqC is compared with the q-axis current feedback IqF from the coordinate converter 25, and the q-axis current integration controller 30 integrates the difference between the q-axis current command IqC and the q-axis current feedback IqF. The sum of the q-axis current command IqC and the q-axis current integrator output is obtained, and the q-axis current command IqCB after the integral compensation is obtained.

On the other hand, with reference to the motor rotational speed ωm, the d-axis current command Idc is given the necessary exciting current, and the d-axis current command IdC is compared with the d-axis current feedback IdF from the coordinate converter 25, and the d-axis current integral controller 35 The difference integral between the d-axis current command IdC and the d-axis current feedback IdF. The sum of the d-axis current command IdC and the d-axis current integrator output is obtained, and the d-axis current command IdCB after the integral compensation is obtained.

The slip frequency calculator 40 calculates the slip frequency command ωs from the q-axis current command IqC and the d-axis current command IdC. The slip frequency command ωs is added to the motor rotation speed ωm output from the speed calculator 15. The primary frequency command ω1 is obtained by adding the slip frequency command ωs to the motor rotational speed ωm. The primary frequency command ω1 is integrated by the integrator 45 so as to become the pole pair Pm of the motor, and the stator position command θmc is obtained.

The coordinate converter 50 converts the q-axis current command IqCB after the integral compensation and the d-axis current command IdCB coordinate after the integral compensation based on the stator position command θmc, and obtains the three-phase current commands Iuc, Ivc, and Iwc.

The phase current controller 55 is based on the three-phase current commands Iuc, Ivc, Iwc and the motor currents Iu and Iv control the currents of the respective phases, and calculate the three-phase voltage commands Vuc, Vvc, and Vwc. The three-phase voltage commands Vuc, Vvc, and Vwc are supplied to the motor 80 through the PWM controller 60 and the power converter 70, and the motor 80 is driven in response to the three-phase voltage commands Vuc, Vvc, and Vwc.

The q-axis current feedback IqF and the d-axis current feedback IdF are obtained by the coordinate converter 25 converting the motor currents Iu and Iv based on the stator position command θmc.

The excitation current command (d-axis current command Idc) obtained by referring to the motor rotation speed ωm is constant in the constant torque field as shown in FIG. 3, and is inversely proportional to the increase in the motor rotation speed ωm in the fixed output field. That is, the field weakening control is performed.

In the field of field weakening in which the excitation current command is inversely proportional to the increase in the motor rotational speed ωm, since the magnetic flux of the motor is reduced with the decrease of the excitation current command, the actual motor for the torque current command (q-axis current command IqC) is reduced. The torque drops.

Because the response of the speed control system is determined by the multiplication of the gain of the speed controller 20 and the actual motor torque of the torque current command (q-axis current command IqC), the response of the speed control system decreases with the weakening of the magnetic field. .

In high-precision milling machines, in order to synchronize the spindle motor with the feed axis motor, the response of the speed control system is expected to be fast. If the response of the speed control system is slow, the machining accuracy of the tapping process will be degraded due to the delay in response to the command. Therefore, the speed control response of the speed range in the motor is prevented from being lowered by setting the base speed of the motor to be high.

4 and 5 are graphs showing the torque-rotation speed characteristics of the motor 80 when the field weakening control is performed. Fig. 4 is a graph showing the torque-rotation speed characteristics used in the conventional tapping process, and the base speed is higher than that of Fig. 5 showing the torque-rotation speed characteristic in the case where the fixed output field is wide, and the field of the output is expanded. . Compared with the torque-rotation speed characteristic of Fig. 5, the torque rotation speed characteristic of Fig. 4 is that the torque that can be output in the low speed range is small. Therefore, there is a problem that low-speed heavy cutting capability requiring a large cutting torque in a low speed region is low.

In recent years, the demand for higher torque and lowering of the rotational speed of high-speed cutting is required to achieve a wider range of output.

However, if the substrate speed is lowered and the output field is expanded, the decrease in the magnetizing current in the medium speed region will become large. Therefore, the actual motor torque reduction for the torque command is increased, and the speed response in the medium speed region is decreased, which has a problem that the high-precision tapping processing cannot be realized.

If the speed gain is increased in order to improve the problem, and the maximum rotation speed is increased in order to improve the machining accuracy of the high-speed milling machine, the motor current is likely to be affected by the fluctuation of the rotation speed.

The core offset caused by the motor 80 having the bearing of the motor itself is offset from the core of the encoder 10, and the rotation of the motor 80 detects the rotational variation of the motor 80. The error of the rotation of the motor 80 is 1 when the rotation is once, because the motor rotation speed ωm by the speed calculator 15 The result of the calculation is the differentiation of the motor position. Therefore, the higher the speed is, the larger the amplitude is, and the speed change occurs once per rotation.

When the speed command is constant, the speed control system outputs a torque command in such a manner as to suppress the speed variation. Since the torque is a slight component of the speed, the torque command (q-axis current Iqc) includes a variation for suppressing the speed variation. This fluctuation is a torque command (q-axis current Iqc) that occurs once per rotation, and the amplitude of the fluctuation is larger as the speed is higher.

Further, since the stator position command θmc is obtained by integrating the primary frequency command ω1 and performing the pole-number multiple, the stator position command θmc is a SIN θ, COS θ signal which is a multiple of the number of poles per motor rotation when the slip is small. Further, since the stator position command θmc and the q-axis and d-axis current commands Iqc and Idc are multiplied by the coordinate converter 50 to calculate the motor current commands Iuc, Ivc, and Iwc of the respective phases, the q-axis current command Iqc is applied to the motor. 1 When there is a change in the rotation, a high-frequency wave appears in the motor current command.

For example, when the motor 80 is 4 poles, the frequency of SIN θ and COS θ is twice the motor rotation frequency. If the q-axis current command Iqc changes once in 1 rotation, the motor current command Iuc in each phase is used. , Ivc, Iwc appeared 3 times high frequency wave. When the gain of the speed controller 20 is high, a large fluctuation occurs in the q-axis current command Iqc, and the high-frequency waves of the motor current commands Iuc, Ivc, and Iwc of the respective phases also become large.

FIG. 6 is a view showing a simulation result when the gain of the speed controller 20 is high and the q-axis current command Iqc is changed once in one rotation.

As shown in the figure, the q-axis current command Iqc exhibits a change in the rotation of the motor 1 once. The frequency of SINθ is twice that of the change. Moreover, U phase The current command Iuc is a harmonic change that is three times the change in the rotation of the motor 1 once.

Patent Document 1 listed below has an example of suppressing the core shift of such an encoder. Patent Document 1 below discloses the following techniques.

The device of Patent Document 1 is a position control device that controls a rotational position by attaching a position detector to a rotating shaft and returning a signal based on a position from a position detector in order to detect a rotational position of the rotating shaft.

The detection error data due to the eccentricity and mounting error of the object in the position detector is calculated based on the value of the output signal of the position detector. When the rotation axis is rotated at a constant speed at which the speed of the drive control motor is not sufficiently fast, the detection error data is calculated based on the occurrence point of the one rotation signal output from the position detector at each predetermined time. The value of the output signal of the position detector is calculated.

The calculated detection error is stored in the position control device, and the detection error data is corrected for the rotation command position of the motor that drives the rotary shaft as the rotation command position to the motor.

Advanced technical literature Patent literature

Patent Document 1 Japanese Patent Laid-Open No. 11-27973

Summary of invention

However, in the case where the rotational variation of the motor 1 is stable, the rotational variation of the motor 1 is the core offset caused by the bearing of the motor itself. It is intertwined with several elements such as the core offset of the encoder, the core offset caused by the eccentricity of the gap, and even the core offset of the mechanical spindle connected to the motor.

Therefore, in the case where a stable core offset cannot be obtained, there is a problem that the influence of the core offset remains.

As described above, it is difficult for the speed control device of the conventional motor to realize stable high-speed rotation control, high-speed response control up to the medium-speed range, and wide-ranging output field in order to meet the demand for increasing processing performance.

The present invention is an invention constructed to solve such a conventional problem, and an object thereof is to provide stable high-speed rotation control at the same time even when the core of the encoder is offset and the core of the spindle is shifted. High-speed response speed control in the speed range and a wide range of speed control devices for the motor in the output field.

The speed control device for the motor related to the present invention for achieving the above object is a speed proportional gain limit value calculator, a speed integral time constant torque command low pass filter limit value calculator and a speed controller, a torque command low pass filter .

The speed proportional gain limit value calculator calculates the speed proportional gain limit value using the motor speed to load inertia ratio. Speed integral time constant torque command Low-pass filter limit value calculator calculates the speed integral time constant limit value and calculates the cutoff frequency limit value by using the speed proportional gain limit value. The speed controller is the input speed command, using the speed proportional gain and speed The integral time constant is output and the torque command is output. The torque command low-pass filter passes a torque command that has a lower frequency than the cutoff frequency, and lowers the harmonics contained in the torque command.

According to the speed control device of the motor according to the present invention constructed as above, the core offset caused by the bearing having the motor itself, the core offset of the encoder, and even the core offset of the mechanical spindle connected to the motor, etc. In the case of various core shifts, the motor can also be stably rotated to high speed. In addition, even if the magnetic field is weakened, the actual motor torque reduction for the torque command is small, and a high speed control response up to the medium speed range can be achieved, and a large low-speed torque can be simultaneously achieved by a wide range of output fields. .

10, 110‧‧‧ encoder

15, 115‧‧‧ speed calculator

20, 120‧‧‧ speed controller

25, 50, 150, 155‧‧‧ coordinate converters

30‧‧‧q axis current integral controller

35‧‧‧d-axis current integral controller

40‧‧‧ slip frequency calculator

45, 255‧‧‧ integrator

55‧‧‧phase current controller

60, 160‧‧‧ PWM controller

70, 170‧‧‧Power Converter

80, 180‧‧‧ motor

100‧‧‧Motor speed control device

121‧‧‧Load inertia ratio memory

122‧‧‧Speed proportional gain limit value calculator

123‧‧‧Speed integral time constant torque command low pass filter limit value calculator

124‧‧‧Speed proportional gain set value memory

125‧‧‧Speed proportional gain calculator

126‧‧‧Speed integral constant torque command low-pass filter set value memory

127‧‧‧Speed integral constant torque command low-pass filter set value calculator

130‧‧‧Restrictor

132‧‧‧Maximum primary current command

134‧‧‧torque limit value calculator

135‧‧‧Torque command low-pass filter

140‧‧‧q-axis current calculator

145‧‧‧q axis current controller

220‧‧‧Magnetic field weakening

225‧‧‧Magnetic flux calculator

230‧‧‧ slip frequency calculator

240‧‧‧Magnetic flux controller

245‧‧‧d axis current controller

Fig. 1 is a block diagram of a speed control device for a motor according to the embodiment.

Fig. 2 is a graph showing the speed proportional gain limit value characteristic of the rotational speed of the motor.

3 is a block diagram of a conventional speed control device for a motor.

Fig. 4 is a graph showing the torque-rotation speed characteristics used in conventional tapping processing.

Fig. 5 is a graph showing the torque-rotation speed characteristics in the case where the output field is wide.

Fig. 6 is a view showing a simulation result when the gain of the speed controller is high and the q-axis current command is changed once in one rotation.

Form for implementing the invention

The speed control device of the motor associated with the present invention is based on a motor speed and load inertia ratio and is limited by a speed controller and a torque command low pass filter. The torque limit value is calculated from the maximum primary current command and the d-axis current command, and the torque command after the torque limit is passed through the low-pass filter to suppress the fluctuation of the core offset caused by the torque command. In addition, the torque command-torque characteristic due to the weakening of the magnetic field is prevented by the q-axis current calculator. Thereby, the motor can be stabilized even in the case of various core offsets such as core offset caused by the bearing of the motor itself, offset of the core of the encoder, and even the core offset of the mechanical spindle connected to the motor. Rotate to high speed. In addition, even if the magnetic field is weakened, the actual motor torque reduction for the torque command is small, and a high speed control response up to the medium speed range can be achieved, and a large low-speed torque can be simultaneously achieved by a wide range of output fields. .

Next, an embodiment of a speed control device that can exhibit the above-described characteristics of the motor according to the present invention will be described with reference to the drawings.

[Configuration of Motor Speed Control Device 100]

Fig. 1 is a block diagram of a speed control device 100 for a motor according to the present embodiment.

The motor speed control device 100 includes a speed controller 120, a load inertia ratio memory unit 121, a speed proportional gain limit value calculator 122, a speed integral time constant torque command low-pass filter limit value calculator 123, and a speed proportional gain set value memory. Portion 124, speed proportional gain calculator 125, speed integral time constant torque command low pass filter set value memory unit 126, the limiter 130, the maximum primary current command unit 132, the torque limit value calculator 134, the torque command low-pass filter 135, the q-axis current calculator 140, and the q-axis current controller 145 are given as the q-axis voltage command VqC. system.

The speed controller 120 compares the speed command with the motor speed ωm and the difference between the input speed command and the motor speed ωm. The speed controller 120 is constructed of a PI controller. The motor speed ωm is calculated by the speed calculator 115 using the position signal of the encoder 110 that detects the rotation of the motor 180. The speed controller 120 is an input speed command and outputs a torque command using the speed proportional gain KVP and the speed integral time constant t.

The load inertia ratio memory unit 121 stores JL/JM which is a ratio of the load inertia JL to the motor inertia JM. This is to change the limit value of the speed proportional gain in response to the magnitude of the load inertia ratio. The speed proportional gain limit value calculator 122 calculates the speed proportional gain limit value in accordance with the motor speed ωm and the load inertia ratio JL/JM. The speed integral time constant torque command low pass filter limit value calculator 123 is based on the speed proportional gain limit value, the calculated speed integral time constant limit value, and the torque command low pass filter limit value.

The speed proportional gain set value storage unit 124 stores the speed proportional gain set value. The speed proportional gain calculator 125 calculates the speed proportional gain KVP using the speed proportional gain limit value based on the speed proportional gain set value. The speed integral time constant torque command low-pass filter set value storage unit 126 stores the speed integral time constant t and the torque command low-pass filter fc. Speed integral time constant torque command low-pass filter set value calculation unit 127 is a speed integral time constant, torque command low-pass filter The set value is based on the speed integral time constant value and the torque command low-pass filter fc limit value, and the speed integral time constant and the torque command low-pass filter fc are calculated. The speed controller 120 outputs a torque command using the difference between the input speed command and the motor speed ωm, the speed proportional gain KVP, and the speed integral time constant TVI.

The limiter 130 limits the magnitude of the torque command output by the speed controller 120. The maximum primary current command unit 132 is a maximum primary current IPC output that the power converter 170 can output. The torque limit value calculator 134 calculates the torque limit value TLIM using the d-axis current command IdC, the maximum primary current IPC, and the magnetic flux φ2 outputted by the magnetic flux calculator 225 to be described later. The torque limit value TLIM is set to the limiter 130.

Therefore, the limiter 130 limits the size of the input torque command to be included in the ±TLIM. The torque command low-pass filter 135 sets the cutoff frequency of the torque command low-pass filter 135 calculated by the speed integral time constant torque command low-pass filter set value calculator 127. Therefore, the torque command low-pass filter 135 passes a torque command that lowers the frequency than the set cutoff frequency, and suppresses fluctuations in higher harmonics due to the core offset of the encoder or the spindle included in the torque command. The torque command low-pass filter 135 is constituted by a secondary low-pass filter.

The q-axis current calculator 140 calculates the q-axis current command IqC from the torque command after passing through the torque command low-pass filter 135. The q-axis current limiter 145 is a deviation of the q-axis current command IqC output from the q-axis current calculator 140 and the q-axis current feedback IqF output from the coordinate converter 155, and outputs a q-axis voltage command VqC. The q-axis current limiter 145 is a proportional integral controller to make.

Further, the motor speed control device 100 includes a magnetic field weakening unit 220, a magnetic flux calculator 225, a magnetic flux controller 240, and a d-axis current controller 245 as a system for giving a d-axis voltage command VdC.

As shown in the figure, the field weakening unit 220 stores a relationship between the magnetic flux command φ2C for the motor speed ωm, and outputs a magnetic flux command φ2C based on the motor speed ωm. The magnetic flux calculator 225 calculates the magnetic flux φ2 by using the d-axis current feedback IdF output from the coordinate converter 155.

The magnetic flux controller 240 is a deviation of the magnetic flux command φ2C output from the input field weakening unit 220 and the magnetic flux φ2 output from the magnetic flux calculator 225, and outputs a d-axis current command Idc. The d-axis current controller 245 is a deviation of the d-axis current command IdC output from the input magnetic flux controller 240 and the d-axis current feedback IdF output from the coordinate converter 155, and outputs a d-axis voltage command VdC. The magnetic flux controller 240 and the d-axis current controller 245 are constituted by a proportional integral controller. Incidentally, the q-axis current feedback IqF and the d-axis current feedback IdF are obtained by the coordinate converter 155 converting the motor currents Iu and Iv coordinates based on a stator position command θmc to be described later.

Further, the motor speed control device 100 has a slip frequency calculator 230, an integrator 255, and coordinate converters 150 and 155 as a system for performing coordinate conversion.

The slip frequency calculator 230 is a q-axis current command IqC output from the q-axis current calculator 140 and a magnetic flux command φ2 output from the magnetic flux calculator 225, and calculates a slip frequency command ωs. The integrator 255 is an input of the slip frequency command ωs output from the slip frequency calculator 230 and the slave speed calculator 115. The primary frequency command ω1 obtained by adding the output motor rotational speed ωm integrates the primary frequency command ω1. The stator position command θmc is obtained by setting the frequency-first frequency command to the pole number Pm of the motor. The stator position command θmc is output to the coordinate converters 150 and 155.

The coordinate converter 150 converts the q-axis voltage command VqC and the d-axis voltage command VdC coordinates based on the input stator position command θmc, and obtains the three-phase voltage commands Vuc, Vvc, and Vwc.

The coordinate converter 155 converts the motor currents Iu and Iv based on the input stator position command θmc, and obtains the q-axis current feedback IqF and the d-axis current feedback IdF.

Further, the motor speed control device 100 has a PWM controller 160 and a power converter 170 as a system for driving the motor 180.

The PWM controller 160 inputs the three-phase voltage commands Vuc, Vvc, and Vwc output from the coordinate converter 150, and outputs a PWM signal for switching the power converter 170 based on the input three-phase voltage commands Vuc, Vvc, and Vwc.

The power converter 170 is a PWM signal output from the PWM controller 160 to switch the semiconductor switching element provided therein to drive the motor 180.

[Operation of Motor Speed Control Device 100]

First, the speed proportional gain limit value calculator 122 is a load inertia ratio which is calculated by using the motor speed ωm output from the speed calculator 115 and the load inertia ratio memory unit 121, and the speed proportional gain limit value is obtained by the following equation. KVPLIM.

KVPLIM={KVPLIM2-K2(|ωm|-ω2)}/(JL/JM)

However, (ω2<|ωm|)

Here, KVPLIM2 is the speed proportional limit value of the rotational speed ω2, JL is the load inertia, JM is the motor inertia, and (JL/JM) is the load inertia ratio.

When the rotational speed is greater than or equal to ω2, as shown in the above equation, the speed proportional gain limit value increases in proportion to the rotational speed and decreases inversely proportional to the load inertia ratio (JL/JM).

In addition, when the rotational speed is ω1 to ω2, the speed proportional gain limit value decreases as the rotational speed increases. The speed proportional gain limit value KVPLIMB before the speed inverse ratio is obtained by the following equation.

KVPLIMB={KVPLIM2+K1(ω2-|ωm|)}

However, (ω1<|ωm| Ω2)

Here, K1 is a coefficient that reduces the speed proportional gain limit value between ω1 and ω2.

Next, the calculation is performed in inverse proportion to the rotation speed and inversely proportional to the load inertia ratio as in the following equation, and the speed proportional gain limit value KVPLIM is obtained.

KVPLIM=KVPLIMB×ω2/|ωm|/(JL/JM)

However, (ω1<|ωm| Ω2)

When the rotational speed is less than or equal to the rotational speed ω1, the speed proportional gain limit value is a constant value as shown in the following equation.

KVPLIM=KVPLIM1

Moreover, ω1 is a rotation speed that is about twice the speed of the substrate, so that the speed is fast. Degrees 0 to ω1 make the speed proportional gain not decrease.

Thus, the upper limit of the speed proportional gain is limited to KVPLIM.

Fig. 2 is a graph showing the speed proportional gain limit value characteristic of the rotational speed of the motor. As shown in the figure, the medium speed ω1 is limited to a certain high speed gain KVPLIM1 up to about twice the base speed ω0, and the accuracy of tapping is improved. When the medium-speed range ω1 is greater than or equal to the increase in the speed proportional gain with the increase in the rotational speed ωm, the harmonics of the motor current are suppressed, and the heat generation of the motor is lowered. In addition, when the load inertia ratio (JL/JM) is large (for example, JL/JM=2), since the torque command for suppressing the speed variation due to the core offset becomes large, the speed proportional gain limit value is decreased ( The graph of JL/JM=2 is used to suppress the generation of higher harmonics of current.

The speed integral time constant limit value is calculated based on KVPLIM and is calculated by the following equation.

TVILIM=1/KVPLIM

The lower limit of the speed integral constant is limited to this value.

The cutoff frequency limit value of the torque command low-pass filter is calculated based on KVPLIM and is calculated by the following equation.

TCLPFLIM=KVPLIM×KTCL

The upper limit of the cutoff frequency of the torque command low pass filter is limited to this value.

The coefficient KTLC is a value that is as small as possible in the range in which the gain of the frequency response characteristic of the speed control system does not occur. This embodiment is 5. Thereby, in the frequency exceeding the response of the speed control system, the torque command low-pass filter is set to a value as small as possible, and the speed proportional gain is matched. According to the limitation, the fluctuation of one rotation of the torque command after the torque command low-pass filter is suppressed, and the generation of the high-frequency wave by the coordinate conversion is suppressed, and the heat of the motor is included in the allowable loss.

Further, by taking the appropriate speed integral time constant value and the torque command low-pass filter cutoff frequency limit value for each speed gain value, the characteristics of the speed control system are such that they do not have an appropriate overshoot. On the other hand, when the load inertia ratio is known, the speed gain limit value is calculated using this value. When it is not known, the torque command and the motor speed are calculated using an iterative least squares method or the like. Alternatively, the speed controller 120 may be configured by an IP controller or the like, and the torque command low-pass filter 135 may be configured by a three-pass low-pass filter or the like. Alternatively, instead of the speed proportional gain limit value, the cutoff frequency limit value or the speed integral time constant limit value of the torque command low-pass filter 135 may be sequentially calculated.

Next, the field weakening unit 220 calculates the magnetic flux command φ2C based on the motor speed ωm output from the speed calculator 115, and reduces the magnetic field at a rotation speed greater than or equal to the base speed ω0.

φ2C=φ2CB (0 |ωm| Ω0)

φ2C=φ2CB. Ω0/|ωm| (ω0<|ωm|)

The magnetic flux command φ2C calculated as described above is compared with the magnetic flux φ2 output from the magnetic flux calculator 225, and the magnetic flux controller 240 outputs the d-axis current command Idc from the deviation between the magnetic flux command φ2C and the magnetic flux φ2.

The torque limit value calculator 134 is based on the following equation and is calculated by the d-axis current command IdC and the maximum primary current command IPC that the inverter can output. Torque limit value TLIM. The calculated torque limit value TLIM is set to the limiter 130, and the limiter 130 limits the torque command output by the speed controller 120 to less than or equal to ±TLIM.

TLIM=Pm×M/L2×φ2×(IqC2-IdC2)1/2

However, L2 is the second inductance, M is the mutual inductance, and Pm is the pole logarithm.

The torque command from the speed controller 120 is passed through the limiter 130 based on the torque limit value TLIM, and the torque command passed through the low-pass filter is obtained by the torque command low-pass filter 135. The q-axis current command IqC is obtained by inputting the torque command passed through the low-pass filter to the q-axis current calculator 140. Then, the q-axis current calculator 145 calculates a q-axis current command based on the following equation.

IqC=L2/(Pm×M×φ2)×(Torque command after passing through the low-pass filter)

In this way, since the q-axis current command is calculated by dividing the magnetic flux φ2, if the magnetic field is weakened and the magnetic flux is reduced, the q-axis current command is increased, and the torque characteristic of the torque command is not reduced by the weakening of the magnetic field.

The q-axis current command is compared with the q-axis current feedback, and the q-axis voltage command VqC is given by the q-axis current controller 145. The d-axis current command is compared with the d-axis current feedback, and the d-axis voltage command VdC is given by the d-axis current controller 245.

The slip frequency calculator 230 calculates the slip frequency command ωs based on the q-axis current command IqC output from the q-axis current calculator 140 and the magnetic flux φ2 output from the magnetic flux calculator 225 based on the following expression.

Ωs=M×R2/L2×(IqC/φ2)

Then, the calculated slip frequency command ωs is added to the motor speed ωm to obtain the primary frequency command ω1.

The primary frequency command ω1 is integrated, and the pole value of the motor is multiplied by Pm to obtain a fixed value position command θmc.

The coordinate converter 150 converts the q-axis voltage command VqC and the d-axis voltage command VdC coordinates based on the stator position command θmc, and obtains the three-phase voltage commands Vuc, Vvc, and Vwc. The three-phase voltage commands Vuc, Vvc, and Vwc are supplied to the motor 180 through the PWM controller 160 and the power converter 170, and the motor 180 is driven in response to the three-phase voltage commands Vuc, Vvc, and Vwc.

The q-axis current feedback IqF and the d-axis current feedback IdF are obtained by the coordinate converter 155 converting the motor currents Iu and Iv based on the stator position command θmc.

The magnetic flux calculator 225 calculates the magnetic flux φ2 from the d-axis current feedback IdF by the following equation.

Φ2=1/(1+L2/R2×S)×M×IdF

Incidentally, the d-axis voltage command and the q-axis voltage command can be used to mount a non-dry controller to control the interference between the d-axis and the q-axis. In addition, the inside of the d-axis and q-axis current control systems can also be constructed by a three-phase current control system.

As described above, the conventional speed control device for a motor is difficult to achieve stable high-speed rotation and high-speed response to a medium-speed range, and to secure a wide output field. However, the speed control device of the motor related to the present invention is based on the motor rotation speed and the load inertia ratio, and is limited by the speed controller and the torque command low-pass filter, and the torque limit value is calculated by the maximum primary current command and the d-axis current command. Torque limit The post-torque torque command suppresses the variation in the torque command due to the core offset of the encoder or the spindle through the low-pass filter. In addition, the torque command-torque characteristic due to the weakening of the magnetic field is prevented by the q-axis current calculator.

Thereby, the motor can be stabilized even in the case of various core offsets such as core offset caused by the bearing of the motor itself, offset of the core of the encoder, and even the core offset of the mechanical spindle connected to the motor. Rotate to high speed. In addition, even if the magnetic field is weakened, the actual motor torque reduction for the torque command is small, and a high speed control response up to the medium speed range can be achieved, and a large low-speed torque can be simultaneously achieved by a wide range of output fields. .

100‧‧‧Motor speed control device

110‧‧‧Encoder

115‧‧‧Speed Calculator

120‧‧‧Speed controller

121‧‧‧Load inertia ratio memory

122‧‧‧Speed proportional gain limit value calculator

123‧‧‧Speed integral time constant torque command low pass filter limit value calculator

124‧‧‧Speed proportional gain set value memory

125‧‧‧Speed proportional gain calculator

126‧‧‧Speed integral constant torque command low-pass filter set value memory

127‧‧‧Speed integral constant torque command low-pass filter set value calculator

130‧‧‧Restrictor

132‧‧‧Maximum primary current command

134‧‧‧torque limit value calculator

135‧‧‧Torque command low-pass filter

140‧‧‧q-axis current calculator

145‧‧‧q axis current controller

150, 155‧‧‧ coordinate converter

160‧‧‧PWM controller

170‧‧‧Power Converter

180‧‧‧ motor

220‧‧‧Magnetic field weakening

225‧‧‧Magnetic flux calculator

230‧‧‧ slip frequency calculator

240‧‧‧Magnetic flux controller

245‧‧‧d axis current controller

255‧‧‧ integrator

Claims (8)

A speed control device for a motor, comprising: a speed proportional gain limit value calculator, calculating a speed proportional gain limit value using a motor speed to a load inertia ratio; and a speed integral time constant torque command low pass filter limit value calculator, Calculate the speed integral time constant limit value and calculate the cutoff frequency limit value by using the aforementioned speed proportional gain limit value; speed controller, input speed command, use speed proportional gain and speed integral time constant to output torque command; torque command low pass filter The torque command having a lower frequency than the cutoff frequency is passed, and the higher harmonics included in the torque command are lowered. The speed control device of the motor of claim 1, wherein the speed proportional gain calculator has a motor speed greater than or equal to ω2 such that the speed proportional gain limit value decreases in proportion to the motor speed and decreases inversely proportional to the magnitude of the load inertia ratio The motor speed between ω1 and ω2 which is slower than the aforementioned ω2 is such that the speed proportional gain limit value decreases as the motor speed becomes faster, and the motor speed less than or equal to ω1 limits the speed proportional gain limit value to a certain value. . In the speed control device for the motor of claim 1, the load inertia ratio is stored in the load inertia ratio memory portion, and the load inertia ratio is a ratio of the load inertia JL to the motor inertia JM, that is, JL/LM. The speed control device for the motor of claim 1, wherein the speed integral time constant torque command low pass filter limit value calculator is at the aforementioned speed ratio The reciprocal of the limit value is used as the aforementioned speed integral time constant limit value. In the speed control device for the motor of claim 1, the speed integral time constant torque command low-pass filter limit value calculator is obtained by multiplying the speed ratio limit value by a value of a certain magnification as the cutoff frequency limit value. The speed control device of the motor of claim 5, wherein the predetermined magnification is selected as small as possible in a range in which the gain of the frequency response characteristic of the speed control does not occur (hump) The speed control device for the motor of claim 1 further includes a q-axis current calculator for calculating a q-axis current command by a torque command after the torque command low-pass filter, and a magnetic flux command and a magnetic flux corresponding to the motor speed. The magnetic flux controller that outputs the d-axis current command is driven to drive the motor using the q-axis current command and the d-axis current command. The speed control device for the motor of claim 1, further comprising a torque limit value calculator for calculating a torque limit value using the aforementioned d-axis current command and the magnetic flux and the maximum primary current, and low-passing from the speed controller to the torque command The limiter for limiting the magnitude of the torque command output by the filter sets the aforementioned torque limit value in the limiter.