JP7162759B2 - electric motor drive - Google Patents

Description

The present invention relates to a motor drive device for driving a synchronous motor.

A servo system using a synchronous motor is used as a power source for various mechanical devices. In a general servo system, a speed controller and a current controller are connected in series. A limiter is provided on the output of each controller to protect the synchronous motor and machinery. In general, a power converter that outputs an AC voltage to a synchronous motor has a limit on the maximum voltage that can be output or a limit on the maximum current that can be output. Such limits in the power converter also act like limiters.

Each controller is provided with an integrator for controlling the output so as to eliminate the steady-state error. When the output of each controller saturates due to the limiter, integration continues and integration becomes excessive, resulting in a windup phenomenon in which the output value does not change from the limit value even if the command value changes. It is known. Sustained oscillations can be excited by the windup phenomenon. The windup phenomenon is a factor that lowers the stability of control by the servo system. As one of the methods to prevent the windup phenomenon, when it is detected that the output of each controller is saturated, the command value input to each controller is lowered so that the saturation state is released. It is mentioned that

Japanese Patent Laid-Open No. 2002-200002 discloses a control method for reducing a speed command value when the output voltage of a power converter reaches an upper limit and the output voltage is saturated, in relation to a speed control device for an electric motor. The speed control device according to Patent Document 1 performs voltage phase control such that the phase of the voltage command value leads the dq rotation coordinate when the output voltage of the power converter is saturated, and the phase angle of the voltage command value exceeds the threshold value, the speed command value is decreased by performing an operation for correcting the speed command value.

Japanese Patent No. 5256009

When the control method described in Patent Literature 1 is applied, the electric motor drive device can prevent occurrence of the windup phenomenon by appropriately adjusting the control parameters. However, according to the control method described in Patent Document 1, adjustment of many control parameters is required. Also, the servo system or plant model includes many nonlinear elements that exhibit complex characteristics. For this reason, according to the control method described in Patent Document 1, the motor driving device adjusts the control parameters by trial and error, so the work load required for the adjustment for stable control of the motor is large. There was a problem.

SUMMARY OF THE INVENTION It is an object of the present invention to provide an electric motor drive device capable of reducing the work load required for adjustment for stably controlling the electric motor.

In order to solve the above-described problems and achieve the object, the electric motor driving device according to the present invention converts the value of the phase current flowing in the electric motor into each value of d-axis current and q-axis current, which are currents in a dq coordinate system. Then, a current controller that controls the phase current by determining a voltage command based on the d-axis current, the d-axis current command, the q-axis current, and the q-axis current command, and the voltage amplitude that is the amplitude of the voltage command is obtained. a voltage amplitude calculator; a speed controller that controls the rotational speed by determining a q-axis current command based on the speed command, the rotational speed of the motor, and a speed droop amount that reduces the speed command; a flux-weakening controller that performs flux control to limit the amplitude of the voltage output to the motor by determining the d-axis current command based on the voltage limit value of and the voltage amplitude and the second voltage limit value and a speed droop controller that controls the amount of speed droop based on and. A speed droop controller determines an amount of speed droop that causes the voltage amplitude to be less than the second voltage limit.

ADVANTAGE OF THE INVENTION According to this invention, the electric motor drive device is effective in the ability to reduce the workload required for the adjustment for performing the stable control of an electric motor.

1 is a block diagram showing a configuration example of an electric motor drive device according to Embodiment 1 of the present invention; FIG. 2 is a block diagram showing a configuration example of a speed controller included in the electric motor drive device according to the first embodiment; FIG. FIG. 4 is a diagram for explaining a voltage vector representing the voltage state of the synchronous motor that is the object of control by the electric motor drive device according to the first embodiment; FIG. 2 is a block diagram showing a configuration example of a flux-weakening controller included in the electric motor drive device according to the first embodiment; FIG. 2 is a block diagram showing a configuration example of a speed droop controller included in the electric motor drive device according to the first embodiment; FIG. 4 is a diagram showing an example of a control model for the motor drive device and the synchronous motor according to the first embodiment; A diagram showing a model obtained by approximating the control model shown in FIG. 6 around the operating point in the high-speed region. A first diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. A second diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. A third diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. A fourth diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. A first diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. A second diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. A third diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. A fourth diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. FIG. 4 shows an example of operating waveforms when the electric motor drive device according to the first embodiment is used; FIG. 2 is a diagram showing an example of hardware configuration of an electric motor drive device according to a second embodiment of the present invention; FIG.

An electric motor drive device according to an embodiment of the present invention will be described in detail below with reference to the drawings. In addition, this invention is not limited by this embodiment.

Embodiment 1.
FIG. 1 is a block diagram showing a configuration example of an electric motor drive device according to Embodiment 1 of the present invention. A motor drive device 100 according to the first embodiment drives a synchronous motor 1 . The electric motor drive device 100 is connected to the power converter 3 . A synchronous motor 1 is mechanically connected to a mechanical device 2 . The synchronous motor 1 is the power source of the mechanical device 2 . The mechanical device 2 operates when the power converter 3 outputs an AC voltage to the synchronous motor 1 . Synchronous motor 1 , power converter 3 , and motor drive device 100 constitute a motor system that drives synchronous motor 1 .

In Embodiment 1, the synchronous motor 1 is a permanent magnet synchronous motor in which a rotor is provided with permanent magnets. The synchronous motor 1 may be a wound-field synchronous motor in which a field winding is wound around the rotor, or may be a reluctance synchronous motor that obtains rotational torque by utilizing the saliency of the rotor. Also good. The arrangement of the permanent magnets in the synchronous motor 1 may be an embedded arrangement or a surface arrangement. In Embodiment 1, the synchronous motor 1 is assumed to be a three-phase synchronous motor. The synchronous motor 1 may be a synchronous motor other than a three-phase synchronous motor. For example, the synchronous motor 1 may be a two-phase synchronous motor or a five-phase synchronous motor.

The mechanical device 2 may be any device that operates by driving the synchronous motor 1 . In Embodiment 1, the mechanical device 2 is a refrigerant compressor, which is a representative example of an application in which control adjustment tends to take a long time. Refrigerant compressors are incorporated in equipment such as air conditioners, chillers, and refrigerators. Most of refrigerant compressors have an integrated structure in which an electric motor is incorporated in order to reduce the number of parts. Therefore, in many refrigerant compressors, it is difficult to control and adjust the electric motor alone. Furthermore, in the refrigerant compressor, since the pressure condition changes slowly over time, it takes time for the pressure to stabilize. Since it takes time for the pressure to stabilize, the control adjustment of the refrigerant compressor tends to take a long time.

Refrigerant compressors include various types of compressors such as rotary compressors, scroll compressors, screw compressors, reciprocating compressors, and turbo compressors. Refrigerant compressors are common in that control adjustment is complicated regardless of the type of compressor. The mechanical device 2, the refrigerant compressor, may be any of various types of compressors. The mechanical device 2 may be a device other than a refrigerant compressor.

The power converter 3 converts power input from a power source (not shown) into power in a prescribed form and outputs the power. In Embodiment 1, the power converter 3 is a general-purpose voltage source inverter. A voltage source inverter is a device that switches a DC voltage supplied from a DC voltage source and converts it into a desired AC voltage. Power converter 3 converts DC voltage into AC voltage based on voltage command 12 output from motor drive device 100 , and outputs the converted AC voltage to synchronous motor 1 . Note that the power converter 3 may be another type of circuit such as a current source inverter or matrix converter, or may be a multi-level converter, as long as it can supply a desired AC power to the synchronous motor 1 .

A current detector 4 detects a phase current flowing through the synchronous motor 1 . The type and arrangement of the current detector 4 are not particularly limited. The current detection unit 4 may be a type current sensor using a transformer called a CT (Current Transformer) or a type current sensor using a shunt resistor. The current detector 4 may be a combination of a CT and a shunt resistor. A current detection unit 4 shown in FIG. 1 is arranged in wiring between the synchronous motor 1 and the power converter 3 and measures a phase current flowing through the synchronous motor 1 . The current detector 4 outputs a signal 11 indicating the value of the phase current. Note that the current detection unit 4 may be arranged at a position other than the position shown in FIG. For example, the current detector 4 may be arranged inside the power converter 3 .

When the current detection unit 4 is arranged inside the power converter 3, as a current detection method, a 1-shunt current detection method in which a shunt resistor is arranged on the N side of the DC bus in the power converter 3; A lower arm shunt current detection method in which a shunt resistor is inserted in series with the lower arm can be used. The one-shunt current detection method and the lower arm shunt current detection method have restrictions on the timing at which current detection is possible, but can reduce the component cost compared to the case of using a CT.

When the synchronous motor 1 is a three-phase synchronous motor, the motor drive device 100 calculates the value of the phase current of any two of the three phases from the value of the phase current of the other phase based on Kirchhoff's current law. can be calculated. Therefore, it is sufficient that a current sensor is arranged in any two of the three phases, and a current sensor does not need to be arranged in the other one phase.

The motor driving device 100 controls the synchronous motor 1 by vector control. The electric motor drive device 100 includes a position/speed identifying unit 5 , a speed controller 6 , a dq-axis current controller 7 , a voltage amplitude calculator 8 , a flux-weakening controller 9 , and a speed droop controller 10 .

In order to vector-control the synchronous motor 1, it is necessary to detect or estimate the magnetic pole position _θe and the rotational speed _ωe of the synchronous motor 1. FIG. The position/speed specifying unit 5 specifies the magnetic pole position θ _e and the rotation speed ω _e of the synchronous motor 1 . Specifically, the position/speed specifying unit 5 determines the magnetic pole position θ _e and the rotational Estimate the velocity ω _e . The position/speed specifying unit 5 outputs the specified magnetic pole position θ _e and the specified rotational speed ω _e .

A position sensor for detecting the magnetic pole position _θe may be attached to the synchronous motor 1 . Rotary encoders or resolvers are used as position sensors. A speed sensor such as a tachogenerator may be attached to the synchronous motor 1 instead of the position sensor. It should be noted that the use of a position sensor or a speed sensor may not be suitable for the synchronous motor 1 due to constraints such as usage environment and cost. In Embodiment 1, it is assumed that the electric motor drive device 100 performs position sensorless control. The electric motor drive device 100 is not limited to one that does not use a position sensor or speed sensor, and may use a position sensor or speed sensor. It should be noted that the refrigerant compressor described above is a representative example of an application in which it is difficult to use a position sensor or speed sensor.

Various methods have been proposed for the position sensorless control of the synchronous motor 1, but basically any method may be used in the first embodiment. As a known method, for example, there is a speed estimation method of estimating the state quantity of the synchronous motor 1 using a state observer and adaptively identifying the rotational speed ω _e using the state quantity estimation error. This method is a method called an adaptive observer, and has the advantage of being able to perform robust speed estimation against changes in the induced voltage constant. If the adaptive observer is not used, the magnetic pole position θ _e may be estimated simply from the arctangent component of the velocity electromotive force. This method is called the arctangent method. The arctangent method has the disadvantage that an error in the induced voltage constant causes a speed estimation error, but the calculation is simpler than that of the adaptive observer. Although many other position sensorless control methods have been proposed, any method may be used as long as the magnetic pole position θ _e and the rotational speed ω _e can be estimated.

The speed controller 6 determines the q-axis current command i _q ^* based on the speed command ω ₁ ^* , which is the first speed command, the speed droop amount Δω, and the specified rotational speed ω _e , thereby controlling the synchronous motor 1 to control the rotational speed ω _e .

FIG. 2 is a block diagram showing a configuration example of a speed controller included in the electric motor drive device according to the first embodiment; The speed controller 6 has adders 21 and 25 , a subtractor 22 , a speed feedback (Feed Back: FB) controller 23 , and a speed feed forward (Feed Forward: FF) controller 24 .

A speed command ω ₁ ^* is input to the speed controller 6 from the outside of the electric motor drive device 100 . The speed command ω ₁ ^* may be obtained by calculation inside the electric motor drive device 100 . A speed command ω ₁ ^* and a speed drooping amount Δω are input to the adder 21 . The adder 21 adds the speed command ω ₁ ^* and the speed droop amount Δω, and outputs a second speed command ω ₂ ^* that is the addition result. The speed droop amount Δω will be described later. The second speed command ω ₂ ^* and the rotational speed ω _e are input to the subtractor 22 . A subtractor 22 outputs the difference between the second speed command ω ₂ ^* and the rotational speed ω _e . The speed FB controller 23 performs FB control so that the difference input from the subtractor 22 becomes zero.

A proportional integral (PI) controller is used as the speed FB controller 23 . It is known that the PI controller has zero steady-state error for the step response. The use of the PI controller facilitates gain design. A controller based on a control law other than PI control may be used as the speed FB controller 23 . A controller having an integrator is used for the speed FB controller 23 in order to make the steady-state error zero. The speed FF controller 24 is connected in parallel with the speed FB controller 23 . A second speed command ω ₂ ^* is input to the speed FF controller 24 . A speed FF controller 24 performs FF control of the rotational speed _ωe . The speed controller 6 can speed up the control response by providing the speed FF controller 24 . Adder 25 adds the output value of speed FB controller 23 and the output value of speed FF controller 24 to generate q-axis current command i _q ^* .

The d-axis current command i _d ^* is determined by the flux-weakening controller 9 . The speed controller 6 may determine the _d -axis current command id ^* by "Maximum Torque Per Ampere control (MTPA)". The d-axis current command i _d ^* will be described later.

A dq-axis current controller 7 , which is a current controller, controls phase currents flowing through the synchronous motor 1 . As the dq-axis current controller 7, a vector controller that performs vector control on the dq rotating coordinates is used. A general vector controller performs current control on the dq rotation coordinates with the magnetic pole position _θe as a reference. When the phase current is converted to a value on the dq rotating coordinates, the alternating current amount becomes a direct current amount, which facilitates control. Therefore, the electric motor driving device 100 performs current control on the dq rotating coordinates. Since coordinate conversion requires information on the magnetic pole position θ _e , the dq-axis current controller 7 receives the magnetic pole position θ _e specified by the position/speed specifying unit 5 .

The dq-axis current controller 7 converts the value of the phase current into a d-axis current value and a q-axis current value, which are currents in the dq coordinate system, by coordinate transformation. Also, the dq-axis current controller 7 determines a voltage command 12 based on the d-axis current and the d-axis current command i _d ^* and the q-axis current and the q-axis current command i _q ^* . The dq-axis current controller 7 adjusts the d-axis voltage command so that the d-axis current matches the d-axis current command i _d ^* . The dq-axis current controller 7 adjusts the q-axis voltage command so that the q-axis current matches the q-axis current command _iq ^* . Thereby, the dq-axis current controller 7 determines a voltage command on the dq rotation coordinates.

The dq-axis current controller 7 includes a PI controller (not shown) that FB-controls the d-axis current, a PI controller (not shown) that FB-controls the q-axis current, and an FF compensation (not shown) for the dq-axis interference component. and a decoupling controller. If the d-axis current can appropriately follow the d-axis current command i _d ^* and the q-axis current can properly follow the q-axis current command i _q ^* , the control method in the dq-axis current controller 7 can be as described above. A method other than the method to be used may be used.

The dq-axis current controller 7 performs coordinate conversion from the voltage command on the dq rotating coordinates to the value on the three-phase stationary coordinates based on the magnetic pole position _θe . The dq-axis current controller 7 outputs a voltage command 12 on the three-phase stationary coordinates to the power converter 3 .

A voltage amplitude calculator 8 obtains a voltage amplitude that is the amplitude of the voltage command. The amplitude of the voltage command is also called the norm of the voltage command vector or the absolute value of the voltage command vector. Various methods are conceivable as methods for calculating the amplitude of the voltage command. The voltage amplitude computing unit 8 computes the amplitude of the voltage command, for example, by computation represented by the following equation (1). The voltage amplitude calculator 8 outputs the calculation result of the voltage amplitude.

|ν _dq ^* | is the voltage amplitude, ν _d ^* is the d-axis voltage command, and ν _q ^* is the q-axis voltage command. When the voltage amplitude calculator 8 performs the calculation of the formula (1), the voltage amplitude calculator 8 receives voltage commands ν _d ^* and ν _q ^* on the dq rotation coordinates from the dq-axis current controller 7 .

Note that the voltage amplitude calculator 8 may calculate the modulation factor instead of the voltage amplitude |ν _dq ^* |. The modulation factor is obtained by normalizing the voltage amplitude |ν _dq ^* | in order to evaluate how large the voltage amplitude |ν _dq ^* | is relative to the output limit of the power converter 3 . The voltage amplitude calculator 8 calculates M, which is the modulation factor, by the calculation shown in the following equation (2).

VDC is the _DC bus voltage of the voltage source inverter which is the power converter 3 . A DC bus voltage is detected by a DC bus voltage detector. Illustration of the DC bus voltage detector is omitted. A voltage region in which the modulation factor obtained by Equation (2) is less than 1 is called an inverter linear region. A voltage region where the modulation factor obtained by Equation (2) is greater than 1 is called an overmodulation region or a voltage saturation region.

The flux-weakening controller 9 determines the amplitude of the voltage output to the synchronous motor 1 by determining the d-axis current command i _d ^* based on the voltage amplitude |ν _dq ^* | and the first voltage limit value V _lim1 . Perform magnetic flux control to control The speed droop controller 10 controls the speed droop amount Δω based on the voltage amplitude |ν _dq ^* | and the second voltage limit value V _lim2 . Details of the flux-weakening controller 9 and the speed droop controller 10 will now be described.

FIG. 3 is a diagram for explaining a voltage vector representing a voltage state of the synchronous motor to be controlled by the motor drive device according to the first embodiment. FIG. 3 shows voltage vectors when the synchronous motor 1, which is an embedded permanent magnet synchronous motor, is rotating in a high speed region. Since the voltage drop due to the winding resistance of the synchronous motor 1 can be ignored in many cases in the high-speed region, the voltage drop due to the winding resistance is omitted in FIG. FIG. 3 shows voltage vectors in a steady state, omitting transient terms.

In the synchronous motor ₁ , the speed electromotive force _ωeΦa increases as the rotational speed _ωe increases. Here, _Φa is the number of dq-axis magnetic flux linkages, and is a value unique to the motor. The speed electromotive force ω _e Φ _a is generated in the q-axis direction. In a permanent magnet synchronous motor, the q-axis current is proportional to the magnet torque of the motor. The synchronous motor 1 normally outputs torque to cause the mechanical device 2 to do some mechanical work. A _q -axis current _iq flows through the synchronous motor 1, and an armature reaction of the _q -axis current _iq generates a voltage _ωeLqiq in the d-axis direction. L _q is the q-axis inductance.

On the other hand, since the d-axis current id has a low degree of contribution to torque, the _d -axis current _id is controlled to a smaller value than in the high speed range in the low and medium speed range where the rotational speed is lower than in the high speed range. Known techniques for determining the d-axis current command i _d ^* in the low-to-medium speed range include techniques such as i _d =0 control or MTPA.

In general, there is a limit to the maximum AC voltage that the power converter 3 can output to the synchronous motor 1 . In the high-speed region, the vector sum of the speed electromotive force ω _e Φ _a and the voltage ω _e L _q i _q may exceed the maximum output voltage of the power converter 3, and it is necessary to use a technique called flux-weakening control.

Assuming that the limit value of the dq-axis voltage is _Vom , the limit value _Vom satisfies the following approximate expression (3) in the high-speed region. Strictly speaking, the output limit range of the power converter 3 is a hexagonal shape, but is approximated by a circle here. In the first embodiment, the discussion is based on the premise of approximation with a circle, but it is needless to say that the discussion may be made strictly considering a hexagon.

In the first embodiment, a circle centered at the origin and having a radius limit value V _om is referred to as a voltage limit circle 30 . It is well known that when the power converter 3 is a PWM (Pulse Width Modulation) inverter, the limit value _Vom varies depending on the value of the DC bus voltage.

Since the speed electromotive force _ωeΦa becomes extremely large in the high-speed region, in order to increase the _q -axis current _iq , the _d -axis current id is caused to flow in the negative direction, and the amplitude of the voltage command vector ν ^* is set to the voltage limit circle Must be within 30. In this way, the control method of generating the _d -axis stator magnetic flux Ld id in the direction opposite to the _dq -axis magnetic flux linkage _Φa to reduce the amplitude of the voltage is generally called flux-weakening control. L _d is the d-axis inductance.

The simplest method for flux-weakening control is to determine the d-axis current command i _d ^* based on the voltage equation. By solving the above equation (3) for the _d -axis current id, the following equation (4) is obtained.

However, the flux-weakening control that obtains the _d -axis current id based on the above equation (4) has the drawback of being susceptible to changes in the motor constants or fluctuations in the motor constants, and is not widely used in the industrial world.

Integral-type flux-weakening control is known as one of techniques used instead of the flux-weakening control based on the above equation (4). For example, a method of determining the d-axis current command i _d ^* by integrally controlling the difference between the voltage amplitude |ν _dq ^* | and the first voltage limit value V _lim1 is known. In the following description, such a method may be referred to as "d-axis current command manipulation type flux-weakening control".

FIG. 4 is a block diagram showing a configuration example of a flux-weakening controller included in the electric motor drive device according to the first embodiment. The flux-weakening controller 9 has a subtractor 41 and an integrator 42 with a limiter. A subtractor 41 outputs a difference obtained by subtracting the voltage amplitude |ν _dq ^* | from the first voltage limit value V _lim1 . The integrator 42 obtains the d-axis current command i _d ^* by integrating the result of multiplying the difference by a control gain (not shown). The flux-weakening controller 9 is a controller that integrates the difference between the first voltage limit value V _lim1 and the voltage amplitude | ^ν _dq ^* | _. It can be automatically adjusted to an appropriate value that is neither too much nor too little.

When the voltage amplitude |ν _dq ^* | is greater than the first voltage limit value V _lim1 , the difference is negative, so the d-axis current command i _d ^* changes in the negative direction. Conversely, when the voltage amplitude |ν _dq ^* | is smaller than the first voltage limit value V _lim1 , the difference is positive, so the d-axis current command i _d ^* changes in the positive direction. In general terms, the d-axis current command i _d ^* is appropriately provided with a limiter. By providing the limiter, divergence of the integration calculation in the integrator 42 is prevented. In addition, demagnetization of the synchronous motor 1 due to excessive d-axis current command i _d ^* is prevented by providing a limiter. Further, a limiter in the positive direction may be provided to prevent the positive _d -axis current id from flowing while the synchronous motor 1 is rotating in the low-to-medium speed range. The limiting value in the positive direction is generally set to zero or "maximum torque/current command value for current control".

In order to explain the usefulness of the electric motor drive device 100 according to the first embodiment, another method widely known as a flux-weakening control method will be explained. The "position error command calculation", which is the method described in Patent Document 1, is considered to be a type of integral-type flux-weakening control. According to the above-described flux-weakening control method, the phase angle of the voltage command advances as a result of manipulating the d-axis current command, but the same effect can be obtained by directly manipulating the phase of the voltage command. can. A method of directly manipulating the phase of the voltage command is called "voltage phase control" or the like. It is presumed that the "voltage phase control" is also used in the "position error command calculation". There is also known a method of shifting the phase of the control coordinate in the forward direction with respect to the magnetic pole position instead of the phase of the voltage command. In the following description, these techniques using phase manipulation may be referred to as "phase manipulation type flux-weakening control". The phase-manipulation type flux-weakening control has the disadvantage that the mathematical outlook is poor and the calculation for determining the control gain is complicated.

In general, the poor mathematical outlook greatly affects the difficulty of control adjustment. Although the classical control engineering approach is a powerful tool for gain design, it does not work well when the plant model or controller contains nonlinear elements. Phase rotation requires trigonometric functions, and many differential equations containing trigonometric functions are nonlinear elements. If the phase manipulation amount is very small, it is possible to linearly approximate the trigonometric function. It is generally recognized that the discussion of nonlinear control is difficult, and control adjustment is not easy. If a suitable gain cannot be found theoretically, trial and error experiments are repeated to adjust the control gain, which requires a lot of labor. In such a point, the phase operation type flux-weakening control can be said to be an undesirable technique.

In the electric motor drive device 100 according to the first embodiment, the "d-axis current command operation type flux weakening control" makes it possible to easily perform gain design compared to the "phase operation type flux weakening control". The gain design in the "d-axis current command manipulation type flux-weakening control" will be described later.

FIG. 5 is a block diagram showing a configuration example of a speed droop controller included in the electric motor drive device according to the first embodiment. Here, a configuration will be described on the assumption that it is applied to an application such as a refrigerant compressor that performs only power running operation in forward rotation. The configuration of the speed droop controller 10 can of course be configured in consideration of reverse rotation or regenerative operation.

The speed droop controller 10 has a subtractor 51 and an integrator 52 with a limiter. A subtractor 51 outputs a difference obtained by subtracting the voltage amplitude |ν _dq ^* | from the second voltage limit value V _lim2 . The integrator 52 obtains the velocity droop amount Δω by integrating the result of multiplying the difference by a control gain (not shown). The speed droop controller 10 is a controller that integrates the difference between the second voltage limit value V _lim2 and the voltage amplitude |ν _dq ^* |. can be automatically adjusted to an appropriate value.

When the voltage amplitude |ν _dq ^* | is greater than the second voltage limit value V _lim2 , the difference is negative, so the velocity drooping amount Δω changes in the negative direction. Conversely, when the voltage amplitude |ν _dq ^* | is smaller than the second voltage limit value V _lim2 , the difference is positive, so the velocity drooping amount Δω changes in the positive direction. The integrator 52 limits the possible range of the velocity drooping amount Δω with a limiter so that the integral calculation does not diverge. By setting the upper limit value of the speed drooping amount Δω to zero, the electric motor drive device 100 can prevent the synchronous motor 1 from performing a decelerating operation under conditions where voltage saturation does not occur. That is, the speed droop controller 10 adjusts the speed droop amount Δω so that the voltage amplitude |ν _dq ^* | does not exceed the second voltage limit value V _lim2 . Thus, the velocity droop controller 10 determines an amount of velocity droop Δω that causes the voltage amplitude |ν _dq ^* | to be less than the second voltage limit value V _lim2 .

An appropriate value may be set for the lower limit value of the speed droop amount Δω. Here, since it is assumed that voltage saturation occurs in the high-speed region, for example, the lower limit value of the speed droop amount Δω is −10% to −20% of the maximum speed ω _Max of the synchronous motor 1. If the degree value is set, it is sufficient in many cases. As described above, in the power running operation in forward rotation, the possible range of the speed droop amount Δω is 0≧Δω≧−0.2ω _Max .

The speed controller 6 reduces the speed command ω ₁ ^* and determines the second speed command ω ₂ ^* based on the speed droop amount Δω thus obtained. When serious voltage saturation occurs, for example, when a load torque larger than the maximum torque that the synchronous motor 1 can output is applied to the synchronous motor 1, the motor drive device 100 outputs the speed command ω ₁ ^* . This reduces voltage saturation. By configuring the flux-weakening controller 9 and the speed droop controller 10 as described above, the gains of the flux-weakening controller 9 and the speed droop controller 10 can be designed very easily.

Next, referring to FIGS. 6 to 15, gain design in the electric motor drive device 100 will be described. FIG. 6 is a diagram illustrating an example of a control model for the motor drive device and the synchronous motor according to the first embodiment; FIG. 6 shows the details of the controller model of the motor drive device 100 and the electrical plant model of the synchronous motor 1 . Here, control design for specifically determining the flux-weakening control gain K _Ifw of the flux-weakening controller 9 and the velocity drooping gain K _lst of the velocity drooping controller 10 will be described.

FIG. 7 is a diagram showing a model obtained by approximating the control model shown in FIG. 6 around the operating point in the high-speed region. When it is determined that the control response of the dq-axis current controller 7 is sufficiently higher than the control responses of the speed controller 6, the flux-weakening controller 9, and the speed droop controller 10, the d-axis current command i _d ^* and the d-axis current i _d substantially match, and the q-axis current command i _q ^* and the q-axis current i _q can be regarded as substantially matching. It is also assumed that the rotation speed ω _e changes slowly near the operating point. Furthermore, it is assumed that the rotation speed ω _e is sufficiently high and that the voltage drop due to the armature resistance R is so small that it can be ignored. Under these conditions, the control model shown in FIG. 6 can be simplified and expressed as shown in FIG.

Here, the design of the flux-weakening control gain _KIfw , which is the control gain in the flux-weakening controller, will be described. FIG. 8 is a first diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. FIG. 9 is a second diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. FIG. 10 is a third diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG. FIG. 11 is a fourth diagram for explaining the design of the flux-weakening control gain in the flux-weakening controller shown in FIG.

By omitting the speed droop controller 10 and the speed controller 6 from the model shown in FIG. 7, the block diagram shown in FIG. 8 is obtained. Consider a transfer function for obtaining the voltage amplitude |ν _dq ^* | based on the first voltage limit value V _lim1 . Since the transfer function is a function expressed as one input and one output, input factors other than the first voltage limit value V _lim1 are assumed to be constant near the operating point. That is, the q-axis current command i _q ^* and the dq-axis magnetic flux linkage number Φa are ignored. Under these conditions, the block diagram shown in FIG. 9 is obtained from the block diagram shown in FIG.

The block diagram shown in FIG. 10 expresses a reference model of the flux-weakening controller 9 based on the block diagram shown in FIG. The flux-weakening controller 9 is desirably designed so that the voltage amplitude |ν _dq ^* | appropriately follows changes in the first voltage limit value V _lim1 . Also, it is desirable to specify the speed until the response converges using an arbitrary time constant. For this reason, the reference model of the flux-weakening controller 9 should be the first-order low-pass filter 60 . Let ω _fw be the cutoff angular frequency of the low-pass filter 60 . The cut-off angular frequency is the reciprocal of the time constant.

It is clear that the low-pass filter 60 shown in FIG. 10 is equivalent to the configuration shown in FIG. 11 by simple modification. A low-pass filter 60 shown in FIG. 11 has a subtractor 61 and an integrator 62 . By designing the flux-weakening control gain K _Ifw so that the open-loop transfer function in the block diagram shown in FIG. 9 and the open-loop transfer function in the block diagram shown in FIG. response characteristics can be obtained. Therefore, the flux-weakening control gain K _Ifw is determined by the following equation (5).

Next, the design of the speed droop control gain, which is the control gain in the speed droop controller 10, will be described. FIG. 12 is a first diagram for explaining the design of speed droop control gains in the speed droop controller shown in FIG. FIG. 13 is a second diagram illustrating the design of speed droop control gains in the speed droop controller shown in FIG. 14 is a third diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. 5. FIG. 15 is a fourth diagram for explaining the design of the speed droop control gain in the speed droop controller shown in FIG. 5. FIG. A speed droop control gain is determined based on the transfer function of the speed controller 6 and the transfer function of the synchronous motor 1 .

By omitting the flux-weakening controller 9 from the model shown in FIG. 7, the block diagram shown in FIG. 12 is obtained. Consider now the transfer function for obtaining the voltage amplitude |ν _dq ^* | based on the second voltage limit value V _lim2 . Since the transfer function is a function expressed as one input and one output, input elements other than the second voltage limit value V _lim2 are assumed to be constant near the operating point. That is, the d-axis current command i _d ^* and the dq-axis magnetic flux linkage number Φa are ignored. Under these conditions, the block diagram shown in FIG. 13 is obtained from the block diagram shown in FIG.

Furthermore, by modifying the block diagram shown in FIG. 13, the block diagram shown in FIG. 14 is obtained. The block diagram shown in FIG. 14 includes the transfer function of speed FB controller 23 . Here, the gain design of the velocity FB controller 23 will be described first.

As a method of designing the proportional gain _KPS of the velocity FB controller 23, for example, a method using the following equation (6) is known. As a method of designing the integral gain K _IS of the velocity FB controller 23, for example, a method using the following equation (7) is known.

J is the inertia, Pm is the pole logarithm, _ωSC is the speed control band, and _ωPI is the _PI breakpoint angular frequency. When ω _PI and ω _SC are determined with a policy of determining the target value response on the proportional control side and operating the integral control only to make the steady-state error zero, ω _PI is 1/5 or less of ω _SC . should be set to

The open-loop transfer function G _O (s) in the block diagram shown in FIG. 14 is represented by the following equation (8). Therefore, the block diagram shown in FIG. 14 can be transformed into the block diagram shown in FIG.

The closed-loop transfer function G _C (s) in the block diagram shown in FIG. 15 is represented by the following equation (9). In equation (9), the order of the complex number s of the transfer function parameter is two.

A general expression for the transfer function of the second-order delay system is expressed by the following expression (10). ζ is the damping coefficient and _ωn is the natural angular frequency.

By comparing the denominator coefficient in equation (9) and the denominator coefficient in equation (10), the velocity droop gain K _lst that makes the intrinsic angular velocity ω _n of the velocity droop controller 10 an arbitrary value is obtained as follows. can be determined by the following equation (11).

Note that the damping coefficient ζ of the speed droop controller 10 is expressed by the following equation (12).

If the damping coefficient ζ is not appropriate, the speed droop control by the speed droop controller 10 becomes unstable. If the damping coefficient ζ is less than 0.5, the fluctuation of the speed drooping amount Δω becomes remarkable until the speed drooping amount Δω converges. Therefore, it is desirable that the damping coefficient ζ is at least 0.5 or more. Since it is clear that ω _n >0 and ω _PI >0, the damping coefficient ζ always takes a positive value. Therefore, it can be said that the transfer function shown in the above equation (9) is stable.

Gain design is difficult in the case of a control configuration in which a flag is determined as to whether the voltage is saturated or not, and the speed command is lowered, as in Patent Document 1 described above. In contrast, according to the first embodiment, the control system configured as shown in FIG. 1 enables clear gain design as described above.

FIG. 16 is a diagram showing an example of operating waveforms when the electric motor drive device according to the first embodiment is used. FIG. 16 graphically shows an example of the relationship between each of the rotation speed ω _e , load torque T, _d -axis current command id ^* , voltage amplitude |ν _dq ^* |, and speed drooping amount Δω and time. there is

Assume that the load torque T is gradually increased from time t1 to time t5 as shown in FIG. 16 while the synchronous motor 1 is rotating at a constant speed. During the period up to time t2, the d-axis current command i _d ^* becomes zero because the voltage amplitude |ν _dq ^* | is smaller than the first voltage limit value V _lim1 . After time t2, the voltage amplitude |ν _dq ^* | exceeds the first voltage limit value V _lim1 . Then, the integral-type flux-weakening controller 9 increases the d-axis current command i _d ^* in the negative direction so that the voltage amplitude |ν _dq ^* | does not increase any further.

Assume that the _d -axis current command id ^* reaches the lower limit value _IdLimL at time t3 by increasing the _d -axis current command id ^* in the negative direction. The lower limit value _IdLimL is set to protect the synchronous motor 1 from demagnetization, heat generation, and the like. A _d -axis current id exceeding the lower limit value _IdLimL cannot flow through the synchronous motor 1 . Therefore, after time t3, it becomes necessary to reduce the speed command ω ₁ ^* in order to alleviate the voltage saturation.

In FIG. 16, the second voltage limit value _Vlim2 is set to a value higher than the first voltage limit value _Vlim1 . The voltage amplitude |ν _dq ^* | increases during the period from time t3 to time t4, but at time t4, the voltage amplitude |ν _dq ^* | reaches the second voltage limit value V _lim2 and the speed droops. A quantity Δω begins to occur. During the period from time t4 to time t5, the rotation speed ω _e decreases due to the occurrence of the speed drooping amount Δω, and the voltage amplitude |ν _dq ^* | stops increasing. After time t5, the load torque T becomes constant, and the rotation speed ω _e stops decreasing.

In Embodiment 1, the first voltage limit value V _lim1 and the second voltage limit value V _lim2 are provided separately, and the second voltage limit value V _lim2 is equal to the first voltage limit value V _lim1 , the electric motor drive device 100 shifts the operation timing of the flux-weakening control and the operation timing of the speed droop control from each other. As a result, the electric motor drive device 100 can maximize the use of the flux-weakening control to increase the output torque of the synchronous motor 1 .

It should be noted that when the overmodulation region of the power converter 3 is utilized to increase the maximum torque and reduce the copper loss, the first voltage limit value V _lim1 and the second 2 voltage limits _{V_lim2} can be set. As a result, the motor driving device 100 can fully utilize the output limit range of the synchronous motor 1 by controlling the speed drooping amount Δω after the voltage amplitude limitation by the flux-weakening control becomes ineffective.

According to Embodiment 1, the electric motor drive device 100 can suppress the phenomenon that the control of the synchronous motor 1 becomes unstable at the time of voltage saturation without performing complicated work for control adjustment. In applications such as refrigerant compressors, labor saving in control adjustment is a great advantage. Furthermore, the electric motor driving device 100 can utilize the overmodulation region of the power converter 3 to increase the maximum torque and reduce the copper loss. As described above, the electric motor drive device 100 has the effect of reducing the workload required for adjustment for stably controlling the electric motor.

Embodiment 2.
In the second embodiment, the hardware configuration of the electric motor drive device 100 will be described. 17 is a diagram illustrating an example of a hardware configuration of an electric motor drive device according to Embodiment 2 of the present invention; FIG. In the second embodiment, the same reference numerals are given to the same components as in the first embodiment. FIG. 17 shows the synchronous motor 1, the power converter 3, the current detector 4, and the mechanical device 2 that operates using the synchronous motor 1 as a power source together with the motor driving device 100, which constitute the motor system. .

The electric motor driving device 100 has a processor 101 and a memory 102 as a hardware configuration. The functions of the position/speed specifying unit 5, the speed controller 6, the dq-axis current controller 7, the voltage amplitude calculator 8, the flux-weakening controller 9, and the speed droop controller 10 shown in FIG. It is implemented by the processor 101 executing a program stored in the memory 102 .

The processor 101 is a CPU (Central Processing Unit), processing device, arithmetic device, microprocessor, microcomputer, or DSP (Digital Signal Processor). The memory 102 includes volatile storage such as random access memory and non-volatile secondary storage such as flash memory. The memory 102 may have an auxiliary storage device such as a hard disk instead of the non-volatile auxiliary storage device. Illustrations of the volatile storage device and the auxiliary storage device are omitted. Processor 101 reads the program stored in the auxiliary storage device via the volatile storage device. The processor 101 outputs data such as calculation results to the volatile storage device. Processor 101 may store data in secondary memory via volatile memory.

Various methods have been studied for the power converter 3 and the current detection unit 4, but basically any method may be used. The electric motor system may be provided with voltage detection means for detecting the input voltage or output voltage of the power converter 3 or voltage detection means for detecting the DC bus voltage.

Basically, any method can be used for transmitting and receiving data between each component. Each component may transmit and receive digital signals or transmit and receive analog signals. Digital signals may be used for parallel communication or serial communication. Analog signals and digital signals may be converted as appropriate by a converter (not shown). For example, when the phase current detected by the current detector 4 is represented by an analog signal, a D/A (Digital to Analog) converter (not shown) converts the analog signal into a digital signal and transmits the data to the processor 101 . The D/A converter (not shown) may be inside the motor drive device 100 or inside the current detection unit 4 .

The voltage command signal that the processor 101 transmits to the power converter 3 may be either an analog signal or a digital signal. Moreover, the processor 101 may have a modulating section such as a carrier comparison modulating section and a space vector modulating section. Processor 101 may transmit a voltage command, which is a pulse train after modulation, to power converter 3 . When voltage detection means for detecting the input voltage or output voltage of the power converter 3 or voltage detection means for detecting the DC bus voltage is provided, the transmission/reception method between the voltage detection means and the motor drive device 100 is basically Any method may be used. When a position sensor is attached to the synchronous motor 1, any transmission/reception method may be used between the position sensor and the motor drive device 100 basically.

The processor 101 determines the voltage command 12 by performing speed control calculation and current control calculation based on the speed command ω ₁ ^* . If the amplitude of the voltage command 12 exceeds the first voltage limit value _Vlim1 , the flux weakening control operates, and if the amplitude of the voltage command 12 exceeds the second voltage limit value _Vlim2 , the speed droop control operates.

Speed command ω ₁ ^* , first voltage limit value V _lim1 , and second voltage limit value V _lim2 are given to motor drive device 100 from a computer external to motor drive device 100 . The illustration of the computer that gives the speed command ω ₁ ^* , the first voltage limit value V _lim1 and the second voltage limit value V _lim2 to the motor drive device 100 is omitted. The speed command ω ₁ ^* , the first voltage limit V _lim1 and the second voltage limit V _lim2 may be calculated inside the processor 101 . Depending on the computing power of the processor 101, the processor 101 may perform computational processes other than computing the speed command ω1 ^* , the _first voltage limit value _Vlim1 , and the second voltage limit value _Vlim2 .

The configuration shown in the above embodiment shows an example of the content of the present invention, and it is possible to combine it with another known technology, and one configuration can be used without departing from the scope of the present invention. It is also possible to omit or change the part.

1 synchronous motor 2 mechanical device 3 electric power converter 4 current detector 5 position speed specifying unit 6 speed controller 7 dq-axis current controller 8 voltage amplitude calculator 9 flux-weakening controller 10 speed Droop controller 11 Signal 12 Voltage command 21,25 Adder 22,41,51,61 Subtractor 23 Speed FB controller 24 Speed FF controller 30 Voltage limit circle 42,52,62 Integral instrument, 60 low-pass filter, 100 motor drive, 101 processor, 102 memory.

Claims (5)

The value of the phase current flowing in the motor is converted into each value of d-axis current and q-axis current, which are currents in the dq coordinate system, and the d-axis current and d-axis current command and the q-axis current and q-axis current command are converted into a current controller that controls the phase current by determining a voltage command based on
a voltage amplitude calculator that obtains a voltage amplitude that is the amplitude of the voltage command;
a speed controller that controls the rotational speed by determining the q-axis current command based on the speed command, the rotational speed of the electric motor, and a speed droop amount that reduces the speed command;
a flux weakening controller that performs flux control for limiting the amplitude of the voltage output to the electric motor by determining the d-axis current command based on the voltage amplitude and a first voltage limit value;
a speed droop controller that controls the speed droop amount based on the voltage amplitude and a second voltage limit value;
The motor drive device, wherein the speed droop controller determines the speed droop amount that makes the voltage amplitude smaller than the second voltage limit value. 2. The motor drive device according to claim 1, wherein said second voltage limit value is a value greater than said first voltage limit value. 3. The motor driving device according to claim 1, wherein the speed droop controller is a controller that integrates a difference between the second voltage limit value and the voltage amplitude. 4. The motor drive device according to claim 1, wherein the flux-weakening controller is a controller that integrates a difference between the first voltage limit value and the voltage amplitude. 5. The electric motor drive according to claim 1, wherein a control gain in said speed droop controller is determined based on a transfer function of said speed controller and a transfer function of said electric motor. Device.

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