RU2608554C2 - High-speed decametric radio communication system - Google Patents

Abstract

FIELD: wireless communications.

SUBSTANCE: invention relates to radio communication and can be used in wide application decameter range radio networks, intended for discrete messages high-speed transmission using angular keying signals. System of high-speed decametric radio communication contains transmitting system, including series-connected messages source, encoder and series-parallel converter, which outputs are connected to corresponding inputs of N channel keying unit, which output of is connected to series-connected radio transmitting device and transmitting antenna, as well as receiving system containing two receiving antennae, each of which output is connected to input of corresponding radio receiving device, as well as N channel demodulators unit, which outputs are connected to corresponding inputs of parallel-serial converter, which output of is connected with in-series connected decoder and messages recipient. Receiving system includes N coherent signals addition (CSA) units, one input of each of which is integrated with one radio receiving device output, and of each CSA other input is combined with other radio receiving device output, wherein of each CSA output is connected corresponding input of N channel demodulators unit, each CSA comprises two phasing units, each of which contains serially connected channel filter, which input is corresponding input of CSA, normalizing amplifier, first multiplier, measuring filter and second multiplier, which other input is connected to first multiplier input, each phasing unit second multiplier output is connected with corresponding input of adder, which output is connected to input of resulting signal filter, which output, which is CSA output, is connected through resultant signal normalizing amplifier to other input of each phasing unit first multiplier.

EFFECT: technical result is each i-th channel signal (i=1, 2,…, N) higher noise immunity of receiving consisting of N-channel group signal and group signal in general under influence of additive radio interferences, as well as communication system broader functional capabilities due to enabling of signals transmission and reception with any angular keying of bearing channel signals both in phase (PDSK, DPDSK, etc), and in frequency (FT, DFT).

1 cl, 1 dwg, 1 tbl

Description

The invention relates to radio communications and can be used in decameter radio networks of wide application, intended for the transmission of high-speed discrete messages using signals with angular manipulation.

A known system of high-speed decameter radio communication with single-channel (sequential) transmission of discrete messages containing a transmitting complex containing a series-connected encoder, modulator, radio transmitting device and a transmitting antenna, as well as a receiving complex containing a series-connected receiving antenna, a radio receiving device, a demodulator and a decoder [1 ], from. 107.

In this system, the data stream from the output of the encoder manipulates a single carrier frequency in the modulator. Depending on the multiplicity of compression of the transmitted signal k [2], p. 573, the modulator can generate signals with angular manipulation such as, for example, signals of single relative phase telegraphy (OFT) or frequency telegraphy (CT) signals with a data rate of V = 1 / T (bit / s) for a selected duration T of the transmitted signal element and at k = 1, or signals of a double OFP (DFT) or double frequency telegraphy (DBT) transmitted with a double speed V = 2 / Τ (bit / s) at k = 2, etc.

Demodulation of these signals can be carried out by traditional methods given in [2] and determining the structure of constructing a demodulator that implements the reconstruction operation of the transmitted binary sequence, the inverse of the modulation operation.

However, single-channel (sequential) decameter radio communication systems with traditional methods of processing received signals [2] and more complex methods of processing signals [3] have the following disadvantages:

1. When using traditional methods of demodulating received signals [2], data transmission over a decameter radio channel with high speed is associated with serious difficulties due to the occurrence of intersymbol interference on the receiving side due to the presence of delayed beams. If you do not take special measures (to reduce or even eliminate the harmful effect of the multipath effect), then the duration Τ of the transmission (signal element) cannot be selected less than 2-3 ms, which limits the maximum transmission speed of about 300-500 bits / s [3].

2. An increase in the data transfer rate by increasing the multiplexing factor k while maintaining the required duration длительн of the element of the transmitted signal leads to a decrease in the noise immunity of receiving discrete information [2].

3. Increasing the data transfer rate by reducing the duration Τ of the element of the transmitted signal when implementing more complex algorithms for processing the received signals, allowing to overcome the consequences of intersymbol interference, for example, as in a communication system with a test pulse and prediction (SIIP) [3], leads to a decrease in noise immunity reception due to a decrease in the energy of the transmitted signal element [2] and the deterioration of the electromagnetic compatibility of the radio communication system due to the expansion of the transmission spectrum emogo signal.

In addition, the expansion of the spectrum of the transmitted signal requires a corresponding increase in the bandwidth when receiving this signal, which further reduces the noise immunity of the reception due to the increased likelihood that the spectral components of the station or spectrum-focused (interference) radio interference get into the wider reception band.

A known system of high-speed decameter radio communication with multi-channel (parallel) transmission of discrete messages containing a transmitting complex containing a series-connected message source, encoder and series-parallel converter, the outputs of which are connected to the corresponding inputs of the block N channel manipulators, the output of which is connected to series-connected radio transmitting device and a transmitting antenna, as well as a receiving complex containing serially connected receiving an antenna, a radio receiver and a block of N channel demodulators, the outputs of which are connected to the corresponding inputs of the parallel-serial converter, the output of which is connected to the decoder and the receiver of messages in series [4].

In transmitting complex of this radio communication system the transmitted binary sequence output from the encoder with a speed V = 1 / T (bit / s) converted to serial-to-parallel converter in kN parallel subsequences with channel rate following binary symbols in each V _CAD = V / N (bits / from).

In the block of N channel manipulators, consisting of N identical channel manipulators, for example, phase or frequency, for each manipulator at the initial moment of each clock interval of duration T _channel = TN, k symbols of the corresponding k subsequences are fed in parallel and synchronously. For example, when sealing of multiplicity k = 1 and for a selected duration T _ch = TN element of the transmitted signal, each channel manipulator can be formed on corresponding to this channel manipulator carrier frequency signals GSF or Th at the transfer rate V _ch = 1 / TN (bit / s), when k = 2 - signals DOFT or DCT with a speed of V _channel = 2 / TN (bit / s), etc.

The output signals of all channel manipulators are summed up in a block of N channel manipulators and a group N-channel (frequency) signal is transmitted to the air using a radio transmitting device and a transmitting antenna.

In the receiving complex of this radio communication system, the received group signal from the output of the radio receiving device is supplied to a block of N channel demodulators, consisting of N of the same type of demodulators, each of which demodulates the corresponding channel signal by the selected traditional method [2]. As a result, at the outputs of a block of N channel demodulators, kN binary subsequences are formed in parallel, which are converted by a parallel-serial converter into one binary sequence, similar to the sequence at the encoder output, which, after decoding in the decoder, is sent to the information receiver.

The duration T of the transmitted binary waveform _kan element in each channel becomes N times greater than the original duration Τ signal on the encoder output element that allows to overcome the negative effects of multipath reception signal at a relatively high data rate.

However, the noise immunity of this decameter radio communication system is insufficient:

1. Increasing the data rate by increasing the multiplicity of seals in each channel while maintaining the necessary duration T _kan element of the transmitted signal reduces the noise immunity of receiving digital information [2].

2. Increasing the data rate by reducing the duration T of the transmitted signal _kan element in each frequency channel or by increasing the number N of frequency channels results in a corresponding expansion of the spectrum of the transmitted baseband signal and the corresponding bandwidth extension at its reception, which leads, as mentioned above, to reduce the noise immunity of receiving discrete information and the deterioration of the electromagnetic compatibility of the communication system.

Of the known high-speed decameter radio communication systems, the closest in essence the tasks to be solved and to the majority of the essential features coinciding is the high-speed decameter radio communication system given in [5], p. 7.

The structure of this radio communication system with multichannel (parallel) transmission of discrete messages basically corresponds to the above communication system [4], except that in the receiving complex of this system not only single reception of signals to one antenna can be carried out, but also more noise-resistant dual reception into two antennas that are spaced apart in space or polarization when a group multi-frequency signal is emitted in a transmitting complex in the upper or lower sideband, as well as dual reception for the same frequency diversity antenna chlorophylls two baseband boot both sidebands of the radio transmission device in the same signal group [5], p. 10.

Consider in more detail the operation of the receiving complex of this system.

Two samples of the group signal received at the separated antennas from the outputs of the linear paths of the respective two radio receivers of the receiving complex of this communication system are simultaneously fed to each of the N = 20 channel blocks (KB), which filter out each of the samples of the group signal to its channel signals and calculate values proportional to the cosines and sines of the phase difference between the received sample channel signal and the corresponding reference oscillation.

Each channel block provides dual reception both when the antennas are separated in space or polarization, and when the frequency diversity, and consists of two identical active filters, which receive samples of the group signal from the outputs of the linear paths of the radio receivers and the same reference oscillation from the generator frequency grids.

Each of the 2N active filters, in turn, consists of two identical correlators, characterized in that the reference oscillations supplied to them are 90 ° out of phase. The correlator contains a multiplier and an integrator, built on the basis of a DC amplifier with a large gain and an RC feedback circuit.

The output signals of the integrators of each channel block are the results of converting the corresponding two samples of the channel signal to zero frequency with the decomposition of each channel signal into two quadrature components, the voltage values of which are recorded and stored in the corresponding cells of the storage device (memory). Information on the quadrature components of each package of each channel signal in the form of analog voltage levels is stored in memory cells (on capacitors) for the duration of two adjacent packages. Moreover, from the values of the levels of each pair of quadrature components, one can calculate the amplitude and phase of the carrier wave of the corresponding channel signal.

In the subsequent phase difference calculation unit (BVRF), each of the N pairs of samples of channel signals received on the corresponding two spaced apart, for example, in the antenna space, is linearly added by summing the corresponding analog levels of the quadrature components recorded in the memory.

Since the radio communication system under consideration uses phase difference manipulation (relative phase telegraphy) signals with a multiplicity of k = 1, or k = 2, or k = 3 to transmit information, to determine the true values of binary symbols upon demodulation of each of the N resulting channel signals (after linear addition), it is required to determine the phase difference between every two time-adjacent bursts of the resulting channel signal. In BVRF, this operation is performed by calculating the values of trigonometric functions according to the data recorded in the memory.

The output binary information, depending on the multiplicity of multiplexing k, is output from the output (s) of each channel demodulator to the input (s) of the subsequent output device (WU) via one, two, or three buses (outputs), i.e. by the number of binary subchannels (corresponding to multiplicity of multiplexing k) in one channel of the system [5], p. 18. The control unit is intended for folding binary information arriving at it via kN outputs into one binary sequence similar to that transmitted.

Thus, in the BVRF, each of N pairs of identical samples of channel signals is linearly added and the phase difference of the signal sendings across all channels is calculated for all manipulation rates, i.e. in fact, the BVRF performs the function of a block of N channel signal demodulators, which, unlike the block of N channel demodulators of the above system [4], provides demodulation of each result of linear addition of two identical samples of a channel signal, the signal only with phase difference manipulation.

WU here actually performs the function of a parallel-serial converter, as in the above radio communication system [4].

The dual diversity reception is an effective means of increasing the noise immunity in radio channels with fading signals [2] and in this case to some extent compensates for the reduction in noise immunity of the reception due to an increase in the data transfer rate.

However, the noise immunity of this decameter radio communication system is insufficient, since it realizes a linear addition of the output levels of the quadrature correlators of channel signals, which are actually phase detectors of the received signals, i.e. in accordance with the terminology given in [6], Section 5.3 implements a linear post-detector addition of each of the N pairs of samples of channel signals of a group N-channel signal received along two diversity branches.

It is also known from [6] that, with diversity reception, linear signal addition provides a smaller gain in noise immunity (in the signal-to-noise ratio in power) with respect to the optimal addition of these same signals. In addition, when implementing optimal or linear addition, it is preferable to perform addition before detecting the signals, since when in-phase addition of oscillations, the signals are added algebraically, while the noise is added geometrically [6], p. 183.

The disadvantage of linear addition is that its implementation puts forward stringent requirements for ensuring equal gain in the diversity branches. A significant difference in the gain in the branches in the limit turns the dual reception into a single one. The allowable spread of gain should provide a spread of no more than 1-2 dB in the entire dynamic range of the linear receiving path, taking into account the influence on the characteristics of the devices destabilizing factors [6], p. 180.

Moreover, a significant drawback of linear addition in this case is the insecurity of each result of the addition of two samples of any channel signal from exposure to at least one of the radio receiving devices of the diversity diversity branches of an additive spectrum-focused (impact) interference, the occupied frequency band of which is part of the frequency band or fully coincides with the frequency band occupied by any channel signal.

In this case, the interference voltage will be linearly added (without reducing (suppressing) its level) with the voltages of the channel signal samples, distorting the result of the summation. When the interference voltage reaches a level commensurate with the resulting level of the summed samples of the channel signal, the demodulation of the summation result by the corresponding channel demodulator can be blocked, i.e. when erroneously received symbols start appearing in the regenerated binary sequence at the output of the channel demodulator (for k = 1) or in each of k regenerated binary subsequences on each of the k outputs of the channel demodulator (for k> 1).

As a result, if one of the N channel signals is concentrated by the interference spectrum concentrated by the spectrum, the output binary sequence at the output of the parallel-serial converter (to the decoder) may be distorted, for example, for k = 1, each N-th binary symbol. When N = 20, error probability (before the decoder) could reach value P _OJJJ ≥0,05, which may be critical for correcting capability of the selected correction code communication system and unacceptable for the recipient information (after the decoder).

Under the influence of two or more concentrated interference of a certain level, radio communication obviously becomes unsuitable.

Under the influence of a comparatively broadband spectrum of additive interference, for example, station interference, the occupied frequency band of which covers the frequency band occupied by more than one channel signal, then when the interference reaches a certain level, two or more parallel channels may be affected, which is also obviously not suitable for communication .

The disadvantage is that in the communication system [5] it is possible to transmit and receive only signals with phase difference modulation at k = 1, k = 2, k = 3, which limits its functionality.

The problems the present invention is directed to, a high-speed decameter radio communication system, is to increase the noise immunity of reception of each i-th channel signal (i = 1, 2, ..., Ν) as part of the N-channel group signal and the group signal as a whole when exposed to additive radio interference.

In addition, an additional object of the invention is to expand the functionality of the communication system by providing transmission and reception of signals with any angular manipulation of the carrier channel signals both in phase (OFT, DOPT, etc.) and frequency (CT, DCT).

The solution of these problems is achieved by the fact that in a high-speed decameter radio communication system containing a transmitting complex containing a serially connected message source, an encoder and a serial-parallel preamplifier, the outputs of which are connected to the corresponding inputs of the block N channel manipulators, the output of which is connected to the radio-transmitting device a transmitting antenna, as well as a receiving complex containing two receiving antennas, the output of each of which is connected to the input the corresponding radio receiver, and a block of N channel demodulators, the outputs of which are connected to the corresponding inputs of the parallel-serial converter, the output of which is connected to the decoder and the receiver of messages in series, N blocks of coherent signal addition (BCS) are introduced, one input of each of which is combined with the output of one radio receiver, and the other input of each BCS is combined with the output of another radio receiver, and the output of each BCS is connected to the corresponding the input input of the block of N channel demodulators, each BCS contains two phasing nodes, each of which contains a channel filter connected in series, the input of which is a corresponding input of the BCS, a normalizing amplifier, a first multiplier, a measuring filter and a second multiplier, the other input of which is connected to the input of the first multiplier , the output of the second multiplier of each phasing unit is connected to the corresponding input of the adder, the output of which is connected to the input of the filter of the resulting signal, the output of which The horn, which is the output of the BCS, is connected through the normalizing amplifier of the resulting signal to another input of the first multiplier of each phasing unit.

In FIG. 1 is an electrical structural diagram of the proposed radio communication system.

The high-speed decameter radio communication system comprises a transmitting complex 1, comprising a message source 2 connected in series, an encoder 3 and a serial-parallel converter 4, the outputs of which are connected to the corresponding inputs of the channel manipulator unit N, the output of which is connected to a radio-transmitting device 6 and a transmitting antenna 7 connected in series , as well as a receiving complex 8, containing two receiving antennas 9 ₁ and 9 ₂ , the output of each of which is connected to the input of the corresponding radio receiver devices 10 ₁ (10 ₂ ), and a block N of channel demodulators 11, the outputs of which are connected to the corresponding inputs of the parallel-serial converter 12, the output of which is connected to the serial connected decoder 13 and the message receiver 14.

One input of each BCS 15 ₁ ... 15 _{Ν is} combined with the output of one radio receiver 10 ₁ , and the other input of each BCS 15 ₁ ... 15 _{N is} combined with the output of another radio receiver 10 ₂ , and the output of each BCS 15 ₁ ... 15 _{N is} connected to the corresponding input block N channel demodulators 11.

Each BCS 15 ₁ ... 15 _N contains two phasing nodes 16 ₁ and 16 ₂ , each of which contains a channel filter 17 connected in series, the input of which is a corresponding input of BCS, a normalizing amplifier 18, a first multiplier 19, a measuring filter 20 and a second multiplier 21, the other input of which is connected to the input of the first multiplier 19.

The output of the second multiplier 21 of each phasing unit 16 ₁ (16 ₂ ) is connected to the corresponding input of the adder 22, the output of which is connected to the input of the filter of the resulting signal 23, the output of which, which is the output of the BCS, is connected through the normalizing amplifier of the resulting signal 24 to another input of the first multiplier 19 each phasing node is 16 ₁ (16 ₂ ).

The high-speed decameter radio communication system operates as follows.

In the transmitting complex 1, the binary sequence from the output of the message source 2 enters the encoder 3, the task of which is to increase the noise immunity of data transmission. Coding, as a rule, is accompanied by two effective procedures - scrambling and interleaving. Scrambling converts a digital signal into a quasi-random one in order to obtain a more uniform energy spectrum of the emitted radio signal. Simple interleaving (temporal permutation) of message symbols allows to decorrelate errors in the channel, i.e. convert long-duration error packets into a series of single ones. The last operation significantly increases the coding efficiency [4].

The binary sequence output from the encoder 3 at a speed V = 1 / T (bits / s), where Τ - duration of the signal element is converted to a serial-parallel converter 4 kN parallel subsequences with channel rate sequence of binary symbols each equal to V _CAD = V / N (bits / s). Here, similarly to the above, N is the number of parallel orthogonal channel signals in the transmitted group signal, k is the multiplicity of compression of each channel signal.

In the block of N channel manipulators 5, consisting of N of the same type of channel manipulators, for example, phase or frequency, for each manipulator at the initial instant of time of each clock interval of duration Τ _kan = TN, k symbols of corresponding k subsequences are sent in parallel and synchronously. For example, when k = 1, each channel manipulator may generate signals GSF or Th with channel V _CAD rate = 1 / TN (bit / s), when k = 2 - signals DOFT or DCHT with velocity V _CAD = 2 / TN (bits / s) etc.

The output signals of all the manipulators are summed up in the block of N channel manipulators 5, and the group N-channel (frequency) signal is radiated using the radio transmitting device 6 and the transmitting antenna 7.

In the reception complex 8 taken at spaced antenna ₉ January and _September 2nd baseband samples and the additive noise with the linear paths of the outputs of two radio receivers (RSUH) 10 ₁ and 10 _2, simultaneously supplied to the respective two inputs of each of the N BCS 15 ₁ ... 15 _Ν. In each i-th BCS 15 _i with a conditional serial number i (i = 1 ... N), from each of the two received samples of the group signal, a corresponding channel signal sample with serial number i (at the channel frequency with serial number i) is extracted using one of two identical channel filters 17 phasing units 16 ₁ and 16 ₂ . The bandwidth of each channel filter is consistent with the spectrum of the allocated channel signal.

The filtered samples of the ith channel signal and additive interference are leveled by normalizing amplifiers 18 and then fed to the first multipliers 19 and second multipliers 21. After adjusting the phase (phasing) of the signals and weighing them in the second multipliers 21 by multiplying the normalized total signal voltage and noise on the voltage from the output of the measuring filter corresponding to the level ("weight") of the signal in the normalized mixture of the signal and noise, voltage from the outputs of the second multiplier nodes phased 16 ₁ and 16 ₂ are fed to the corresponding inputs of the adder 22.

The summation result is filtered by the filter of the resulting signal 23 and fed to the input of the corresponding channel demodulator of the channel demodulator unit 11 and to the input of the normalizing amplifier of the resulting signal 24, from the output of which the resulting oscillation is fed to the second inputs of the first multipliers 19.

Demodulation of each i-th resulting channel signal with angular manipulation from the output of the i-th BCS can be carried out by the corresponding channel demodulator of the channel demodulator block 11 using one of the known methods [2, 6].

As a result, at the outputs of the block N channel demodulators 11, kN binary subsequences are formed in parallel, which are converted by the parallel-serial converter 12 into one binary sequence similar to that transmitted from the output of encoder 3, which, after decoding in decoder 13, is then transmitted to the information receiver 14.

To conduct a comparative assessment of the noise immunity of the proposed communication system and the known communication system [5], the prototype, we consider in more detail the operation of the i-th BCS 15 _i , which ensures optimal coherent addition of two samples of the corresponding i-th channel signal as in the absence of interference at its inputs, and when the appearance of interference concentrated in the spectrum at one of its inputs.

In order to simplify analysis operation BCS shall consider its work within any of the time intervals Δt stationarity reception baseband signal, much longer than the duration of a binary element at the respective node input signal T _kan within which the amplitude of each sample i-th channel signal phasing January ₁₆ (16 ₂ ) can be considered constant.

In addition, at the first stage of the analysis, we assume that, when receiving signal samples, additive spectrum-concentrated spectral interference in the group signal reception band of each diversity branch is absent, and the level of fluctuation noise at the output of the linear path of each RPU 10 ₁ and 10 _{2 is} relatively small with respect to signal level and reception quality is practically not affected.

Let the samples of a group signal come from the outputs of the linear paths of the RPU 10 ₁ and 10 ₂ to the first and second inputs of the BCS 15 _i :

- at the first entrance -

- to the second entrance -

Here U _1Gy (t) and U _2Gy (t) are the voltages of the first and second samples of the group signal, respectively;

U _1Ci (t) is the voltage of the first sample of the i-th channel signal;

U _2Ci (t) is the voltage of the second sample of the i-th channel signal;

U _1Ci and ϕ _{1Ci are the} amplitude and phase of the first sample of the i-th channel signal;

U _2Ci and ϕ _{2Ci are the} amplitude and phase of the second sample of the i-th channel signal;

ω _Ci is the carrier frequency of the i-th channel signal;

θ _Ci (t) is a function that determines the type of angular manipulation [7] of the i-th channel signal;

N is the number of channel signals in the received group signal.

To simplify the analysis, we assume that the transfer coefficient of any of the filters (17, 20, 24), as well as the adder 22 in each BCS 15 ₁ ... 15 _N is equal to one. In addition, due to the fact that the structure of the BCS is a closed loop system of self-regulation with feedback, signal delays or changes in their initial phases when passing through these BCS filters will not be taken into account.

In view of the foregoing, samples of the i-th channel signal at the inputs of the normalizing amplifiers 18 of the i-th BCS filtered by channel filters 17 phasing nodes 16 ₁ and 16 ₂ can be represented as follows:

- for node 16 ₁ -

- for node 16 ₂ -

It should be noted that in the proposed communication system, RPUs 10 ₁ and 10 ₂ should operate in the off mode of their own automatic gain control (AGC) system, since the AGC of the radio receiver can only adjust the level of the group signal received in the corresponding wide band, and not each channel signal with a reception band in N of a smaller reception band of the group signal.

The AGC system of each normalizing amplifier 18 and 24 BCS 15 ₁ ... 15 _N can be characterized by the coefficient of regulation of the AGC system. The AGC control coefficient shows how many times the range of the signal at the output of the normalizing amplifier is less than at its input [6]:

where U _{IN MIN} and U _{OUT MIN} are the minimum input and minimum output voltages, which are limited by the value of the real sensitivity of the normalizing amplifier 18 of the phasing unit 16 ₁ (16 ₂ ), and U _{IN MAN} and U _{OUT MAN} are limited by the maximum value of input oscillations at which the level of combination components at the output of the normalizing amplifier 18 does not exceed the permissible.

For each of the identical normalizing amplifiers 18 and 24, BCS 15 ₁ ... l5 _N will be considered acceptable, for example, a 100 dB change in the signal at the input of the normalizing amplifier filtered by the corresponding channel filter by 100 dB when the signal at its output changes by no more than 3 dB. AGC systems with such parameters are implemented in modern RPUs [8].

At the output of the normalizing amplifiers 18 of each phasing unit 16 ₁ and 16 _{2, the} filtered samples of the i-th channel signal are leveled and fed to the inputs of the first multipliers 19, the other inputs of which come from the normalizing amplifier of the resulting signal 24, the resulting oscillation:

where U _Pi , ω _Pi , ϕ _Pi are the amplitude, angular frequency, and phase of the resulting signal, respectively.

The output product of the first multiplier 19 of the first phasing unit 16 ₁ , to one input of which the filtered and normalized voltage of the first sample of the i-th channel signal is supplied, and to its other input, the resulting oscillation, can be represented as:

where K ₁ is the value of the transfer coefficient of the normalizing amplifier 18 of the first phasing unit 16 ₁ , at which normalization of the first sample of the input signal with amplitude U _{1Ci is provided} .

The first term in braces is easily eliminated by the measuring filter 20 of the first phasing unit 16 ₁ , because its spectrum is much higher than the spectrum of the second term.

The second term in curly brackets (7) is harmonic oscillation (without manipulation) at the difference circular frequency ω _Фi = ω _Ci -ω _Pi , which coincides with the central frequency of the measuring filter of 20 phasing nodes 16 ₁ and 16 ₂ . Since this oscillation is directly proportional to the amplitude of the received signal V _1Ci and the transmission coefficient K ₁ (for U _Pi ≈const and K ₁ U _1Ci ≈const), then in the absence of interference at the inputs of the i-BCS under consideration, the amplitude of this oscillation will be the amplitude of this oscillation maximum and correspond to the maximum "weight" of the voltage of the received first sample of the i-th channel signal in the normalized oscillation at the output of the normalizing amplifier 18.

The output voltage of the measuring filter 20 of the first phasing unit 16 ₁ , taking into account the foregoing, can be represented as:

For a more accurate assessment in the phasing unit 16 ₁ (16 ₂ ) of the level or “weight” of the channel signal sample in the normalized mixture of signal and noise at the output of the normalizing amplifier 18, the passband of the measuring filter 20 of each phasing unit, on the one hand, should be extremely small, and on the other hand, it is necessary that this band provides the ability to "track" the signal level during its fading and changes in the carrier frequency of the channel signal during its reception. In the practical implementation of the communication system, this band can be selected on the order of (20-25) Hz.

Similarly to (8), one can imagine the output voltage of the measuring filter 20 of the second phasing unit 16 ₂ , which in this case also corresponds to the maximum “weight” of the voltage of the received second sample of the ith channel signal:

where K ₂ is the value of the transfer coefficient of the normalizing amplifier 18 of the second phasing unit 16 ₂ , at which normalization of the second sample of the input signal is provided.

The output of the second multiplier 21 of the first phasing node 16] will be:

Similarly, the output of the second multiplier 21 of the second phasing unit 16 _{2 is} recorded:

The first terms in braces (10) and (11) are eliminated during further filtering of the output product of the adder 22 by the filter of the resulting signal 23 and can be ignored. Therefore, the voltage of the first sample of the channel signal at the first input of the adder 22, which must be taken into account when summing, can be represented as:

Similarly, you can imagine the voltage of the second sample of the channel signal at the second input of the adder 22, which must be taken into account when summing:

In this case, the output voltage of the filter of the resulting i-th signal 23 is written in the form:

It follows from (14) that, in the absence of interference in the diversity branches at the phasing nodes 16 ₁ and 16 _{2, the} amplitudes of the corresponding samples of the ith channel signal are aligned by the normalizing amplifiers 18 to a certain value, the maximum range of which can reach 3 dB when the signal amplitude changes by input up to 100 dB. Moreover, within the considered range of the amplitude variation of the input oscillations, for example, limited to 40 dB, the amplitude of the output normalized oscillation U _CH can be considered constant:

Further, the voltage of the signals of the diversity branches is reduced to a single resulting frequency ω _Pi and phase ϕ _Pi , squared and fed to the corresponding inputs of the adder 22 for in-phase quadratic addition.

In view of (15), expression (14) can be represented as:

Using the normalizing amplifier 24, the resulting voltage U _{p CiФ} (t) is normalized by level, i.e. reduced to the form (6).

At the second stage of the analysis, we consider the operation of the i-th BCS under the above conditions, but taking into account the fact that at its first input there was additionally a spectrum-focused radio noise, for example, the so-called in-band sinusoidal noise (falling into the passband of the channel filter 17 BCS) of the form:

where U _1Пi and ϕ _1Пi are the amplitude and phase of the interference,

ω _1Pi is the carrier frequency of the interference falling into the passband of the channel filter 17 of the first phasing unit 16 _{1 of the} i-th BCS.

Thus, at the output of the channel filter 17 of the first phasing unit 16 ₁ , two oscillations will act - the first sample of the i-th channel signal U _1Ci (t) and interference U _1Пi (t), defined by expressions (3) and (17), respectively:

and at the output of the channel filter 17 of the second phasing unit 16 ₂ , a second sample of the i-th channel signal U _2Ci (t), defined by expression (4).

Such a situation may well occur in practice, for example, when the frequency diversity of the signals is used when loading the upper and lower side bands of the transmitter 6 with the same signal [5]. In this case, the narrow-target sighting radio noise falls into the passband of the channel filter of the i-th BCS, the inputs of which simultaneously receive samples of the group signal of the form (1) and (2) from the outputs of the RPU 10 ₁ and 10 ₂ , which separately receive the group signal in the upper and lower side strip accordingly.

In addition, a similar situation can arise under the influence of relatively broadband radio interference, individual spectral components of which fall into the passband of the channel filters of several BCS 15 ₁ ... 15 _N , affecting several channels simultaneously.

полагается настолько малой по сравнению со средней круговой частотой

, что результирующее колебание (18) можно считать узкополосным процессом [7].In expression (18), the detuning

relies so small compared to the average circular frequency

that the resulting oscillation (18) can be considered a narrow-band process [7].

Let M _i denote the initial channel signal-to-noise ratio:

then the envelope A (t) of the narrow-band process (18) taking into account (19) can be represented according to [7], p. 119, in the form:

In this case, the resulting oscillation (18) taking into account (20) can be represented in the form [7]:

where θ (t) is the function that describes the phase changes of the resulting oscillation with the central noise frequency ω _1Pi for U _1Пi > U _1Ci [7].

For U _1Ci > U _{1Pi, the} resulting oscillation (18) will have the form:

where R _i = U _1Pi / U _1Ci for R _i <1;

θ '(t) is a function that describes the phase changes of the resulting oscillation with the central signal frequency ω _Ci [7].

In order to simplify the analysis, we will further consider the operation of the compared communication systems in the most difficult communication conditions (19), i.e. when U _1Pi > U _1Ci , M _i <1.

The time constant of the AGC system of the normalizing amplifiers 18 should be much larger than the period of change of the envelope of the signal during angular manipulation, i.e. a signal element T _channel (including the beats of the resulting oscillation with a frequency Δω) and at the same time is less than the period of signal fading [6].

From (21) it follows that the average value of the amplitude of the resulting oscillation at the input of the normalizing amplifier 18, taking into account the maximum and minimum values of the variable cosΔωt, will be equal to:

The amplitude of the resulting oscillations at the output of the normalizing amplifier 18, taking into account (15) and (22), will be equal to:

Here K ₃ is the value of the transfer coefficient of the normalizing amplifier 18 of the first phasing unit 16 ₁ , at which its output level is normalized by the AGC system to almost the same level that was in the absence of impact interference, i.e. to a known value of U _CH (15).

From (23) we determine the value of K ₃ :

From (15) and (24) we can find how many times the value of the transfer coefficient K _{1 of the} normalizing amplifier 18 of the first phasing unit 16 ₁ corresponding to the reception of the first sample of the channel signal without interference will be greater than the value of the transfer coefficient K ₃ corresponding to the reception of the mixture of the same signal with sinusoidal interference, 1 / M _i times higher than the level of the first channel signal sample:

Using the principle of superposition, the output product of the first multiplier 19 of the first phasing unit 16 ₁ in this case will be:

Here, the first term characterizing the voltage of the first signal sample at the output of the first multiplier 19 of the first phasing branch 16 ₁ can be represented in the form of expression (7) with the replacement of K ₁ by K ₃ :

In this case, the output voltage of the measuring filter 20 of the first phasing unit 16 and corresponding to the "weight" of the received first sample of the i-th channel signal in a normalized mixture of signal voltage and noise, by analogy with (8) and taking into account (24), can be represented as:

The second term in (26), which characterizes the interference voltage at the output of the first multiplier 19 of the first phasing branch 16 _1, will have the form:

From consideration (29) it follows that in the first multiplier, the harmonic oscillation of the noise U _1Пi (t) is converted into two phase-shifted signals: at the upper carrier frequency ω _в = ω _Пi + ω _Pi (the first term in parentheses) and at the lower carrier frequency ω _{Pi n} = ω -ω _Pi (the second term in parentheses). The spectral components of the first phase-manipulated signal at the carrier frequency ω _are much higher than the center frequency ω _{Фi of the} measuring filter 20 and are easily eliminated by them, and individual spectral components of the second phase-manipulated signal at the carrier frequency ω _n can fall into the passband of the measuring filter 20.

However, taking into account that the passband of the measuring filter 20 is much smaller than the passband of the channel filter 17 corresponding to the spectrum of the received signal, the response of the narrow-band measuring filter 20 to the above individual spectral components of the phase-shifted signal can be ignored.

Thus, the output voltage of the measuring filter 20 when receiving a mixture of signal and sinusoidal interference with a certain degree of accuracy will correspond to the expression (28):

Using the principle of superposition, the output product of the second multiplier 21 of the first phasing unit 16 ₁ , taking into account (30), will have the form:

The first term in (31) determines the voltage of the first channel signal sample at the output of the second multiplier 21 of the first phasing branch 16 ₁ . By analogy with (10) and taking into account (24) and (28), this term can be written in the form:

Considering that the first term in (32) is eliminated during further filtering, we will take into account, as before, only the second term at the first input of adder 22, which characterizes the voltage of the first channel signal sample:

The second term in (31) determines the interference voltage at the first input of the adder 22. Taking into account (24) and (28), this term will have the form:

By analogy with (32) in (34), we will take into account only the second term at the first input of adder 22, which characterizes the interference voltage:

In the adder 22, the amplitudes of the signals from the outputs of the phasing nodes 16 ₁ , 16 ₂ and determined by expressions (33) and (13) taking into account (15) are added algebraically, the resulting signal amplitude at the output of the filter of the resulting signal 23 will be equal to:

Here, the first term determines the normalized amplitude of the second sample of the channel signal at the second input of the adder 22, provided that its amplitude U _2Ci at the input of the normalizing amplifier 18 of the second phasing node 16 ₂ exceeds the minimum amplitude U _{BX MIN of} the input signal range of the AGC of the normalizing amplifier 18, which is characterized by a coefficient of regulation (5).

From (35) the magnitude of the noise amplitude at the output of the filter of the resulting signal 23 will be equal to:

The resulting channel signal-to-noise ratio at the output of the filter of the resulting signal 23 or at the input of the i-th channel demodulator of the channel demodulator block 11 with optimal (weight) addition of spaced samples of the i-th channel signal taking into account (36) and (37) will be equal to:

To compare the noise immunity of the proposed communication system with the well-known communication system - the prototype [5] - we define the resulting channel signal-to-noise ratio with linear addition and with the similar (discussed above) action of two samples of the channel signal and in-band sinusoidal noise at the corresponding two inputs of the device providing linear addition of vibrations.

As noted above, with linear addition, strict requirements are put forward to ensure the equality of gain in diversity branches, i.e. radio receivers of the diversity branches must be strictly identical in gain [6]. Assume that this condition is met.

Similarly to the above we consider operation of the communication prototype system within one of the time intervals Δt stationarity reception baseband signal, much longer than the duration of a binary element T _kan signal within which the amplitude of each sample i-th channel signal at the corresponding input node of the linear addition can be considered constant .

Let, as before, the samples of the group signal U _1C (t) and U _2C (t) of the form (1) and (2), as well as the in-band sinusoidal interference U _1Pi (t) of the form (13) from the outputs of the linear paths corresponding to the RPU the receiving complex of the known communication system of the prototype, arrive at N channel blocks (KB), each of which is designed for dual reception and consists of two identical active filters [5].

To simplify the analysis, we will accept the following model of a device for linear addition of diversity signals instead of real devices for linear addition of a prototype communication system:

- in each i-th (i = 1, 2, ..., N) channel block of the prototype communication system, two samples of the i-th channel signal are filtered by the traditional method (using bandpass filters without switching to zero frequency and decomposing the signal into quadrature components);

- in the phase difference calculation unit (BVRF), in-phase linear addition of two samples of each i-th channel signal is performed, followed by demodulation of the i-th channel signal in a known manner (the same as in the proposed communication system).

With this simplification, the noise immunity of the prototype communication system with the adopted linear addition model cannot be worse than the noise immunity of the known communication system that uses the less preferred post-detector addition of diversity signals [6].

The voltage of the signal samples and the interference of the diversity branches allocated by the filters of the ith channel block can be represented as:

- along the first branch -

on the second branch -

The amplitude of the resulting signal after in-phase linear addition in the BVRF will be equal to (with the transfer coefficient of the summing device equal to 1):

From (19) the voltage of the additive in-band sinusoidal noise at the output of the summing device is equal to:

From (41) and (42) we find the resulting channel signal-to-noise ratio at the output of the ith linear addition device or at the input of the ith channel demodulator:

Let be

where L _i is a number indicating how many times the magnitude of the amplitude U _{2Ci is} greater (less) than the amplitude U _1Ci .

Then (43) takes the form:

Thus, expressions (38) and (45) make it possible to evaluate the resulting channel signal-to-noise ratios at the output of any ith addition device (optimal and linear) of spaced samples of the ith channel signal (or at the input of the ith channel demodulator) of each of the compared communication systems, depending on the initial channel signal-to-noise ratio at the input (output) of the linear path of one of the RPU of the communication system in the presence of a signal without interference at the input (output) of the linear path of another RPU.

However, when comparing the values of H _Pionm and H _Rilin, it is necessary to know the critical value of the signal-to-noise ratio (in-band sinusoidal) N _cr , below which the demodulator malfunctions, as explained above. Since we previously agreed that the demodulators of the compared communication systems are identical, the value of N _cr for the compared communication systems should be the same.

The numerical value of N _cr for the demodulators that are used as part of the RPU is determined by the noise immunity parameter of the RPU telegraph channels under the influence of in-band sinusoidal interference. The requirement for noise immunity is presented to any modern industrially produced RPU, for example, to any RPU given in [8]. For modern RPUs, the critical signal-to-noise ratio (in-band sinusoidal) at the input of the RPU (in dB) is not more than 4.5 dB when receiving CT signals and not more than 8 dB when receiving signals of OFT at different operating speeds.

In fact, this is a requirement for the noise immunity of the demodulator itself as part of the RPM, since the total signal voltage and interference when it passes through the linear path (from the antenna input of the RPU to the input of the demodulator in the RPU) is not subject to significant nonlinear distortions [6, 8].

We will take the ratio N _cr equal to 8 dB (2.51 times) as the noise immunity parameter of each channel demodulator of the compared communication systems, and calculate the resulting channel signal-to-noise voltage ratios (H _Pionm and H _Pilin ) at the input of each of the channel demodulators of both systems according to formulas (38) and (45) for various initial channel signal-to-noise voltage ratios M _i at the input (output) of the linear path of one of the two RPFs of each of the systems in the passband of the i-th channel signal into which it passes a sinusoidal interference noise, and for different voltage ratios of two samples of the i-th channel signal L _i at the outputs (inputs) of the corresponding RPU.

The calculated data are given in the table.

From the analysis of the table it follows that the proposed high-speed decameter radio communication system significantly exceeds the well-known radio communication system - a prototype for the noise immunity of receiving any i-th channel signal with the simultaneous action at the input (output) of the linear path of one of two RPUs 10 ₁ or 10 ₂ , in-band sinusoidal interference within the frequency band occupied by the spectrum of the corresponding sample of this signal.

The above calculations show that the known communication system provides the reception of any i-th channel signal without errors only in one case - when the voltage of one sample of the channel signal free of interference is exceeded, the voltage of another sample of this signal affected by the interference is not less than 2 times , with M _i = 0.9 and L _i = 2.

Under other communication conditions (for different values of M _i and L _i ), the resulting channel relationship H _Pilin <H _cr = 8 dB (2.51 times), i.e. the operation of each of the channel demodulators is blocked by the in-band sinusoidal interference and the reception of a group signal is not possible.

However, under the same communication conditions, the proposed communication system provides the resulting channel relationship H _Pionm > H _cr , which allows you to receive any of the channel signals without errors, and accordingly to receive the group signal as a whole also without errors.

In conclusion, based on the above comparative analysis, we can note the following advantages of the proposed high-speed decameter radio communication system in relation to the known prototype communication system:

1. When the input voltage of one of the RPU 10 ₁ or 10 ₂ voltage concentrated on the spectrum (in-band) additive interference falling into the frequency band occupied by the spectrum of any i-th channel signal (as part of the N-channel group signal), wherein the value of i-th channel stress ratio signal / noise ratio M _i <1, is achieved a considerable gain of noise immunity receiving i-th channel signal by achieving a higher value of the resulting channel voltage ratio signal / noise ratio H _Pionm> 1-i-r output BCS 15 _i, and thus the inlet of the corresponding i-th channel demodulator unit channel demodulators 11, substantially greater than the critical value of H _Pionm = H _cr at which the failure of the demodulator.

2. In view of the mutual independence of the reception of each i-th channel signal, a higher noise immunity of the reception of the group signal as a whole is achieved when one of the in-band interference and G in-band interference (G≤N) act on the input of one of two RPUs 10 ₁ or 10 ₂ , each of which falls into the frequency band occupied by the corresponding one of the N channel signals in the group signal.

3. A gain is also achieved in the noise immunity of the reception and under the influence of fluctuation interference acting within the frequency bands occupied by channel signals due to the implementation in each BCS 15 ₁ ... 15 _{N of the} optimal coherent addition of samples of channel signals performed prior to the operation of detecting these signals [ 2, 6].

4. Expanding the functionality for transmitting and receiving a wider class of signals with angular manipulation by providing each BCS 15 ₁ ... 15 _{Ν the} optimal coherent addition of images of the corresponding channel signal with phase difference and frequency manipulation at any multiplicity to signal compression, and also demodulating these signals with a block of N channel demodulators 11 by known methods [2, 6].

Information sources

1. Klovsky D.D. Theory of signal transmission. Textbook for high schools. - M .: Communication. 1973. 376 p.

2. Fink L.M. Theory of discrete message transmission. - M .: Soviet radio. 1970.728 s.

3. Klovsky D. D., Nikolaev B. I. Engineering implementation of radio circuits (in discrete message transmission systems under conditions of intersymbol interference). - M .: Communication. 1975.200 p.

4. Kiselev A.M., Mahotin V.V., Ryzhov N.Yu., Shatalova G.V. A method for implementing a high-speed parallel modem // Radio engineering. 2006. Issue. 11. S. 5-15.

5. Ginsburg VV, Girshov B.C., Zayezny A.M., Kagan BD, Kustov OV, Okunev Yu.B. and other equipment for the transmission of discrete information MS-5 / Ed. Arrival A.M. and Okuneva Yu.B. - M .: Communication. 1970.

6. N.A. Sartasov, V.M. Edvabny, V.V. Mushroom. Short-wave trunk radio receivers. - M .: Communication. 1971.288 s.

7. I.S. Honorovsky. Radio circuits and signals. Textbook for high schools. Ed. 3rd, rev. and add. - M: “Owls. radio". 1977. 608 p.

8. Berezovsky V.A., Dulkeit I.V., Savitsky O.K. Modern decameter radio communication: equipment, systems and complexes. Ed. V.A. Berezovsky. - M: Radio engineering, 2011 .-- 444 p.

Claims (1)

A high-speed decameter radio communication system comprising a transmitting complex comprising a serially connected message source, an encoder and a serial-parallel converter, the outputs of which are connected to the corresponding inputs of a block N of channel manipulators, the output of which is connected to a serially connected radio transmitting device and a transmitting antenna, as well as a receiving complex, comprising two receiving antennas, the output of each of which is connected to the input of the corresponding radio receiving device, as well as a block of N channel demodulators, the outputs of which are connected to the corresponding inputs of the parallel-serial converter, the output of which is connected to the decoder and the receiver of messages in series, characterized in that N blocks of coherent signal addition (BCS) are introduced into the receiver complex, one input of each of which is combined with the output of one radio receiver, and the other input of each BCS is combined with the output of another radio receiver, and the output of each BCS is connected to by the corresponding input of the block of N channel demodulators, each BCS contains two phasing units, each of which contains a channel filter connected in series, the input of which is a corresponding input of the BCS, a normalizing amplifier, a first multiplier, a measuring filter and a second multiplier, the other input of which is connected to the input of the first multiplier , the output of the second multiplier of each phasing unit is connected to the corresponding input of the adder, the output of which is connected to the input of the filter of the resulting signal, in course of which being output BCS is connected via a normalizing amplifier resultant signal to another input of the first multiplier of each node phasing.

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