CN112803968B - Airborne measurement and control method for unmanned aerial vehicle - Google Patents
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Abstract
The invention discloses an airborne measurement and control method of an unmanned aerial vehicle, which comprises a carrier synchronization step, a pseudo code synchronization step and a multi-set receiving step, wherein a received uplink signal is a direct sequence spread spectrum signal, and a local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal; carrying out spread spectrum code capture on the baseband spread spectrum signal, and outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful; and outputting a plurality of paths of local spread spectrum codes with different phases to perform correlation operation with the baseband spread spectrum signals respectively, and then performing selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand. The method has strong anti-interference, anti-multipath and anti-flash capabilities.
Description
Technical Field
The invention relates to the field of airborne measurement and control communication, in particular to an airborne measurement and control method of an unmanned aerial vehicle.
Background
The airborne measurement and control of the unmanned aerial vehicle mainly complete the remote control of the flight state of the unmanned aerial vehicle, the remote measurement of flight parameters and the tracking of the unmanned aerial vehicle. Along with the flying speed of the unmanned aerial vehicle is faster and faster, the flying distance is further and further, the wireless mobile communication between the airborne measurement and control of the unmanned aerial vehicle and the ground measurement and control works in the environment with long distance, low elevation angle and high dynamic state most of the time, therefore, in the flying process of the unmanned aerial vehicle, electromagnetic waves transmitted between the unmanned aerial vehicle and a ground station can be reflected, refracted, diffracted and even shielded by objects such as trees, buildings and the like, so that serious multipath effect and flash phenomenon are caused, meanwhile, the interference caused by various communication systems on the earth surface can not be ignored, and therefore, the airborne measurement and control method of the unmanned aerial vehicle must have certain anti-interference, anti-multipath and anti-flash capabilities.
Disclosure of Invention
The invention mainly solves the technical problem of providing an airborne measurement and control method of an unmanned aerial vehicle, and solves the problem that the airborne measurement and control method of the unmanned aerial vehicle in the prior art has insufficient capacity in the aspects of interference resistance, multipath resistance and flash resistance in the process of receiving uplink measurement and control signals.
In order to solve the technical problems, the invention adopts a technical scheme that an airborne measurement and control method of an unmanned aerial vehicle is provided, and the airborne measurement and control method comprises the following steps: carrier synchronization: the received uplink signal is a direct sequence spread spectrum signal, and the local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal; pseudo code synchronization: carrying out spread spectrum code capture on the baseband spread spectrum signal, and outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful; receiving a plurality of sets: and outputting a plurality of paths of local spread spectrum codes with different phases to perform correlation operation with the baseband spread spectrum signals respectively, and then performing selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand.
Preferably, in the multi-set receiving step, a demodulation tracking carrier frequency offset value is further output, and is also fed back and input to the local carrier for frequency offset correction.
Preferably, the pseudo code synchronization step includes a first FFT transformation process, a second FFT transformation process, a conjugate multiplication process, an IFFT transformation process, and a capture judgment process; the first FFT conversion processing comprises receiving the baseband spread spectrum signal and carrying out FFT conversion on the baseband spread spectrum signal, and the second FFT conversion processing comprises receiving a local pseudo code sequence output by a local pseudo code generator and carrying out FFT conversion on the local pseudo code sequence; the output results of the two FFT conversion processes further complete complex conjugation and complex multiplication operations in the conjugate multiplication process, then are input into the IFFT conversion process for IFFT operation, the capture judgment process carries out capture judgment and identification on the IFFT operation results to obtain a pseudo code capture carrier frequency offset value and a pseudo code capture code phase value, the pseudo code capture carrier frequency offset value is output to carrier synchronization for carrier frequency offset correction, and the pseudo code capture code phase value is output to the local pseudo code generator for pseudo code phase correction.
Preferably, in the pseudo code synchronization step, a data framing process is further included before the first FFT process, and a pseudo code zero padding framing process is further included before the second FFT process.
Preferably, in the pseudo code synchronization step, the conjugate multiplication processing performs complex conjugate calculation on the result output by the second FFT processing, and then performs complex multiplication calculation on the result output by the first FFT processing.
Preferably, in the multi-set receiving step, a plurality of correlation operation channel processes are included, and correlation operations are performed respectively with the local spread spectrum codes of different phases output in the pseudo code synchronizing step, and then the operation results are subjected to selective combining processing, and the results of the selective combining processing are output to pseudo code tracking processing and frequency-locked loop differential demodulation processing respectively.
Preferably, in the multi-set receiving step, in each correlation operation channel processing, the delay conjugate processing and the conjugate multiplication processing are included, and the output expression after the correlation operation processing is performed on the baseband spread spectrum signal may be represented as:
wherein A isnRepresenting the amplitude of a symbol in a received signal, D (N) representing a spreading code, N representing the serial number of the spreading code, i representing the serial number of a correlation operation channel, comprising N correlation operation channels, i is more than or equal to 1 and less than or equal to N,
representing the carrier phase difference brought by the multipath time delay of each correlation operation channel;
in each correlation operation channel processing, after delaying one code phase by delay processing, the following steps are carried out:
wherein, T represents a symbol period delayed by one spreading code, and after conjugate multiplication, the output result is:
ri′＝ri(n)*ri *(n+1)＝An 2*D(n)*D(n+1)e-j2πfT。
preferably, in the multi-set receiving step, the results respectively output by the N delay conjugation processes in the N correlation operation channel processes are compared and selected in the selection combining process, and a signal with a large signal-to-noise ratio is selected from the results and combined.
Preferably, in the multi-set receiving step, the pseudo code tracking processing includes phase demodulation processing and code loop filtering processing, and the frequency-locked loop differential demodulation processing includes frequency-locked loop processing and differential demodulation processing.
The invention has the beneficial effects that: the invention discloses an airborne measurement and control method of an unmanned aerial vehicle, which comprises a carrier synchronization step, a pseudo code synchronization step and a multi-set receiving step, wherein a received uplink signal is a direct sequence spread spectrum signal, and a local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal; carrying out spread spectrum code capture on the baseband spread spectrum signal, and outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful; and outputting a plurality of paths of local spread spectrum codes with different phases to perform correlation operation with the baseband spread spectrum signals respectively, and then performing selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand. The method has strong anti-interference, anti-multipath and anti-flash capabilities.
Drawings
FIG. 1 is a block diagram of components according to one embodiment of an airborne test and control terminal for an unmanned aerial vehicle;
FIG. 2 is a detailed block diagram of another embodiment of an airborne test and control terminal of an unmanned aerial vehicle;
FIG. 3 is a block diagram of pseudo code synchronization modules according to another embodiment of an airborne test and control terminal of an unmanned aerial vehicle;
fig. 4 and 5 are frequency spectrums generated by a pseudo code synchronization module according to another embodiment of an airborne measurement and control terminal of an unmanned aerial vehicle;
FIG. 6 is a block diagram of a multi-set receiving module according to another embodiment of an airborne test and control terminal of an unmanned aerial vehicle;
FIG. 7 is a block diagram of a selective combiner in a multi-set receive module according to another embodiment of an airborne test and control terminal for an unmanned aerial vehicle;
FIG. 8 is a block diagram of a pseudo code tracker in a multi-set receiving module according to another embodiment of an airborne test and control terminal of an unmanned aerial vehicle;
fig. 9 is a block diagram of a frequency-locked loop differential demodulator in a multi-set receiving module according to another embodiment of an airborne measurement and control terminal of an unmanned aerial vehicle;
fig. 10 is a flowchart of an embodiment of an airborne measurement and control method for an unmanned aerial vehicle according to the present invention.
Detailed Description
In order to facilitate an understanding of the invention, the invention is described in more detail below with reference to the accompanying drawings and specific examples. Preferred embodiments of the present invention are shown in the drawings. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete.
It is to be noted that, unless defined otherwise, all technical and scientific terms used herein have the same meaning as commonly understood by one of ordinary skill in the art to which this invention belongs. The terminology used in the description of the invention herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the invention. As used herein, the term "and/or" includes any and all combinations of one or more of the associated listed items.
Fig. 1 shows a block diagram of an embodiment of an airborne measurement and control terminal of an unmanned aerial vehicle. In fig. 1, the airborne measurement and control terminal of the unmanned aerial vehicle includes an uplink signal demodulation unit, the uplink signal is a direct sequence spread spectrum signal, the uplink signal demodulation unit further includes a carrier synchronization module 1, a pseudo code synchronization module 2 and a multi-set receiving module 3, the carrier synchronization module 1 receives a pseudo code output from the pseudo code synchronization module 2 to capture a carrier frequency offset value, so as to perform frequency offset correction on a local carrier generated by the carrier synchronization module, and perform down-conversion on a carrier of the uplink signal by using the local carrier after the frequency offset correction to obtain a baseband spread spectrum signal; the baseband spread spectrum signal is input to the pseudo code synchronization module 2 for spread spectrum code capture, after the spread spectrum code capture is successful, the pseudo code synchronization module 2 outputs a plurality of local spread spectrum codes with different phases to the multi-set receiving module 3, and performs correlation operation with the baseband spread spectrum signal input to the multi-set receiving module respectively, and then performs selection and combination, and the output result is used for pseudo code tracking on one hand, and on the other hand, information demodulation outputs remote control information, and demodulation also outputs the demodulation tracking carrier frequency offset value to the carrier synchronization module 1.
Preferably, the multiple-set receiving module 3 outputs a demodulation tracking carrier frequency offset value, and also feeds back the demodulation tracking carrier frequency offset value to the carrier synchronization module for local carrier frequency offset correction.
Further, on the basis of the embodiment shown in fig. 1, the internal components of each component block are further shown in fig. 2, wherein the pseudo code synchronization block 2 comprises a first FFT transformer 21, a second FFT transformer 22, a local pseudo code generator 23, a conjugate multiplier 24, an IFFT transformer 25 and an acquisition decider 26. The first FFT converter 21 receives the baseband spread spectrum signal from the down converter 11, and performs FFT conversion on the baseband spread spectrum signal, and the second FFT converter 22 receives the local pseudo code sequence output from the local pseudo code generator 23, and performs FFT conversion on the local pseudo code sequence. The output results of the two FFT converters complete complex conjugate and complex multiplication operations in the conjugate multiplier 24, specifically, the complex conjugate calculation is performed on the result output by the second FFT converter 22, then the complex multiplication calculation is performed on the result output by the first FFT converter 21, the multiplied result is input into the IFFT converter 25 for IFFT operation, the capturing decision unit 26 performs capturing decision and identification on the IFFT operation result, so as to obtain a pseudo code capturing carrier frequency offset value and a pseudo code capturing code phase value, the pseudo code capturing carrier frequency offset value is output to the down converter 11 for correcting the local carrier, and the pseudo code capturing code phase value is output to the local pseudo code generator 23 for adjusting the code phase of the local pseudo code sequence output by the local pseudo code generator 23, thereby completing the pseudo code synchronization of the baseband spread spectrum signal, i.e., achieving the pseudo code capturing.
Preferably, fig. 3 further shows the internal components of the pseudo code synchronization module, wherein a data framer 211 is further included before the first FFT transformer 21, and a pseudo code zero padding framer 221 is further included before the second FFT transformer 22. It can be seen that since the number of data bits processed by the first FFT converter 21 and the second FFT converter 22 is the same, and the subsequent complex multiplication operation is also facilitated, the number of data bits input to the first FFT converter 21 is required to be the same as the number of data bits of the pseudo code sequence input to the second FFT converter 22. Preferably, the number of data bits FFT-ed by the second FFT converter 22 is close to 2 times the number of cycle bits of the pseudo code sequence, for example, the number of data bits FFT-ed by the second FFT converter 22 is 2048 bits, the number of data bits is usually an integer power of 2, for example 2048 is 11 times 2, and the number of cycle bits of the local pseudo code sequence is 1023, obviously 2048 is close to 2 times 1023. Thus, for the local pseudo-code sequence, the remaining number of zeros, for example, 1025 zeros, need to be complemented at the end of the local pseudo-code sequence by the pseudo-code zero-padding framer 221. Further, the bit number of the data of the baseband spread spectrum signal included in the data framer 211 before the first FFT converter 21 corresponds to an integer multiple of the bit number of the local pseudo code sequence period, and insufficient data bits are also processed by zero padding, for example, if the bit number of the local pseudo code sequence period is 1023, the bit number of the data of the baseband spread spectrum signal included in the data framer 211 is two times 1023, which is just 2046, but is not enough 2048 bits, and insufficient 2 bits are obtained by zero padding.
Further, based on the embodiment shown in fig. 3, the data framing of the baseband spread spectrum signal and the optimization of the code-taking and zero-filling framing part of the local pseudo code sequence are as follows: the method comprises the steps of collecting data of two code segments for a baseband spread spectrum signal, wherein the length of each code segment is 1023, the storage depth is 2046, 2048-point FFT calculation is completed after 2 zeros are complemented, 2048-point FFT calculation is completed after 1025 zeros are complemented by a pseudo code of 1023 periods locally, then complex conjugation is carried out on an FFT result of the pseudo code, the complex multiplication result is multiplied with the FFT result of the data of the baseband spread spectrum signal, the complex multiplication result is used as the input of IFFT calculation, the first 1023 values of the 2048 data output by the IFFT calculation represent an autocorrelation function of a pseudo code sequence, and the last 1025 values need to be discarded completely due to the fact that the periodic characteristic of the pseudo code sequence is damaged.
Although the calculation length of the FFT is doubled in the embodiment shown in fig. 3, since only one pseudo code sequence period 1023 is reserved when the local pseudo code is FFT calculated, all the rest is zero-filled, that is, all the rest 1025 points have zero influence on the autocorrelation value, and meanwhile, two pseudo code sequence periods are buffered in the data of the received baseband spread spectrum signal, the period characteristic of the pseudo code sequence is not destroyed by the cyclic convolution period of the FFT calculation, and the autocorrelation function is not affected thereby. The processing method realizes FFT-IFFT parallel pseudo code correlation operation, can quickly capture the pseudo code phase without influencing the capture correlation peak value and reducing the capture performance by only adding a small amount of hardware resources, and obtains the frequency offset value.
The corresponding calculation result of the pseudo code capturing scheme shown in fig. 3 is shown in fig. 4, the result of the capturing by the correlation operation using the conventional multi-path correlator is shown in fig. 5, and the comparison result shows that the capturing results of the two diagrams are basically consistent, and a positive maximum value can be seen near the abscissa of 300, because the latter half of the data in the calculation result of the invention is useless and discarded, only the front 1023-bit correlation value is needed, the 2048-bit calculation result is shown in fig. 4, only the 1023-bit result is shown in fig. 5, and only the 1023-bit result before the comparative analysis is needed.
Further, as shown in fig. 2, the multi-set receiving module 3 includes a plurality of correlation operation channels, and the correlation operation channels perform correlation operations with the plurality of local spreading codes with different phases output by the pseudo code synchronization module 2 respectively. It can be seen that the first correlation operation channel includes a first correlator 311 and a first delay conjugator 312, the second correlation operation channel includes a second correlator 321 and a second delay conjugator 322, and so on, so that the nth correlation operation channel includes an nth correlator 331 and an nth delay conjugator 332, where N is greater than or equal to 2. The calculation results of these multiple correlation operation channels are input to the selection combiner 34 for result selection and combination, the selection combiner 34 outputs the results to the pseudo code tracker 35 and the frequency-locked loop differential demodulator 36 respectively, the pseudo code tracker 35 is used for pseudo code synchronous tracking, outputs a correction signal to the local pseudo code generator 23 for phase correction of the output pseudo code sequence, the frequency-locked loop differential demodulator 36 is used for information demodulation and output of remote control information, and simultaneously, also uses the frequency difference information of demodulation and tracking as a demodulation and tracking carrier frequency offset value to the down converter 11 for regulating and controlling the frequency of the local carrier.
Further, as shown in fig. 6, a composition of the multi-set receiving module 3 is shown, wherein for each correlation operation channel, the delay conjugator includes a delay device and a conjugate multiplier, and the output expression of the baseband spread spectrum signal output by the down converter after passing through the correlator can be expressed as:
wherein A isnRepresenting the amplitude of a symbol in a received signal, D (N) representing a spreading code, i representing the serial number of the correlation operation channels, N correlation operation channels in total, i is more than or equal to 1 and less than or equal to N,
representing the carrier phase difference brought by the multipath time delay of each correlation operation channel;
in each channel, after being delayed by one code phase by a delayer, the following steps are carried out:
where T represents a delay time duration, which is usually a symbol period of a spreading code, and after passing through the conjugate multiplier, the expression of the output result is:
ri′＝ri(n)*ri *(n+1)＝An 2*D(n)*D(n+1)e-j2πfT
it can be seen that the carrier phase difference of the multipath component caused by the time delay of the multipath component is cancelled after conjugate multiplication of the two symbols before and after the carrier phase difference of the multipath component, so that diversity combining can be realized without performing parameter estimation of the multipath component.
Further, fig. 7 shows the internal components of the selection combiner in fig. 2 and fig. 6, and the results respectively output from the N delay conjugators in the N correlation operation channels are selected by the energy comparator in the selection combiner, from which the signal with a large signal-to-noise ratio is selected for combining. Specifically, according to the principle that the larger the signal-to-noise ratio of the signal output by the delay conjugator is, the larger the weight is, and the smaller the signal-to-noise ratio is, the smaller the weight is, the corresponding weighting compensation is made, so that the signal-to-noise ratio of the combined output is the largest, and the weighting coefficient can be expressed as:
wherein alpha islFor the signal amplitude of the l branch in the selected M branches, R is the sum of the signal energies of the selected M branches. It can be seen that the amplitude a of the signal component is good when the channel conditions are goodlBecomes large, the weighting coefficient wlThe larger the increase, the greater the total signal contribution to the combination; similarly, when the channel condition is poor, the amplitude α of the signal componentlBecomes smaller, the weighting coefficient wlThe smaller the total signal contribution to the combining. Thus, the weighting factor wlCan suppress noise and enhance signal amplitude to make signal-to-noise ratio gamma of combined outputMRCMaximum, i.e.:
preferably, as shown in fig. 2, one path of the result output by the combiner is output to the pseudo code tracker, and the other path of the result is output to the frequency-locked loop differential demodulator. As for the pseudo code tracker, as shown in fig. 8, the pseudo code tracker includes a phase detector and a code loop filter, when the pseudo code phase of the spread spectrum signal is successfully captured, due to the existence of carrier frequency offset, the locally generated spread spectrum code and the pseudo code of the received spread spectrum signal are not completely consistent in phase, and it is also necessary to accurately track the change of the code phase of the input spread spectrum signal and calibrate the code phase error within an allowable range, which requires the pseudo code tracker. One input end of the phase detector receives signals output by the selective combiner, the signals comprise a signal set subjected to selective combining, and noise is mixed in the signals, the other input end of the phase detector is from a code loop filter, the code loop filter performs loop filtering on the results of the phase detector and outputs a pseudo code sequence subjected to phase smoothing, and phase comparison and correction of the input signals are kept. Meanwhile, the phase discrimination output result is also used for regulating and controlling the phase of the local pseudo code, and the tracking synchronization of the local pseudo code and the pseudo code in the received baseband spread spectrum signal is realized.
Preferably, as shown in fig. 9, the frequency-locked loop differential demodulator includes a feedback-type frequency-locked loop and a differential demodulator. The frequency-locking loop can complete frequency tracking and locking of a local carrier and a received signal, and the differential demodulator outputs differential demodulation data from the carrier-synchronous frequency-locking loop to obtain uplink remote control information. The frequency-locking loop also feeds back a demodulation tracking carrier frequency offset value obtained in real time by tracking and locking to the down converter for frequency offset error correction of the front end.
The frequency-locked loop differential demodulator has strong adaptability and high signal tracking sensitivity, can adapt to Doppler effect caused by speed, acceleration and jerk of the unmanned aerial vehicle relative to a ground station in the high-speed flight process by using a stable low-order loop filter, replaces a channel estimation module on each branch of a selective combiner with a delay conjugate multiplication module by using a simple structure of the differential demodulator, and provides technical support for the miniaturization design of the unmanned aerial vehicle. Moreover, aiming at a plurality of branches of the selective combiner, even if the direct component or the multipath components on any number of branches are shielded by an object or are strongly interfered by the outside, the phenomenon of flash occurs, the frequency-locked loop can be ensured to always work in a stable state, and the signal-to-noise ratio of the input end of the frequency-locked loop is improved.
Furthermore, the frequency-locked loop is used instead of the phase-locked loop, because the phase-locked loop needs to adopt a high-order loop structure to track the large dynamic Doppler change, the loop parameter setting is complex, and the loop is easy to be unstable due to improper setting. And the adaptability of the frequency locking ring is strengthened, the signal tracking sensitivity is higher, and the frequency locking ring can adapt to the Doppler effect caused by the speed, the acceleration and the acceleration rate of the unmanned aerial vehicle relative to the ground equipment in the high-speed flight process.
Based on the same concept, the invention discloses an airborne measurement and control method of an unmanned aerial vehicle, which comprises the following steps as shown in figure 10:
carrier synchronization S1: the received uplink signal is a direct sequence spread spectrum signal, and the local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal;
pseudo code synchronization S2: carrying out spread spectrum code capture on the baseband spread spectrum signal, and outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful;
multi-set reception S3: and outputting a plurality of paths of local spread spectrum codes with different phases to perform correlation operation with the baseband spread spectrum signals respectively, and then performing selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand.
Preferably, in the step of S3, the demodulation tracking carrier frequency offset value is also output, and is also fed back to the local carrier for frequency offset correction.
The above three steps correspond to the carrier synchronization module 1, the pseudo code synchronization module 2, and the multiple-set receiving module 3 in fig. 1, and specific reference may be made to the description of the embodiment in fig. 1, which is not described herein again.
Preferably, in the pseudo code synchronization S2 step, a first FFT transformation process, a second FFT transformation process, a conjugate multiplication process, an IFFT transformation process, and a capture judgment process are included; the first FFT conversion processing comprises receiving the baseband spread spectrum signal and carrying out FFT conversion on the baseband spread spectrum signal, and the second FFT conversion processing comprises receiving a local pseudo code sequence output by a local pseudo code generator and carrying out FFT conversion on the local pseudo code sequence; the output results of the two FFT conversion processes further complete complex conjugation and complex multiplication operations in the conjugate multiplication process, then are input into the IFFT conversion process for IFFT operation, the capture judgment process carries out capture judgment and identification on the IFFT operation results to obtain a pseudo code capture carrier frequency offset value and a pseudo code capture code phase value, the pseudo code capture carrier frequency offset value is output to carrier synchronization for carrier frequency offset correction, and the pseudo code capture code phase value is output to the local pseudo code generator for pseudo code phase correction. For related matters, reference may be made to the foregoing description of the first FFT transformer, the second FFT transformer, the local pseudo code generator, the conjugate multiplier, the IFFT transformer, and the capture decider in the embodiment shown in fig. 2, which is not repeated herein.
Preferably, in the pseudo code synchronization S2, a data framing process is further included before the first FFT process, and a pseudo code zero padding framing process is further included before the second FFT process. For related content, reference may be made to the foregoing description of the data framer before the first FFT transformer and the pseudo code zero padding framer before the second FFT transformer in the embodiment shown in fig. 3, which is not described herein again.
Preferably, in the pseudo code synchronization S2, the conjugate multiplication process performs complex conjugate calculation on the result output by the second FFT process, and then performs complex multiplication on the result output by the first FFT process. For related matters, reference may be made to the description of the conjugate multiplier in the foregoing embodiment shown in fig. 3, which is not described herein again.
Preferably, in the step of S3, the method includes a plurality of correlation operation channel processes, and performs correlation operations with the local spreading codes with different phases output by the pseudo code synchronization step, and then performs selective combining processing on the operation results, and the results of the selective combining processing are output to pseudo code tracking processing and frequency-locked loop differential demodulation processing, respectively. For related content, reference may be made to the foregoing description of the multiple correlation operation channels, the correlator, the selective combiner, the pseudo code tracker, and the frequency-locked loop differential demodulator in the embodiments shown in fig. 2 and fig. 6, which is not described herein again.
Preferably, in the step of multi-set receiving S3, in each correlation operation channel, the delay conjugate processing includes delay processing and conjugate multiplication processing, and the output expression after the correlation operation processing is performed on the baseband spread spectrum signal may be represented as:
wherein A isnRepresenting the amplitude of a symbol in a received signal, D (N) representing a spreading code, N representing the serial number of the spreading code, i representing the serial number of a correlation operation channel, comprising N correlation operation channels, i is more than or equal to 1 and less than or equal to N,
representing the carrier phase difference brought by the multipath time delay of each correlation operation channel;
in each correlation operation channel processing, after delaying one code phase by delay processing, the following steps are carried out:
wherein, T represents a symbol period delayed by one spreading code, and after conjugate multiplication, the output result is:
ri′＝ri(n)*ri *(n+1)＝An 2*D(n)*D(n+1)e-j2πfT。
for related matters, reference may be made to the foregoing description of the delayer and the conjugate multiplier in the embodiment shown in fig. 6, which is not described herein again.
Preferably, in the step of S3, the results output by the N delay conjugation processes in the N correlation operation channel processes are respectively subjected to energy comparison and selection in a selection combining process, and a signal with a large signal-to-noise ratio is selected from the results and combined. For related matters, reference may be made to the foregoing description of the selection combiner, which is not described in detail herein.
Preferably, in the step of multi-set receiving S3, the pseudo code tracking processing includes phase demodulation processing and code loop filtering processing, and the frequency-locked loop differential demodulation processing includes frequency-locked loop processing and differential demodulation processing. For the related content, reference may be made to the foregoing descriptions of the pseudo code tracker, the phase detector, and the code loop filter, and the frequency-locked loop differential demodulator, the frequency-locked loop, and the differential demodulator, which are not described herein again.
Therefore, the invention discloses an airborne measurement and control method of an unmanned aerial vehicle, which comprises a carrier synchronization step, a pseudo code synchronization step and a multi-set receiving step, wherein a received uplink signal is a direct sequence spread spectrum signal, and a local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal; carrying out spread spectrum code capture on the baseband spread spectrum signal, and outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful; and outputting a plurality of paths of local spread spectrum codes with different phases to perform correlation operation with the baseband spread spectrum signals respectively, and then performing selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand. The method has strong anti-interference, anti-multipath and anti-flash capabilities.
The above description is only an embodiment of the present invention, and not intended to limit the scope of the present invention, and all equivalent structural changes made by using the contents of the present specification and the drawings, or applied directly or indirectly to other related technical fields, are included in the scope of the present invention.
Claims (6)
1. An airborne measurement and control method of an unmanned aerial vehicle is characterized by comprising the following steps:
carrier synchronization: the received uplink signal is a direct sequence spread spectrum signal, and the local carrier is utilized to carry out down-conversion on the uplink signal to obtain a baseband spread spectrum signal;
pseudo code synchronization: carrying out spread spectrum code capture on the baseband spread spectrum signal, outputting a pseudo code to capture a carrier frequency offset value to carry out frequency offset correction on a local carrier after the spread spectrum code capture is successful, and outputting a plurality of paths of local spread spectrum codes with different phases;
receiving a plurality of sets: carrying out correlation operation on a plurality of paths of local spread spectrum codes with different phases and baseband spread spectrum signals respectively, and then carrying out selection and combination, wherein the output result is used for pseudo code tracking on one hand, and is demodulated to output remote control information on the other hand;
in the multi-set receiving step, a plurality of correlation operation channel processing is included, correlation operation is respectively carried out on the correlation operation channel processing and the local spread spectrum codes with different phases output in the pseudo code synchronizing step, the operation result is then subjected to selective combination processing, and the result of the selective combination processing is respectively output to pseudo code tracking processing and frequency locking loop differential demodulation processing;
in each correlation operation channel processing, the delay conjugation processing comprises delay processing and conjugation multiplication processing; the method comprises the steps that the results respectively output by N time delay conjugation processes in N correlation operation channel processes are sent to a selection combination process for energy comparison and selection, and signals with large signal-to-noise ratios are selected from the results and combined; the pseudo code tracking processing comprises phase discrimination processing and code loop filtering processing, and the frequency-locked loop differential demodulation processing comprises frequency-locked loop processing and differential demodulation processing.
2. The airborne measurement and control method of unmanned aerial vehicle of claim 1, wherein in the multi-set receiving step, a demodulation tracking carrier frequency offset value is further output, and is also fed back and input to a local carrier for frequency offset correction.
3. The airborne measurement and control method of the unmanned aerial vehicle of claim 1, wherein the pseudo code synchronization step comprises a first FFT transformation process, a second FFT transformation process, a conjugate multiplication process, an IFFT transform process and a capture judgment process; the first FFT conversion processing comprises receiving the baseband spread spectrum signal and carrying out FFT conversion on the baseband spread spectrum signal, and the second FFT conversion processing comprises receiving a local pseudo code sequence output by a local pseudo code generator and carrying out FFT conversion on the local pseudo code sequence; the output results of the two FFT conversion processes further complete complex conjugation and complex multiplication operations in the conjugate multiplication process, then are input into the IFFT conversion process for IFFT operation, the capture judgment process carries out capture judgment and identification on the IFFT operation results to obtain a pseudo code capture carrier frequency offset value and a pseudo code capture code phase value, the pseudo code capture carrier frequency offset value is output to carrier synchronization for carrier frequency offset correction, and the pseudo code capture code phase value is output to the local pseudo code generator for pseudo code phase correction.
4. The airborne measurement and control method of unmanned aerial vehicle of claim 3, wherein in the pseudo code synchronization step, data framing processing is further included before the first FFT transformation processing, and pseudo code zero padding framing processing is further included before the second FFT transformation processing.
5. The airborne measurement and control method of unmanned aerial vehicle of claim 4, wherein in the pseudo code synchronization step, the conjugate multiplication processing performs complex conjugate calculation on the result output by the second FFT processing, and then performs complex multiplication calculation on the result output by the first FFT processing.
6. The airborne measurement and control method of the unmanned aerial vehicle of claim 1, wherein the output expression of the baseband spread spectrum signal after the relevant operation processing is expressed as:
wherein,
representing the amplitude of one symbol in the received signal,
which represents a spreading code, is used to indicate,
indicates the sequence number of the spreading code,
the serial number of the relevant operation channel is shown,Included
a plurality of related operation channels are arranged in the parallel channel,
，
representing the carrier phase difference brought by the multipath time delay of each correlation operation channel;
in each correlation operation channel processing, after delaying one code phase by delay processing, the following steps are carried out:
wherein, T represents a symbol period delayed by one spreading code, and after conjugate multiplication, the output result is:
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