RU2546312C1 - Radio receiver for detecting phase-shift keyed broadband signals - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: radio receiver for detecting phase-shift keyed broadband signals comprises a preselector 1, a frequency converter 2, first 3 and second 17 intermediate frequency amplifiers, band-pass filters 4.i and 5.i, a nonlinear element 6.i, a narrow band-pass filter 7.i, an envelope detector 8.i, switches 9.i (i=1, 2, … n), a decision unit 10, an adder 11, a recording unit 12, first 13 and second 14 mixers, first 15 and second 16 heterodynes, a correlator 18, a threshold unit 19, a switch 20, a multiplier 21, first 22 and second 24 narrow band-pass filters, a phase doubler 23, a phase detector 25 and an inverted amplifier 26.

EFFECT: high noise-immunity and reliability of detecting phase-shift keyed broadband signals.

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Description

The proposed device relates to radio engineering and can be used in equipment designed to receive and analyze phase-shifted (PSK) signals with a binary phase value.

Known radio receivers for detecting signals against a background of noise and interference (ed. Certificate of the USSR No. 211.599, 309.326, 540.230, 1.718.695, 1.758.883, 1.785.410, 1.799.226, 1.799.227, 1.840.539, 1.840 .708; RF patents Nos. 2,001.533, 2.007.046, 2.181.528, 2.196.395, 2.379.837, 2.479.120; US patents Nos. 3,702.475, 3.815.028, 3.510.313, 7.742.914 ; EP patent No. 1.947.642; VA Sabinov, Digital device for detecting and coarse measuring the signal frequency. - Transactions of MAI, 1970, issue 201 and others).

Of the known devices, the closest to the proposed one is "Radio receiving device for detecting broadband signals with phase shift keying" (RF patent No. 2,479.120, H03K 7/00, 2011), which is selected as a prototype.

The known device provides the detection of broadband signals with phase shift keying against the background of noise and narrowband interference. It is built according to a superheterodyne circuit in which there are additional (mirror and Raman) reception channels. The suppression of false signals (interference) received via mirror and Raman channels is based on the use of two local oscillators 15 and 16, the frequencies f _G1 and f _{G2 of} which are spaced apart by a double value of the intermediate frequency

f _r2 -f _r1 = 2f _etc.

and selected symmetrical with respect to the frequency f _{c of the} main receiving channel, (Fig.2):

f _C -f _G1 = f _G2 -f _C = f _ave.

This circumstance leads to a doubling of the number of additional receiving channels, but creates favorable conditions for their suppression due to the correlation processing of channel voltages.

However, under the influence of various destabilizing factors, including the Doppler effect, when the radiation source of the FMN signals and the radio receiver are mutually moved, the indicated symmetry is broken and the noise immunity and reliability of detection of broadband signals with phase shift keying are reduced.

An object of the invention is to increase the noise immunity and reliability of detection of broadband signals with phase shift keying by ensuring the symmetry of the frequencies f _G1 and f _{G2 of the} first and second local oscillators relative to the frequency f _{C of the} main receiving channel

f _C -f _G1 = f _G2 -f _C = f _ave .

The problem is solved in that a radio receiving device for detecting broadband signals with phase shift keying, comprising, in accordance with the closest analogue, a series-connected preselector and a frequency converter, n channels of non-linear processing, each of which consists of a series-connected non-linear element, a narrow-band filter and a key as well as two band-pass filters, the outputs of which are connected to the inputs of the nonlinear element, while the output of the narrow-band filter through the envelope detector with it is single from one of the inputs of the deciding unit, the corresponding output of which is connected to the control input of the key, the outputs of the keys are connected to the inputs of the adder, the output of which is connected to the input of the registration unit, the frequency converter is made in the form of series-connected first local oscillator, first mixer, the second input of which is connected to the output of the preselector, the first intermediate frequency amplifier, correlator, threshold block and key, the second input of which is connected to the output of the first intermediate frequency amplifier, and the output is connected to the inputs of bandpass filters, serially connected to the second local oscillator, the second mixer, the second input of which is connected to the output of the preselector and the second intermediate frequency amplifier, the output of which is connected to the second input of the correlator, while the frequencies f _G1 and f _{G2 of the} first and second local oscillators are doubled intermediate frequency value

f _r2 -f _r1 = 2f _etc.

and are chosen symmetrical with respect to the carrier frequency f _{C of the} main receiving channel

f _C -f _G1 = f _G2 -f _C = f _ave. ,

differs from the closest analogue in that it is equipped with a multiplier, two narrow-band filters, a phase doubler, a phase detector and an inverse amplifier, with a multiplier connected in series to the output of the first local oscillator, the second input of which is connected to the output of the second local oscillator, the first narrow-band filter, phase detector and inverse an amplifier, the two outputs of which are connected to the outputs of the first and second local oscillators, respectively; a phase doubler is connected in series to the output of the frequency converter key the second narrow-band filter, the output of which is connected to the second input of the phase detector.

The structural diagram of the proposed device is presented in figure 1. A frequency diagram explaining the principle of formation of additional reception channels is shown in FIG.

A phase shift keyed radio receiver for detecting broadband signals comprises a preselector 1, a frequency converter 2 and n non-linear processing channels, each of which consists of a non-linear element 6.i, a narrow-band filter 7.i and a key 9.i, connected in series, and two band-pass filters 4.i and 5.i, the outputs of which are connected to the inputs of the nonlinear element 6.i, and the output of the narrow-band filter 7.i through the envelope detector 8.i is connected to one of the inputs of the decision unit 10, corresponding th output is connected to the control input 9.i keys, 9.i outputs connected to the outputs of the adder 11, whose output is connected to the input 12 of the recording unit (i = 1, 2, ..., n).

The frequency converter 2 is made in the form of series-connected first local oscillator 15, the first mixer 13, the second input of which is connected to the output of the preselector 1, the first intermediate frequency amplifier 3, the correlator 18, the threshold unit 19 and the key 20, the second input of which is connected to the output of the first amplifier 3 intermediate frequency, and the output is connected to bandpass filters 4.i and 5.i, serially connected to the second local oscillator 16, the second mixer 14, the second input of which is connected to the output of the preselector 1, and the second amplifier 17 intermediate frequency, the output of which is connected to the second input of the correlator 18, connected in series to the output of the first local oscillator 15 of the multiplier 21, the second input of which is connected to the output of the second local oscillator 16, the first narrow-band filter 22, phase detector 25 and inverse amplifier 26, the two outputs of which are connected to the inputs of the first 15 and second 16 local oscillators, respectively, connected in series to the output of the key 20 of the frequency converter 2 of the frequency doubler 23 and the second narrow-band filter 24, the output of which is connected to the second phase input Vågå detector 25.

The device operates as follows. If the broadband PSK signal is received on the main channel at a frequency f _C

u _C (t) = U _C cos [2πf _C t + φ _k (t) + φ _С ], 0≤t≤T _C ,

where φ _k (t) = {0, π} is the manipulated component of the phase that displays the law of phase manipulation,

then it goes from the output of the preselector 1 to the first input of the first 13 and second 14 mixers, the second input of which supplies the voltage of the first 15 and second 16 local oscillators, respectively

u _Г1 (t) = U _Г1 cos (2πf _Г1 t + φ _Г1 ),

u _Г2 (t) = U _Г1 cos (2πf _Г2 t + φ _Г2 ).

In this case, the frequencies f _G1 and f _{G2 are} spaced twice the value of the intermediate frequency (figure 2)

f _r2 -f _r1 = 2f _etc.

and are chosen symmetrical with respect to the frequency f _{C of the} main receiving channel

f _C -f _G1 = f _G2 -f _C = f _ave .

This circumstance leads to a doubling of the number of additional receiving channels, but creates favorable conditions for their suppression due to the correlation processing of channel voltages.

At the output of the mixers 13 and 14, voltages of combination frequencies are generated. Amplifiers 3 and 7 of the intermediate frequency are allocated voltage intermediate frequency:

u _pr1 (t) = U _pr1 cos [2πf _pr t + φ _k (t) + φ _pr1 ],

_np2 u (t) = U _np2 cos [2πf _{pr t-φ k (t) +} φ _np2], 0≤t≤T _C

;Where $$U_{PR\mathrm{one}}=\frac{\mathrm{one}}{2}{U}_{\mathrm{FROM}}{U}_{G\mathrm{one}}$$

;

; $$U_{PR2}=\frac{\mathrm{one}}{2}{U}_{\mathrm{FROM}}{U}_{G2}$$

;

f _CR = f _C -f _G1 = f _G2 -f _C is the intermediate frequency;

φ _pr1 = φ _C -φ _G1 ; φ _pr2 = φ _Г2 -φ _С ,

which enter the two inputs of the correlator 18, the output of which produces a voltage proportional to the correlation function R (τ).

It should be noted that the correlation function R (τ) of FMN signals has a remarkable property: it has a significant main lobe and a relatively low level of side lobes. This property is used to suppress false signals (interference) received via mirror and Raman channels. The voltage proportional to the correlation function R (τ) is compared with the threshold level U _por1 in the threshold block 19. The threshold voltage U _{por1 is} exceeded only at the maximum value of the correlation function R (τ). Since the correlation voltages u _pr1 (t) and u _pr2 (t) are formed by the same QPSK signal u _С (t) received through two channels at the same frequency f _С , there is a strong correlation between the indicated channel voltages communication. The correlation function R (τ) has a pronounced main lobe that exceeds the threshold level U _pores1 in the threshold block 19. When the threshold level U _pores1 is exceeded, a constant voltage is generated in the threshold block 19, which is supplied to the control input of the key 20, opening it.

полосы обработки и равныIn the initial state, the key 20 is always closed. In this case, the channel voltage of the intermediate frequency u _pr1 (t) from the output of the first amplifier 3 of the intermediate frequency through the public key 20 is supplied to n channels of nonlinear processing. In each i-th channel, two bandpass filters 4.i and 5.i are included in parallel, having a passband ΔF. Their center frequencies are located symmetrically with respect to the center frequency $$\frac{f_J}{\Delta f\ge \Delta F}$$

processing bands and equal

и $$\left[f_{о}+\left(i-\frac{1}{2}\right)\Delta f\right]$$

, $$\left[f_{\mathrm{about}}+\left(i-\frac{\mathrm{one}}{2}\right)\Delta f\right]$$

and $$\left[f_{\mathrm{about}}+\left(i-\frac{\mathrm{one}}{2}\right)\Delta f\right]$$

,

where the output signals of the band-pass filters of the multiplier in the nonlinear element 6.i and are filtered in the narrow-band filter 7i, the tuning frequency of which is 2f _o .

The output signals of the narrow-band filters 7i are summed in the adder 11, resulting in the formation of a second harmonic of the signal, the carrier frequency of which is equal to f _o . The second harmonic of the signal is fixed by the block 12 registration.

The channel into which the powerful narrow-band noise gets (the result of multiplying two narrow-band interference) is turned off by the decider 10 from the adder 11. In the decider 10, the presence of interference is determined by a significant excess of the signal level in the channel at the average channel level.

From the outputs of narrow-band filters 7.i, the signals through the keys 9.i (i = 1, 2, ..., n) are fed to the inputs of the adder 11, from the output of which the sum of the signals goes to the registration unit 12, where its level is compared with the threshold U _por1 . When the threshold is exceeded U _por2 signal detection is recorded.

From the outputs of the narrow-band filters 7.i, the signals are also sent to the envelope detectors 8.i, in which the envelopes of the signals supplied to the inputs of the decision block 10 are distinguished. In the decision block 10, the signal levels of all n channels are compared for channels in which the signal levels are significantly higher the average level across the channels, a control signal is generated that is supplied to the keys 9.i (i = 1, 2, ..., n) of these channels, as a result of which the keys are opened and the corresponding channels are disconnected from the adder 11.

The quantitative gain in noise immunity substantially depends on the level and number of interfering components.

To ensure symmetry

f _C -f _G1 = f _G2 -f _C = f _ave.

a phase locked loop (PLL) is used, consisting of a multiplier 21, a first narrow-band filter 22, a doubler 23 of a phase, a second narrow-band filter 24, a phase detector 25, and an inverse amplifier 26.

Voltages u _Г1 (t) and u _Г2 (t) from the outputs of the first 15 and second 16 local oscillators are supplied to two inputs of the first multiplier 21, at the output of which a harmonic voltage is generated

u ₁ (t) = U ₁ cos [2π (f _Г2 -f _Г1 + φ _Г ] = U ₁ cos (4πf _pr t + 2φ _pr ), 0≤t≤T _С ,

;Where $$U_{\mathrm{one}}=\frac{\mathrm{one}}{2}{U}_{G\mathrm{one}}{U}_{G2}$$

;

f _r2 -f _r1 = 2f _straight;

φ _U = φ -φ _{r2 r1} = 2φ _prosp,

which is allocated by the first narrow-band filter 22 and supplied to the first input of the phase detector 25. The voltage u _pr1 (t) from the output of the first intermediate frequency amplifier 3 is supplied through the open switch 20 to the input of the phase doubler 23, at the output of which a harmonic voltage is generated

u ₂ (t) = U ₂ cos [4πf _pr t + 2φ _k (t) + 2φ _pr1 ] = U ₂ cos (4πf _pr t + φ _pr1 ), 0≤t≤T _С ,

;Where $$U_2=\frac{\mathrm{one}}{2}2U_{PR\mathrm{one}}$$

;

2φ _l (t) = {0.2π},

which is allocated by the second narrow-band filter 24 and fed to the second input of the phase detector 25.

If this symmetry is broken, then a control voltage is generated at the output of the phase detector 25. Moreover, the amplitude and polarity of the control voltage depend on the degree and direction of deviation of the carrier frequency f _C from the frequencies f _G1 and f _{G2 of the} first 15 and second 16 local oscillators. The specified voltage through the inverse amplifier 26 acts on the control inputs of the first 15 and second 16 local oscillators so that the symmetry condition is satisfied

f _C -f _G1 = f _G2 -f _C = f _ave.

The operation of the device described above corresponds to the case of receiving useful QPSK signals on the main channel at a frequency f _C (FIG. 2).

If a complex signal (interference) is received through the first mirror channel at a frequency f _З1 , then in the first 13 and second 14 mixers it is converted to voltages of the following frequencies:

f ₁₁ = f _Г1 -f _З1 = f _ol ; f ₁₂ = f _Г2 -f _З1 -f _ave .

However, only voltage with a frequency of f ₁₁ falls into the passband of the first intermediate frequency amplifier 3. The output signal of the correlator 18 is equal to zero, the key 20 does not open, and a false signal (interference) received on the first mirror channel at a frequency f _Z1 is suppressed.

If a false signal (interference) is received through the second mirror channel at a frequency f _З2 , then in the first 13 and second 14 mixers it is converted to voltages of the following frequencies:

f ₂₁ = f _r1 -f _P2 = 3f _etc., f ₂₂ = f _S2 = f _r2 -f _pr.

However, only voltage with a frequency f ₂₂ fall into the passband of the second intermediate-frequency amplifier 17, the output signal of the correlator 18 is also zero, the key 20 does not open, and a false signal (interference) received through the second mirror channel at a frequency f _Z2 is suppressed.

For a similar reason, a false signal (interference) received on the first combinational channel at a frequency f _k1 , on a second combinational channel at a frequency f _k2, or on any other additional channel, is also suppressed.

If false signals (interference) are simultaneously received through the first f _Z1 and second f _Z2 mirror channels, then in the first 13 and second 14 mixers they are converted to voltages of the following frequencies:

f ₁₁ = f _Г1 -f _З1 = f _ol ; f ₁₂ = f _Г2 -f _З1 -f _ave .

f ₂₁ = f _r1 -f _P2 = 3f _etc., f ₂₂ = f _S2 = f _r2 -f _pr.

In this case, the voltage with frequencies f ₁₁ and f ₂₂ fall into the passband of the first 3 and second 17 amplifiers of intermediate frequency, and then fed to the two inputs of the correlator 18. But the key 20 in this case does not open. This is because different false signals (interference) are received at different frequencies f _Z1 and f _Z2 , so there is a weak correlation between channel voltages with frequencies f ₁₁ and f ₂₂ .

In addition, it should be noted that the correlation function of interference does not have a pronounced main lobe, as is the case with complex PSK signals. The output voltage of the correlator 18 in this case does not exceed the first level U _pore1 in the threshold block 19, the key 20 does not open and false signals (interference) received simultaneously on the first f _Z1 and second f _Z2 mirror channels are suppressed.

For a similar reason, false signals (interference), received simultaneously on the first Raman channel at a frequency of f ₁₁ and the second Raman channel at a frequency of f ₂₂ , or any other additional channels, are also suppressed.

Thus, the proposed device in comparison with the prototype provides increased noise immunity and reliability of detection of broadband signals with phase shift keying. This is achieved by ensuring the symmetry of the frequencies f _G1 and f _{G2 of the} first and second local oscillators relative to the frequency f _{C of the} main receiving channel f _C -f _G1 = f _G2 -f _C = f _pp. through the use of a phase locked loop.

Claims (1)

A radio receiver for detecting broadband signals with phase shift keying, containing serially connected preselector and frequency converter, n channels of non-linear processing, each of which consists of sequentially connected non-linear element, narrow-band filter and key, as well as two band-pass filters, the outputs of which are connected to non-linear inputs element, while the output of the narrow-band filter through the envelope detector is connected to one of the inputs of the decision unit, the corresponding output of which is it is connected to the control input of the key, the outputs of the keys are connected to the inputs of the adder, the output of which is connected to the input of the registration unit, the frequency converter is made in the form of series-connected first local oscillator, first mixer, the second input of which is connected to the output of the preselector, the first intermediate frequency amplifier, correlator, threshold block and key, the second input of which is connected to the output of the first intermediate frequency amplifier, and the output is connected to the inputs of bandpass filters, sequentially connected to the second get the oscillator, the second mixer, the second input of which is connected to the output of the preselector, and the second amplifier of the intermediate frequency, the output of which is connected to the second input of the correlator, while the frequencies f _Г1 and f _{Г2 of the} first and second local oscillators are spaced apart by a double value of the intermediate frequency
f _r2 -f _r1 = 2f _etc.
and are chosen symmetrical with respect to the carrier frequency f _{C of the} main receiving channel
f _C -f _G1 = f _G2 -f _C = f _ave. ,
characterized in that it is equipped with a multiplier, two narrow-band filters, a phase doubler, a phase detector and an inverse amplifier, with a multiplier connected in series to the output of the first local oscillator, the second input of which is connected to the output of the second local oscillator, the first narrow-band filter, a phase detector and an inverse amplifier, two the output of which is connected to the inputs of the first and second local oscillators, respectively, the phase doubler and the second narrow-band phi are connected in series to the output of the frequency converter key tr, the output of which is connected to the second input of the phase detector.

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