RU2541723C1 - Precision analogue-digital interface for working with resistive micro- and nanospheres - Google Patents

Abstract

FIELD: measurement equipment.

SUBSTANCE: invention relates to the field of measurement equipment and may be used in structure of various sensor systems, which use resistive sensors that vary their resistance under physical impact of environment (pressure, deformation, light, temperature, radiation, composition of various gases, moisture, etc.). The device comprises a measurement bridge, the first (1) output of the power supply diagonal of which is connected to the first (2) bus of the power supply source, the second (3) output of the power supply diagonal is connected to the second (4) common bus of the power supply source, and the first (5) and second (6) outputs of the measurement diagonal are connected to inputs of the first (7) differential instrument amplifier (DIA), the first (8) resistive sensor connected between the first (5) output of the measurement diagonal and the first (1) output of the power supply diagonal, the second (9) resistive sensor connected between the first (5) output of the measurement diagonal and the second (3) output of the power supply diagonal, the third (10) resistive sensor connected between the second (6) output of the measurement diagonal and the first (1) output of the power supply diagonal, the fourth (11) resistive sensor connected between the second (6) output of the measurement diagonal and the second (3) output of the power supply diagonal, the first (12) and second (13) auxiliary resistors, connected in series between the output (14) of the DIA (7) and the non-inverting input of the auxiliary OA (15), the inverting input of which is connected to the output (16) of this auxiliary OA (15), the first (17) correcting capacitor connected between the common unit (18) of the first (12) and second (13) auxiliary resistors and the output (16) of the auxiliary OA (15), the second (19) correcting capacitor connected between the non-inverting input of the auxiliary OA (15) and the second (4) common bus of the power supply bus, the first (20) ADC, the input of which is connected with the outlet (16) of the auxiliary OA (15). The circuit includes an additional DIA (21), the output of which (22) is connected to the input of the second (23) ADC, the first (24) input of the additional DIA (21) is connected to the common unit (18) of the first (12) and second (13) auxiliary resistors, and the second (25) input of the additional DIA (21) is connected to the non-inverting input of the auxiliary OA (15).

EFFECT: possibility to create both a digital equivalent of an input measured value (x) and a digital equivalent of its first derivative $$(\dot{x}),$$

and also production of a digital temperature value of sensors.

3 cl, 17 dwg

Description

The present invention relates to the field of measurement technology and can be used in the structure of various sensor systems that use resistive sensors that change their resistance under the physical influence of the environment (pressure, deformation, light, temperature, radiation, the composition of various gases, humidity, etc. .).

To measure the parameters of gaseous media, temperature, bending, deformations of various parts, sensitive elements are resistive micro- and nanosensors, which are included in the structure of the so-called measuring bridges [1-11]. This technical solution, as a rule, involves the use of precision measuring amplifiers, which are connected through the low-pass filters to the input of analog-to-digital converters [1-6] or digital signal processing devices. Such an architecture is classical [1-11].

The closest prototype of the claimed device is the analog-to-digital interface of figure 1, presented in patent US 4.484.146. It contains a measuring bridge, the first 1 output of the power diagonal of which is connected to the first 2 bus of the power source, the second 3 output of the power diagonal is connected to the second 4 common bus of the power source, and the first 5 and second 6 outputs of the measuring diagonal are connected to the inputs of the first 7 differential instrumental amplifier , the first 8 resistive sensor connected between the first 5 output of the measuring diagonal and the first 1 output of the power diagonal, the second 9 resistive sensor connected between the first 5 output of the measuring diagonal and the second 3 output of the power diagonal, the third 10 resistive sensor connected between the second 6 output of the measuring diagonal and the first 1 output of the power diagonal, the fourth 11 resistive sensor connected between the second 6 output of the measuring diagonal and the second 3 output of the power diagonal, the first 12 and 13 auxiliary resistors connected in series between the output 14 of the differential instrumental amplifier 7 and the non-inverting input of the auxiliary operational amplifier 15, the inverting input of which is connected to the output m 16 of this auxiliary operational amplifier 15, the first 17 correction capacitor included between the common node 18 of the first 12 and second 13 auxiliary resistors and the output 16 of the auxiliary operational amplifier 15, the second 19 correction capacitor connected between the non-inverting input of the auxiliary operational amplifier 15 and the second 4 common bus power source, the first 20 analog-to-digital Converter, the input of which is connected to the output 16 of the auxiliary operational amplifier 15.

A significant disadvantage of the known device of figure 1 is that it does not provide the formation of a signal proportional to the derivative of the input measured value. This does not allow the use of this technical solution in new and promising adaptive control systems, for the effective functioning of which it is necessary to have information about the rate of change of the input signal (its derivative).

In addition, the known circuit is characterized by a nonlinear temperature dependence of the output signals, which is associated with the instability of the properties of micro- and nanosensors when exposed to this destabilizing factor.

), а также получении цифрового значения температуры сенсоров. Данная информация может использоваться в дальнейшем для введения соответствующих коррекций в измерительные характеристики конкретной датчиковой системы, которые на практике реализуются микропроцессорами.The main objective of the invention is to form not only the digital equivalent of the input measured quantity (x), but also the digital equivalent of its first derivative ( $$\dot{x}$$

), as well as obtaining a digital temperature value of the sensors. This information can be used in the future to introduce appropriate corrections into the measuring characteristics of a particular sensor system, which are in practice implemented by microprocessors.

The problem is achieved in that in a precision analog-to-digital interface for working with resistive micro- and nanosensors of Fig. 1, containing a measuring bridge, the first 1 output of the power diagonal of which is connected to the first 2 bus of the power source, the second 3 output of the diagonal of power is connected to the second 4 by a common power supply bus, and the first 5 and second 6 outputs of the measuring diagonal are connected to the inputs of the first 7 differential instrumental amplifier, the first 8 is a resistive sensor connected between the first 5 output of the meter diagonal and the first 1 output of the power diagonal, the second 9 resistive sensor connected between the first 5 output of the measuring diagonal and the second 3 output of the power diagonal, the third 10 resistive sensor connected between the second 6 output of the measuring diagonal and the first 1 output of the power diagonal, the fourth 11 resistive a sensor connected between the second 6 output of the measuring diagonal and the second 3 output of the power diagonal, the first 12 and second 13 auxiliary resistors connected in series between the output 14 of the differential instrument mental amplifier 7 and a non-inverting input of the auxiliary operational amplifier 15, the inverting input of which is connected to the output 16 of this auxiliary operational amplifier 15, the first 17 correction capacitor connected between the common node 18 of the first 12 and second 13 auxiliary resistors and the output 16 of the auxiliary operational amplifier 15, the second 19 correction capacitor connected between the non-inverting input of the auxiliary operational amplifier 15 and the second 4 common bus power supply, the first 20 the analog-to-digital converter, the input of which is connected to the output 16 of the auxiliary operational amplifier 15, new elements and connections are provided - an additional differential instrument amplifier 21 is introduced into the circuit, the output of which 22 is connected to the input of the second 23 analog-to-digital converter, the first 24 input of the additional differential the amplifier 21 is connected to a common node 18 of the first 12 and second 13 auxiliary resistors, and the second 25 input of an additional differential instrumental amplifier Ithel 21 is connected to the noninverting input of an auxiliary operational amplifier 15.

The drawing of figure 1 shows a diagram of a precision analog-to-digital interface of the prototype.

The drawing of figure 2 presents a diagram of the inventive device in accordance with claim 1 of the claims.

The drawing of figure 3 presents a diagram of the inventive device in accordance with claim 2 of the claims.

The drawing of figure 4 presents a diagram of the inventive device in accordance with claim 3 of the claims.

The drawing of figure 5 shows an example of the construction of the main functional nodes of the circuit of figure 4 using the so-called multi-differential operational amplifiers, the circuitry of which is widely represented in the modern technical literature [12].

The drawing of Fig.6 shows the amplitude-frequency characteristic (AFC) of the measurement channel in the operating frequency range of a physical quantity acting on the sensors.

The drawing of Fig.7 shows the phase-frequency characteristic (PFC) of the measuring channel of a physical quantity in the operating frequency range, demonstrating high linearity and, therefore, low measurement error of the shape of the corresponding signal.

The drawing of Fig.8 shows the frequency error of the phase response of the channel for measuring physical quantities in the range of workers.

The drawing of Fig.9 shows the dependence of the error of the phase response of the measuring channel of the derivative on the differential transmission coefficient (Kd) of the additional differential instrumental amplifier 21 of the measuring channel of the derivative of a physical quantity.

The drawing of figure 10 shows the effect of the deviation of the capacitance of the third 28 correction capacitor (± 1%) on the non-uniformity of the frequency response of the measurement channel of a physical quantity.

The drawing of Fig.11 shows the effect of the deviation of the capacitance of the third 28 correction capacitor (± 1%) on the phase response error of the measurement channel of a physical quantity.

The drawing of Fig. 12 shows the effect of the deviation h (± 0.5%) on the non-uniformity of the frequency response of the physical quantity measuring channel, where h is the ratio of the capacitances of the first 17 and third 28 correction capacitors.

The drawing of Fig.13 shows the effect of the deviation of the parameter h (± 0.5%) on the phase response error of the measurement channel of a physical quantity.

The drawing of Fig.14 shows the frequency response of the measurement channel of the derivative of the measured physical quantity.

The drawing of Fig. 15 shows the phase response of a measuring channel of a derivative of a measured physical quantity with a transmission coefficient Kd = 15.208, where Kd is the differential transmission coefficient of an additional differential instrumentation amplifier 21.

In the drawing of Fig.16 shows the simulation results of the measurement channel of a physical quantity in the time domain.

In the drawing of Fig.17 shows the results of modeling the channel for measuring the derivative in the time domain.

The precision analog-to-digital interface for working with resistive micro- and nanosensors of FIG. 2 contains a measuring bridge, the first 1 output of the power diagonal of which is connected to the first 2 bus of the power source, the second 3 output of the diagonal of power is connected to the second 4 common bus of the power source, and the first 5 and second 6 outputs of the measuring diagonal are connected to the inputs of the first 7 differential instrumental amplifier, the first 8 resistive sensor connected between the first 5 output of the measuring diagonal and the first 1 output of the diagonal the second 9 resistive sensor connected between the first 5 output of the measuring diagonal and the second 3 output of the power diagonal, the third 10 resistive sensor connected between the second 6 output of the measuring diagonal and the first 1 output of the power diagonal, the fourth 11 resistive sensor connected between the second 6 output measuring diagonal and the second 3 output diagonal of the power supply, the first 12 and second 13 auxiliary resistors connected in series between the output 14 of the differential instrumental amplifier 7 and non-inverting input the house of the auxiliary operational amplifier 15, the inverting input of which is connected to the output 16 of this auxiliary operational amplifier 15, the first 17 correction capacitor included between the common node 18 of the first 12 and second 13 auxiliary resistors and the output 16 of the auxiliary operational amplifier 15, the second 19 correction capacitor included between the non-inverting input of the auxiliary operational amplifier 15 and the second 4 common bus power supply, the first 20 analog-to-digital Converter, the input of which о is connected to the output 16 of the auxiliary operational amplifier 15. An additional differential instrumentation amplifier 21 is introduced into the circuit, the output of which 22 is connected to the input of the second 23 analog-to-digital converter, the first 24 input of the additional differential instrumentation amplifier 21 is connected to a common node 18 of the first 12 and second 13 auxiliary resistors, and the second 25 input of the additional differential instrumentation amplifier 21 is connected to the non-inverting input of the auxiliary operational amplifier i'm 15.

In the drawing of FIG. 2, a low-ohmic auxiliary resistor can be connected in series with the third 10 resistive sensor, providing a given level of asymmetry of the measuring bridge.

передаются в микропроцессор для последующей обработки.Digital equivalents of the input measured quantity D1.x ... Dn.x and its derivatives $$D\mathrm{one.}\dot{x}...Dn.\dot{x}$$

transferred to the microprocessor for further processing.

In the drawing of FIG. 3, in accordance with claim 2, between the output 14 of the first 7 differential instrumentation amplifier and the first 26 output of the first 12 auxiliary resistor not connected to the second 13 auxiliary resistor, a third 27 auxiliary resistor is connected, and between the first 26 the output of the first 12 auxiliary resistor and the second 4 common bus power supply included the third 28 correction capacitor.

In the drawing of figure 4, in accordance with paragraph 3 of the claims, a temperature sensor 29 of the first 8, second 9, third 10 and fourth 11 resistance sensors is connected to the input of the temperature-voltage measuring transducer 30, the output of which is connected with the input of the low-pass filter 31, and the output 32 of the low-pass filter 31 is connected to the input of the third 33 analog-to-digital Converter.

The drawing of figure 5 shows an example of practical construction of the inventive device on a modern element base. Here, the differential instrumentation amplifier 7 is implemented on the basis of a multidifferential op-amp (MOA) [12] and includes feedback resistors 34, 35. An additional differential instrumentation amplifier 21 is implemented on the MOA 36 and feedback resistors 37, 38. A temperature-voltage measuring transducer 30 made on the basis of resistors 39, 40, 41, 43 and operational amplifier 42. The low-pass filter 31 is implemented in accordance with figure 2 and contains resistors 44, 45, operational amplifier 46, capacitors 47 and 48. Input 5, 6 and output 22 , 16, 32, the nodes of the circuit of FIG. 5 have the same designations as the corresponding nodes of the circuit of FIG. 4.

Consider the operation of the device of figure 2.

The influence of the measured physical quantity on the resistance of the resistive sensors 8-10 leads to a change in the differential voltage at the outputs 5, 6 of the measuring diagonal of the bridge and at the corresponding inputs of the differential instrument amplifier 7. Due to the identity of the resistances of the resistors 8-10 (micro- or nanosensors) in-phase voltages, caused by the action of the reference voltage source 2 at the same inputs are identical. Isolation and amplification by differential tool amplifier 7 of differential voltage is accompanied by a weakening of the common-mode signal to a level corresponding to the methodological accuracy of the first 20 and second 23 ADCs. Thus, the input of the spectrum limiter (elements 12, 13, 15, 17, 19) receives an amplified differential voltage proportional to the measured physical quantity. Along with the differential voltage, the differential instrumental amplifier 7 amplifies the intrinsic noise of the sensors 8-10 of the measuring bridge, which during the analog-to-digital conversion leads to the appearance of difference spectral components between the sampling frequency and the frequencies amplified by the differential instrumental amplifier 7 noise components of the total spectrum of the measured process. To reduce the amplitudes of these difference components, a third-order low-pass filter (low-pass filter, spectrum limiter) is used, implemented on the first 12 and second 13 auxiliary resistors, the first 17 and second 19 correction capacitors and auxiliary operational amplifier 15. The presence of the feedback circuit by connecting the first 17 correction capacitor to the output of the auxiliary operational amplifier 15 allows you to effectively use the order of this filter in the transition frequency range by increasing its "direct ougolnosti "and, therefore, reduce the error transform of spectral components in the bandwidth (range of operating frequencies of the sensors).

By connecting the second 13 auxiliary resistor to the differential inputs of the additional differential instrumentation amplifier 21, it is possible to isolate the differential component of the measured physical quantity. Moreover, the operating frequency range for this component will be determined by the passband of the low-pass filter (elements 12, 13, 15, 17, 19).

Indeed, as follows from the circuit of FIG. 2, the voltage at the second correction capacitor 19 and at the output of the auxiliary operational amplifier 15 is determined by the integral of the current flowing through the second 13 auxiliary resistor. Provided that the input resistance of the auxiliary operational amplifier 15 is significantly greater than the resistance of the second 13 auxiliary resistor, the voltage drop across this resistor will correspond to the voltage differential at the output 16.

Mathematical analysis of the proposed device is feasible for the interface of figure 3, because in the absence of the third 27 auxiliary resistor and the third 28 correction capacitor (the output of the differential instrumentation amplifier 7 is directly connected to the first 12 auxiliary resistor), the frequency filtering channel shown in FIG. 2 is implemented. Using the method of analysis of linear electronic circuits, it can be shown that the following transfer functions are realized at outputs 16 and 22

where K ₇ , K ₂₁ - differential gains of differential instrumental amplifiers 7 and 21;

R ₁₃ , C ₁₉ - resistance and capacitance of circuit elements 13 and 19.

In these relations, the coefficients a _{i of the} transfer functions are determined as follows

;where R _ij , C _ij are resistances and capacitances of circuit elements with number ij $$(i=\overline{\mathrm{one},2},j=\overline{\mathrm{one},2})$$

;

It follows from relations (3) that, due to the additive principle of the formation of all the transfer function coefficients (1) and (2), the channels for measuring a physical quantity and its derivative are characterized by low parametric sensitivity

In addition, the requirement of a small non-uniformity of the frequency response of the spectrum limiter in the passband (the working frequency range of the sensors), in accordance with the properties of their approximating functions (Gauss, Butterworth, Chebyshev), is associated with the implementation of a low Q factor of the dominant pole (1.0-1.5 units) . That is why the proposed solutions to the problem provide low parametric sensitivity to all elements of the circuit.

In this regard, in the practical implementation of the interface, additional parametric conditions can be used:

Then

Note that relations (6), up to the ratios of the nominal values of the same type of elements, correspond to the structure of the low-pass filter (theoretically optimal in terms of parametric sensitivity) of the low-pass filter. This conclusion is confirmed by the results of modeling a practical interface circuit shown in the drawings of Figures 10-13. Moreover, it can be rigorously shown that the last conclusion is valid when the inequality

where f ₁ is the frequency of unity gain of the amplifier 15;

f _c is the operating frequency range of the sensor (sensitive element). From the transfer functions (2) and (1) (due to the properties of the Laplace transform) it follows that

where u _out.16 (t) and u _out.22 (t) are the output voltages at nodes 16 and 22. Thus, the proposed structures and circuit diagrams of the interfaces provide not only high accuracy of measuring a physical quantity, but also a measurement (or estimate) its derivative. This conclusion is shown in the drawings of Fig.6 and Fig.7. Here, assessment is understood in the general case as inconsistency in the duration of transients (functions (1) and (2)) of the considered interface channels. It also follows from relation (8) that the absence of difference terms preserves the low parametric sensitivity of the derivative estimation channel

-ой гармонической составляющей входного сигнала и, следовательно, измеряемой величины, максимальное отклонение определяется из соотношенияThe resulting accuracy of the measurement (estimation) of the derivative of the measured quantity is also affected by the phase error of the conversion channel, which according to (1) is due to the differentiation of the signal only in the passband of the low-pass filter (Fig. 14). It can be shown quite strictly that for $$\ell$$

th harmonic component of the input signal and, therefore, the measured quantity, the maximum deviation is determined from the ratio

- амплитудное значение производной l-й гармонической составляющей;Where $$U_{\ell \max }^{\text{'}\text{'}}$$

- the amplitude value of the derivative of the l-th harmonic component;

, $$\Delta {\varphi }_{\ell }$$

- круговая частота и паразитный фазовый сдвиг $$\ell$$

-й гармонической составляющей. $${\omega }_{\ell }$$

, $$\Delta {\varphi }_{\ell }$$

- circular frequency and spurious phase shift $$\ell$$

-th harmonic component.

) возможно путем уменьшения эффективной полосы пропускания ФНЧ и, следовательно, при заданной (требуемой) селективности путем повышения порядка его передаточной функции. ФЧХ, приведенные на фиг.8 и фиг.15, наглядно демонстрируют этот вывод. К этому же результату приводит и дополнительная погрешность дополнительного дифференциального инструментального усилителя 21 (фиг.9), связанная с влиянием на ФЧХ коэффициента ослабления входного синфазного напряжения. Результаты моделирования практической схемы интерфейса, подтверждающие этот вывод, показаны на чертежах фиг.16 и фиг.17. При этом, как видно из временных диаграмм фиг.17, для неизменной производной (базовый случай при измерении физических величин) погрешность ее оценки связана переходными процессами в канале ее измерения.From the above relation it is seen that the maximum error corresponds to the absolute minimum of the derivative, and the increase in accuracy (decrease $$\Delta {\varphi }_{\ell }$$

) is possible by reducing the effective passband of the low-pass filter and, therefore, at a given (required) selectivity by increasing the order of its transfer function. The phase response curves shown in Fig. 8 and Fig. 15 clearly demonstrate this conclusion. The additional error of the additional differential instrumentation amplifier 21 (Fig. 9) leads to the same result, due to the influence of the attenuation coefficient of the input common-mode voltage on the phase response curve. The simulation results of the practical interface circuitry, confirming this conclusion, are shown in the drawings of Fig.16 and Fig.17. Moreover, as can be seen from the time diagrams of FIG. 17, for an unchanged derivative (the base case when measuring physical quantities), the error of its estimation is associated with transients in the channel of its measurement.

From the graphs of Fig. 7 in particular, it follows that the implementation of the linear phase response provides high accuracy for measuring short-term (pulsed) input actions.

Thus, the claimed device is characterized by relatively small values of the errors of measurement of a physical quantity and evaluation of its derivative.

The analysis performed above, as well as the results of computer simulations show that in the proposed precision analog-to-digital interface (Fig. 2) one of the problems of modern measuring equipment is solved - obtaining the digital equivalent of the derivative of the measured physical quantity, the information about which significantly expands the possibilities of building on its basis adaptive control systems for various objects, as well as the digital equivalent temperature of sensors, the latter property allows you to enter using a microprocessor n The necessary correction of temperature errors in measuring a physical quantity. In addition, the circuit can provide diagnostics of the state of resistive nanosensors 8-10.

Information sources

1. Patent RU No. 2.247.325

2. Patent RU No. 2.380.714

3. Patent RU No. 2.265.229

4. US patent No. 8.330.537

5. Application for US patent No. 2012/01860091

6. EP patent No. 1,703.262

7. US patent No. 4.063.447

8. Patent SU No. 1.830.463

9. Patent RU No. 2,304.284

10. Application for US patent No. 2001/0035758

11. Application for patent US No. 2003/0916033

12. Multidifferential operational amplifier in the mode of an instrument amplifier [Text] / S.G. Krutchinsky, Titov A.E. // Scientific and technical statements of SPbSPU “Informatics, Telecommunications and Management”, No. 3 (101), 2010. - P.200-204.

Claims (3)

1. A precision analog-digital interface for working with resistive micro- and nanosensors, comprising a measuring bridge, the first (1) output of the power diagonal of which is connected to the first (2) bus of the power source, the second (3) output of the diagonal of power is connected to the second (4 ) a common power supply bus, and the first (5) and second (6) outputs of the measuring diagonal are connected to the inputs of the first (7) differential instrument amplifier, the first (8) resistive sensor connected between the first (5) output of the measuring diagonal and the first (1 ) with dia drove the power supply, the second (9) resistive sensor connected between the first (5) output of the measuring diagonal and the second (3) output of the power diagonal, the third (10) resistive sensor connected between the second (6) output of the measuring diagonal and the first (1) output power diagonals, a fourth (11) resistive sensor connected between the second (6) output of the measuring diagonal and the second (3) output of the power diagonal, the first (12) and second (13) auxiliary resistors connected in series between the output (14) of the differential instrumental amplifier (7) and the non-inverting input of the auxiliary operational amplifier (15), the inverting input of which is connected to the output (16) of this auxiliary operational amplifier (15), the first (17) correction capacitor connected between the common node (18) of the first (12) and second (13) auxiliary resistors and the output (16) of the auxiliary operational amplifier (15), the second (19) correction capacitor connected between the non-inverting input of the auxiliary operational amplifier (15) and the second (4) common bus of the power source, the first (20) analog a digital converter, the input of which is connected to the output (16) of the auxiliary operational amplifier (15), characterized in that an additional differential instrument amplifier (21) is introduced into the circuit, the output of which (22) is connected to the input of the second (23) analog-to-digital converter, the first (24) input of the additional differential instrumentation amplifier (21) is connected to a common node (18) of the first (12) and second (13) auxiliary resistors, and the second (25) input of the additional differential instrumentation amplifier (21) under for prison staff to the non-inverting input of the auxiliary operational amplifier (15). 2. A precision analog-to-digital interface for working with resistive micro- and nanosensors according to claim 1, characterized in that between the output (14) of the first (7) differential instrument amplifier and the first (26) output of the first (12) auxiliary resistor connected to the second (13) auxiliary resistor, a third (27) auxiliary resistor is connected, and a third (28) correction capacitor is connected between the first (26) terminal of the first (12) auxiliary resistor and the second (4) common power supply bus. 3. The precision analog-to-digital interface for working with resistive micro- and nanosensors according to claim 1, characterized in that a temperature sensor (29) of the first (8), second (9), third (10) and fourth (11) is introduced into the circuit ) resistive sensors associated with the input of the temperature-voltage measuring transducer (30), the output of which is connected to the input of the low-pass filter (31), and the output (32) of the low-pass filter (31) is connected to the input of the third (33) analog digital converter.

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