RU2518428C2 - Direction finding phase method and phase direction finder for implementing said method - Google Patents

Abstract

FIELD: physics, navigation.

SUBSTANCE: disclosed method and apparatus relate to radioelectronics and can be used to determine coordinates of radiation sources of composite signals with combined phase and frequency keying, mounted on-board an aircraft (airplane, helicopter, airship, probe etc) and determine parameters thereof. The phase direction finder which realises the disclosed direction finding phase method comprises receiving antennae, three receivers, a reference generator, a pulse generator, an electronic switch, two 90° phase changers, eight phase detectors, an indicator, a heterodyne, a mixer, an intermediate frequency amplifier, four multipliers, three band-pass filters, a delay line, two squaring devices, a scaling multiplier, a subtractor, a phase doubler, three phase-locked loop units, two phase halvers, three narrow band-pass filters, a frequency demodulator, an adder and a recording unit, connected to each other in a certain manner.

EFFECT: broader functional capabilities of the existing method and apparatus through accurate and unique determination of the azimuth and elevation angle of the radiation source of a composite signal with combined phase and frequency keying, mounted on-board an aircraft, and synchronous detection thereof.

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Description

The proposed method and device relates to the field of electronics and can be used to determine the coordinates of the radiation sources of complex signals with combined phase and frequency manipulation (FMN-FMN), placed on board the aircraft (aircraft, helicopter, airship, probe, etc.) , and definitions of their parameters.

Known phase methods of direction finding and phase direction finders (RF patents Nos. 2,003.131, 2.006.872, 2.010.258, 2.012.010, 2.134.429, 2.155.352, 2.175.770, 2.290.658, 2.296.432, 2.303. 274, 2.311.656, 2.365.931, 2.427.853 ;; US patents Nos. 4,380.010, 7.084.812 ;; UK patents Nos. 1,395.599, 1,598.325; German patents Nos. 2,127.087, 2.710. 955; Kinkulkin I.E. et al. Phase method for determining coordinates. M: Sov. Radio, 1979 and others).

Of the known methods and devices closest to the proposed are the "Phase method of direction finding and phase direction finder for its implementation" (RF patent No. 2.427.853, G01S3 / 46.2010), which are selected as prototypes.

Known technical solutions are invariant to instability of the carrier frequency of the received signals, the type of modulation (manipulation) and the width of the spectrum, and the exact and unambiguous measurement of the angular coordinates α (azimuth) and β (elevation angle) of the radiation source of the signal placed on board the aircraft (aircraft, helicopter, airship, probe, etc.), is carried out at a stable frequency Ω of the reference generator.

However, the known technical solutions do not allow measuring the angular coordinates α and β of the radiation source of a complex signal with combined phase and frequency manipulation (QPSK-FSK) located on board the aircraft, and carry out its synchronous detection. These signals are widely used in various electronic devices placed on board aircraft.

An object of the invention is to expand the functionality of the known method and device by accurately and unequivocally determining the azimuth and elevation angle of the radiation source of a complex signal with combined phase and frequency manipulation, placed on board the aircraft, and its synchronous detection.

The problem is solved in that the phase direction finding method, based, in accordance with the closest analogue, on the fact that they receive signals, amplify and limit them in amplitude, compare signals that have passed two channels in phase, while the signal of one of the channels is pre-shifted in phase by 90 °, n receiving antennas are installed in the azimuthal plane n of a circle of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receiving antennas are switched, times displaced around the circle, alternately with frequency Ω, the signal received by the antenna located in the center of the circle is converted in frequency using the local oscillator frequency, the intermediate frequency voltage is extracted, it is multiplied with signals alternately received by n receiving antennas located around the circle, phase-modulated voltage is isolated multiply it with the local oscillator voltage, isolate the first low-frequency voltage with a frequency of Ω and compare it in phase with the reference voltage, forming an accurate, but not unique the direction finding scale of the signal source in the azimuthal plane, the phase-modulated voltage is subjected to autocorrelation processing at the same time, a second low-frequency voltage is extracted with a frequency Ω, it is compared in phase with the reference voltage, forming a rough but unambiguous direction-finding scale for the signal source in the azimuthal plane, set in the elevation plane the second receiving antenna at a distance d ₂ from the first receiving antenna, receive a signal on it, amplify and limit it in amplitude multiply with the voltage of the intermediate frequency, isolate the harmonic voltage at the local oscillator frequency, multiply it with the voltage of the local oscillator, isolate the voltage proportional to the phase difference between the signals received by the first and second receiving antennas, forming a rough but unambiguous direction finding scale for the signal source in the elevation plane, the indicated voltage is squared, multiplied with the initial voltage, at the same time the initial voltage is proportional to the phase difference between the signals received with the first and second antennas, they are phase shifted 90 °, squared, multiplied with the generated product using a scaling factor of three, and the resulting product is subtracted from the generated product, forming an accurate but ambiguous direction finding scale for the signal source in the elevation plane differs from the closest analogue in that they double the phase of the received signal with combined phase and frequency manipulation at an intermediate frequency, eliminating the phase and frequency manipulation and transforming its continuous spectrum into three discrete components at frequencies 2ω ₁ , 2ω ₂ and 2ω ₃ , filter the indicated discrete components and monitor them, divide the phase of the discrete components into two, and distinguish harmonic voltages at symbol frequencies ω ₁ , ω ₂ and ω ₃ , which are selected as follows:

ω ₁ = ω ₃ -1 / 4τ _e - signal frequency corresponding to the symbol "+1";

ω ₂ = ω ₃ + 1 / 4τ _e - signal frequency corresponding to the symbol "-1";

ω ₃ = ω _ol = Ω = (ω ₁ + ω ₂ ) / 2 - the average imaginary "signal frequency";

where τ _e - the duration of the elementary premises;

ω _ol - intermediate frequency,

carry out phase demodulation of the received signal with combined phase and frequency manipulations at an intermediate frequency using harmonic voltages at the first ω ₁ and second ω ₂ symbol frequencies, respectively, select low-frequency voltages at frequencies ω ₃ -ω ₁ and ω ₂ -ω _3, respectively, summarize them , phase demodulation is carried out of the total low-frequency voltage with a voltage at the third harmonic frequency of the symbol ω _3, isolated low-frequency voltage, proportional to the first th modulating code M ₁ (t), used for phase-shift keying, and record it, is performed frequency demodulation of the received signal with combination of the phase and frequency shift keying at an intermediate frequency using harmonic voltages on the first ω ₁ and a second ω ₂ symbol frequencies emit low-frequency voltage proportional to the second modulating code M ₂ (t), used for frequency shift keying, and it is recorded, comparing the phase of the third harmonic voltage symbol frequency ω ₃ GCO nym voltage for Ω the frequency, if said voltages are different from each other in phase, forming a control voltage whose amplitude and polarity of which depends on the extent and direction of deviation of the third symbol frequency ω ₃ of the frequency Ω reference voltage affect them to a frequency ω _g oscillator so so that the symmetry of the frequency Ω of the reference voltage with respect to the symbol frequencies ω ₁ and ω _{2 is} maintained.

The problem is solved in that a phase direction finder, comprising, in accordance with the closest analogue, a first receiving antenna, a first receiver, a mixer, the second input of which is connected to the local oscillator output, an intermediate frequency amplifier, a first multiplier, a first bandpass filter, a delay line, the second phase detector, the second input of which is connected to the output of the first bandpass filter, the first phase shifter 90 °, the first phase detector, the second input of which is connected to the second output of the reference gene a rotor, and an indicator, a reference generator, a pulse generator, an electronic switch, connected in series with n outputs of n receiving antennas arranged in a circle of radius d with the possibility of electronic rotation around the first receiving antenna located in the center of the circle, and a second receiver, output which is connected to the second input of the first multiplier, the second multiplier is connected in series to the output of the first bandpass filter, the second input of which is connected to the output of the local oscillator, the second an axial filter and a third phase detector, the second input of which is connected to the third output of the reference oscillator, and the output is connected to the second input of the indicator, the second receiving antenna, the third receiver, the third multiplier, the second input of which is connected to the output of the intermediate frequency amplifier, the third bandpass filter the fourth phase detector, the second input of which is connected to the output of the local oscillator, the first quadrator, the second input of which is connected to the output of the fourth phase detector, the fourth multiplier, the second input of which is connected to the output of the fourth phase detector, and the subtractor, the output of which is connected to the third input of the indicator, the fourth input of which is connected to the output of the fourth phase detector, the second phase shifter 90 ° connected in series to the output of the fourth phase detector, the second quadrator, the second input of which connected to the output of the second phase shifter by 90 °, and a scaling multiplier, the second input of which is connected to the output of the fourth phase detector, and the output is connected to the second input of the subtraction ator, the second receiving antenna mounted in the azimuth plane at a distance d ₂ from the first reception antenna differs from the closest analog by the fact that it is provided with a doubler phase, three blocks of the phase-locked loop (PLL), three dividers phase two, three narrow band filters frequency demodulator, fifth, sixth, seventh and eighth phase detectors, an adder and a registration unit, and a phase doubler, a first PLL, a first divider are connected in series to the output of the intermediate frequency amplifier There are two steps, the first narrow-band filter, a frequency demodulator, the second input of which is connected to the output of the second narrow-band filter, and the third input - with the output of the intermediate-frequency amplifier, and the recording unit, the second PLL, the second phase divider are connected in series to the phase doubler output , a second narrow-band filter, a sixth phase detector, the second input of which is connected to the output of the intermediate frequency amplifier, an adder and a seventh phase detector, the output of which is connected to the second input of the recording unit, to the output a third PLL, a third phase divider into two and a third narrow-band filter, the output of which is connected to the second input of the seventh phase detector, the fifth phase detector is connected to the output of the first narrow-band filter, the second input of which is connected to the output of the intermediate frequency amplifier, and the output connected to the second input of the adder, the eighth phase detector is connected to the output of the third narrow-band filter, the second input of which is connected to the third output of the reference generator, and the output is connected oscillator output.

The structural diagram of the phase direction finder that implements the proposed phase direction finding method, is presented in figure 1. The relative position of the receiving antennas 1,21,2.i (i = 1, 2, ..., n) and the radio emission source of the IRI is shown in FIG. 2. The relative position of the symbolic frequencies of complex signals with combined phase and frequency manipulation is shown in Fig.3. Timing diagrams illustrating the demodulation of complex QPSK-FSK signals are shown in FIG. 4.

The phase direction finder comprises in series a first receiving antenna 1, a first receiver 3, a mixer 12, a second input of which is connected to the output of the local oscillator 11, an intermediate frequency amplifier 13, a first multiplier 14, a first band-pass filter 15, a delay line 16, a second phase detector 17, and a second the input of which is connected to the output of the first band-pass filter 15, the first phase shifter 8 by 90 °, the first phase detector 9, the second input of which is connected to the second output of the reference generator 5, and the indicator 10. To the first output of the reference generator 5 a pulse generator 6, an electronic switch 7, the n inputs of which are connected to the outputs of n receiving antennas 2.i (i = 1, 2, ..., n) arranged in a circle of radius d with the possibility of their electronic rotation with speed Ω around the first receiving antenna 1, located in the center of the circle, and a second receiver 4, the output of which is connected to the input of the first multiplier 14. To the output of the first bandpass filter 15, a second multiplier 18 is connected in series, the second input of which is connected to the output of the local oscillator 11, the second bandpass filter 19 and a third phase detector 20, the second input of which is connected to the third output of the reference generator 5, and the output is connected to the second input of the indicator 10. The third receiver 22, the third multiplier 23, the second input of which is connected to the output, are connected in series to the output of the second receiving antenna 21 an intermediate frequency amplifier 13, a third bandpass filter 24, a fourth phase detector 25, the second input of which is connected to the output of the local oscillator 11, the first quadrator 26, the second input of which is connected to the output of the fourth phase detector 25, four the second multiplier 29, the second input of which is connected to the output of the fourth phase detector 25, and the subtractor 31, the output of which is connected to the third input of the indicator 10, the fourth input of which is connected to the output of the fourth phase detector 25. The second phase shifter 27 is connected in series to the output of the fourth phase detector 25 90 °, the second quadrator 28, the second input of which is connected to the output of the second phase shifter 27 by 90 °, and a scaling multiplier 30, the second input of which is connected to the output of the fourth phase detector 25, and the output The od is connected to the second input of the subtractor 31. A phase doubler 32, a first PLL block 33, a first phase divider 36 into two, a first narrow-band filter 39, a frequency demodulator 42, the second input of which is connected to the output of the second narrow-band filter, are connected in series to the output of the intermediate frequency amplifier 13. 40, and the third input with the output of the intermediate frequency amplifier 13, and the registration unit 47. The second PLL block 34, the second phase divider 37 into two, the second narrow-band filter 40, the sixth phase detector 44, the second input of which is connected to the output of the intermediate frequency amplifier 13, the adder 45 and the seventh detector 46, the output of which is connected to the output of the phase doubler 32, are sequentially connected with the second input of the registration unit 47. A third PLL block 35, a third phase divider 38 into two and a third narrow-band filter 41, the output of which is connected to the second input of the seventh phase detector 46, are connected in series to the output of the phase doubler 31. A fifth phase detector 43 is connected to the output of the first narrow-band filter 39, the second input of which connected to the output of the intermediate frequency amplifier 13, and the output is connected to the second input of the adder 45. The eighth phase detector 48 is connected to the output of the third narrow-band filter 41, the second input of which is connected to the third output of the reference eratora 5, and an output coupled to the input of oscillator 11.

The proposed method is implemented as follows.

Received complex signals with combined phase and frequency manipulation (QPSK-FSK):

U ₁ (t) = υ ₁ cos [{ω _m (t) ± Δω} t + φ _к (t) + φ ₁ ],

, $$U_2\left(t\right)={\upsilon }_2\cos \left[\left\{{\omega }_m\left(t\right)\pm \Delta \omega \right\}t+{\varphi }_{\mathrm{to}}\left(t\right)+2\pi \frac{d}{\lambda }\cos \left(\Omega t-\alpha \right)\right]$$

,

U ₃ (t) = υ ₃ cos [{(ω _m (t) ± Δω)} t + φ _к (t) + φ ₂ ], 0≤t≤T _c ,

where υ ₁ , υ ₂ , υ ₃ , φ ₁ , φ ₂ , T _c - amplitudes, initial phases and signal duration;

ω _m (t) = {ω ₁ , ω ₂ } is the manipulating component of the frequency that displays the law of frequency manipulation in accordance with the modulating code M ₂ (t) (Fig.4.b), and ω _m (t) = const at mτ _e <t <(m + 1) τ _e and can change abruptly at t = mτ _e , i.e. at the borders between elementary premises (m = 1,2, ..., N-1);

τ _e , N is the duration and number of chips that make up a signal of duration T _c (T _c = Nτ _e );

± Δω is the instability of the carrier frequencies of the signals due to various destabilizing factors, including the Doppler effect;

φ _к (t) = {0, π} is the manipulated phase component that displays the phase manipulation law in accordance with the modulating code M ₁ (t) (Fig. 4, a)

φ _k (t) = const for k τ _e <t <(k + 1) τ _e

and can change abruptly at t = k τ _e , i.e. at the borders between elementary premises (k = 1,2, ..., N-1);

d is the radius of the circle on which the receiving antennas are located 2.i (i = 1,2, ..., n) (measuring base);

Ω is the speed of the electronic rotation of the receiving antennas 2.i (i = 1,2, ..., n) around the first receiving antenna 1;

α - bearing (azimuth) to the signal source;

from the outputs of the receiving antennas 1,2.i (i = 1,2, ..., n) and 21 directly and through the electronic switch 7 are fed to the inputs of the receivers 3, 4 and 22, and then to the first inputs of the mixer 12, the multipliers 14 and 23 respectively. The second input of the mixer 12 from the output of the local oscillator 11 receives voltage

U _g (t) = υ _g cos (ω _g t + φ _g ).

At the output of the mixer 12, voltages of combination frequencies are generated. The amplifier 13 is allocated an intermediate frequency voltage (figure 4, c)

U _pr (t) = υ _pr cos [{ω _pp (t) ± Δω} t + φ _k (t) + φ _pr ], 0≤t≤T _c ,

;Where $${\upsilon }_{PR}=\frac{\mathrm{one}}{2}{\upsilon }_{\mathrm{one}}{\upsilon }_g$$

;

ω _pp (t) = ω _m (t) ω _g is the intermediate frequency;

φ _ol = φ ₁ -φ _g .

In the spectrum of this signal with a continuous phase and an index of frequency manipulation

m _f = (ω ₂ -ω ₁ ) τ _e = 0.5

the symbol frequencies ω ₁ and ω _{2 are} suppressed. The indicated symbolic frequencies are determined as follows (figure 3):

- частота сигнала, соответствующая символу «+1», $${\omega }_{\mathrm{one}}={\omega }_3-\frac{\mathrm{one}}{\mathrm{four}{\tau }_{\mathrm{uh}}}$$

- the frequency of the signal corresponding to the symbol "+1",

- частоты сигнала, соответствующая символу «-1», $${\omega }_2={\omega }_3-\frac{\mathrm{one}}{\mathrm{four}{\tau }_{\mathrm{uh}}}$$

- signal frequency corresponding to the symbol "-1",

- средняя «мнимая»частота сигнала. $${\omega }_{\mathrm{one}}={\omega }_{PR}=\Omega \frac{{\omega }_{\mathrm{one}}+{\omega }_2}{2}$$

- average “imaginary” signal frequency.

Since the spectrum of the received composite FM -CHM _{n n} - character signal frequency ω ₁ and ω ₂ are depressed, the receiver 3 performs tracking of the middle ( "imaginary") frequency ω ₃ = ω _ave = Ω.

The voltage U _CR (t) from the output of the intermediate frequency amplifier 13 is supplied to the second inputs of the multipliers 14 and 23. At the output of the multiplier 14 a phase-modulated (FM) oscillation is generated at the frequency ω _{g of the} local oscillator 11

, 0≤t≤T_c $$U_{\mathrm{four}}\left(t\right)={\upsilon }_{\mathrm{four}}\cos \left[{\omega }_{g}t+{\varphi }_g+2\pi \frac{d}{\lambda }\cos \left(\Omega t-\alpha \right)\right]$$

, 0≤t≤T _c

,Where $${\upsilon }_{\mathrm{four}}=\frac{\mathrm{one}}{2}{\upsilon }_{\mathrm{one}}{\upsilon }_{PR}$$

,

which is allocated by the band-pass filter 15 and supplied to the first inputs of the phase detector 17, delay lines 16 and multipliers 18, the second input of the latter is supplied with the voltage U _g (t) of the local oscillator 11. A harmonic voltage is generated at the output of the multiplier 18

, 0≤t≤T_c, $$U_5\left(t\right)={\upsilon }_5\cos \left[2\pi \frac{d}{\lambda }\cos \left(\Omega t-\alpha \right)\right]$$

, 0≤t≤T _c ,

Where $${\upsilon }_5=\frac{\mathrm{one}}{2}{\upsilon }_{\mathrm{four}}{\upsilon }_g$$

which is allocated by the band-pass filter 19 and supplied to the first input of the phase detector 20. The reference voltage is applied to the second input of the phase detector 20 from the third output of the reference generator 5

U ₀ (t) = υ ₀ cosΩt.

The output of the phase detector 20 produces a low-frequency voltage

U _н1 (α) = υ _н1 сosα,

Where $${\upsilon }_{n\mathrm{one}}\left(\alpha \right)=\frac{\mathrm{one}}{2}{\upsilon }_5{\upsilon }_0$$ ,

which is fixed by indicator 10. Thus, the direction-finding scale of the signal radiation source is formed in the azimuthal plane, which is an accurate but ambiguous scale.

At the same time, the phase-modulated oscillation U ₄ (t) is subjected to autocorrelation processing using an autocorrelator consisting of a delay line 16 and a phase detector 17.

, называемая индексом фазовой модуляции, характеризует максимальное значение отклонения фазы от нулевого значения, происходящего при электронном вращении приемных антенн 2.i (i=1, 2, …, n) вокруг приемной антенны 1. Приемные антенны 2.i (i=1, 2, …, n) поочередно с частотой Ω коммутируются с помощью электронного коммутатора 7, управляемого n-фазным генератором 6 импульсов. Управляющие импульсы формируются генератором 6 импульсов из гармонического напряжения, вырабатываемого опорным генератором 5In phase-modulated voltage U ₄ (t) the value $$m_{\varphi }=2\pi \frac{d}{\lambda }$$

, called the phase modulation index, characterizes the maximum value of the phase deviation from the zero value that occurs during the electronic rotation of the receiving antennas 2.i (i = 1, 2, ..., n) around the receiving antenna 1. Receiving antennas 2.i (i = 1, 2, ..., n) are switched alternately with a frequency Ω using an electronic switch 7 controlled by an n-phase pulse generator 6. The control pulses are generated by the 6 pulse generator from the harmonic voltage generated by the reference generator 5

U ₀ (t) = υ ₀ cosΩt.

However, for d / λ> 1/2, the azimuth count α is ambiguous. The elimination of this ambiguity by reducing the d / λ ratio usually does not justify itself, since the main advantage of the wide-base direction finder is lost. In addition, in the ranges of meter and especially decimeter waves, it is often not possible to take small d / λ values due to design considerations.

In connection with the stated reason, the problem arises of decreasing the phase modulation index without decreasing the relative size of the measuring base d / λ. This is achieved by autocorrelation processing of the phase-modulated voltage U ₄ (t) using the delay line 16 and the phase detector 17. Moreover, the delay time τ _{3 of} the delay line 16 is chosen so as to reduce the phase modulation index to the value:

$$m_{\varphi \mathrm{one}}=2\pi \frac{d_{\mathrm{one}}}{\lambda }$$

where d ₁ > d,

in which the inequality is true:

d ₁ / λ <1/2,

providing unambiguous direction finding of the signal source in the azimuthal plane.

The output of the phase detector 17 produces a voltage

U ₆ (α) = u ₆ cos (Ω-α), 0≤t≤T _c ,

Where $${\mathrm{<figure-callout\; id="6"\; label="\upsilon "\; filenames="00000027.png"\; state="\{\{state\}\}">\upsilon </figure-callout>}}_6=\frac{\mathrm{one}}{2}{\upsilon }_{\mathrm{four}}^2$$

which through the phase shifter 8 through 90 ° enters the first input of the phase detector 9, the second input of which from the second output of the reference generator 5 is supplied with a reference voltage U ₀ (t). The output of the phase detector 9 produces a low-frequency voltage

U _n2 (α) = υ _n2 sinα,

Where $${\upsilon }_{n2}=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="6"\; label="\upsilon "\; filenames="00000027.png"\; state="\{\{state\}\}">\upsilon </figure-callout>}}_6{\upsilon }_0$$

which is fixed by indicator 10. Thus, a rough, but unequivocal scale of direction finding of the signal source in the azimuthal plane is formed.

The voltage U _CR (t) from the output of the amplifier 13 of the intermediate frequency is simultaneously supplied to the second output of the multiplier 23, the output of which produces a voltage at a frequency ω _{g of the} local oscillator 11

U ₇ (t) = υ ₇ cos (ω _g t + φ _g + Δφ), 0≤t≤T _c ,

Where $${\upsilon }_7=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="3"\; label="\upsilon "\; filenames="00000026.png"\; state="\{\{state\}\}">\upsilon </figure-callout>}}_3{\upsilon }_{PR}$$

Δφ = φ ₂ -φ ₁ = 2πd ₂ / λcosβ,

where d ₂ is the distance between the receiving antennas 1 and 21 (measuring base) (figure 2);

λ is the wavelength;

β is the elevation angle of the IRI radiation source;

h is the flight altitude of the aircraft (aircraft, helicopter, airship, probe, etc.), on board which is located the source of radio emission IRI (Fig.2.2),

which is allocated by a band-pass filter 24 and fed to the first (information) input of the phase detector 25, the second (reference) input of which is supplied with the voltage U _g (t) of the local oscillator 11. A low-frequency voltage is generated at the output of the phase detector 25

U _H3 (β) = υ _H3 cosΔφ,

Where $${\mathrm{<figure-callout\; id="3"\; label="\upsilon "\; filenames="00000026.png"\; state="\{\{state\}\}">\upsilon </figure-callout>}}_3=\frac{\mathrm{one}}{2}{\upsilon }_7{\upsilon }_g$$

Δφ = 2πd2 / λcosβ,

which is fixed by indicator 10.

This forms a rough, but unequivocal direction finding scale for the signal source in the elevation (vertical) plane.

The voltage AND _Н3 (β) from the output of the phase detector 25 simultaneously enters the inputs of the first quadrator 26, the fourth multiplier 29 and the second phase shifter 27 by 90 °. At the output of the quadrator 26, which is a multiplier, at the two inputs of which the same voltage U _Н3 (β) is supplied, the following voltage is formed.

U ₈ (β) = υ8cos ² Δφ,

Where $${\upsilon }_8=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="3"\; label="\upsilon \; N"\; filenames="00000026.png"\; state="\{\{state\}\}">\upsilon \; </figure-callout>}}_N^3$$

which is supplied to the first input of the fourth multiplier 29, to the second input of which the voltage U _NC (β) is supplied from the output of the fourth phase detector 25. The output of the fourth multiplier 29 voltage is generated

U ₉ (β) = υ ₃ cos ³ Δφ,

which is fed to the first input of the subtractor 31.

The output of the second phase shifter 27 to 90 ° produces a voltage:

U ₁₀ (β) = - υ _H3 sinΔφ,

which is supplied to the two inputs of the second quadrator 28. The voltage is formed at the output of the latter: U ₁₁ (β) = υ ₁₁ sin ² Δφ,

Where $${\upsilon }_{\mathrm{eleven}}=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="3"\; label="\upsilon \; N"\; filenames="00000026.png"\; state="\{\{state\}\}">\upsilon \; </figure-callout>}}_N^3$$

This voltage is supplied to the first input of the scaling multiplier 30, the second input of which is supplied with voltage U _Н3 (β) from the output of the phase detector 25. The scaling factor K _{m of the} scaling multiplier 30 is chosen equal to 3 (K _m = 3). The output of the scaling multiplier 30 is formed voltage:

U ₁₂ (β) = 3υ ₁₂ cosΔφsin ² Δφ,

Where $${\upsilon }_{12}=\frac{\mathrm{one}}{2}{\mathrm{<figure-callout\; id="3"\; label="\upsilon \; N"\; filenames="00000026.png"\; state="\{\{state\}\}">\upsilon \; </figure-callout>}}_N{\upsilon }_{\mathrm{eleven}}$$

which goes to the second input of the subtractor 31. At the output of the last voltage is formed:

U ₁₃ (β) = υ ₁₃ cos3Δφ,

where U ₁₃ = υ ₉ -3υ ₁₂ ,

Δφ = 2πd ₂ / λсosβ,

3Δφ = 2π3d ₂ / λсosβ,

which is fixed by indicator 10. This voltage is proportional to the triple value of the phase difference between the signals received by the two receiving antennas 1 and 21, and corresponds to the measuring base 3d ₂ , which is constructed indirectly.

Thus, an accurate, but ambiguous, direction-finding scale of the signal radiation source in the elevation (vertical) plane is formed. Moreover, between the measuring bases establish the following inequality:

$$\frac{d_2}{\lambda }<\frac{\mathrm{one}}{2}\le \frac{3d_2}{\lambda }$$

The voltage U _p (t) (Fig. 4, c) from the output of the intermediate frequency amplifier 13 is supplied to the first input of the frequency demodulator 42 and to the input of the phase doubler 32.

Upon doubling the phase, the received complex QPSK-FSK signal of intermediate frequency acquires the frequency modulation index mf = l and its continuous spectrum is transformed into three discrete components at frequencies 2ω ₁ , 2ω ₂ , 2ω ₃ . Using PLLs 33, 34, and 35, these discrete components are filtered and tracked, and phase dividers 36, 37, and 38 ensure that the frequencies of the synchronization signals correspond to the received FMN-FMN signal. At the output of the dividers 36, 37 and 38 of the phase into two, harmonic oscillations are formed (Fig. 4, d, e, e):

U ₁₄ (t) = υ ₁₄ cos (ω ₁ t + φ ₁ ),

U ₁₅ (t) = υ ₁₅ cos (ω ₂ t + φ ₂ ),

U ₁₆ (t) = U ₁₆ cos (ω ₃ t + φ ₃ ),

which are highlighted by narrow-band filters 39, 40 and 41, respectively. Harmonic oscillations U14 (t) and U15 (t) are fed to the first inputs of the phase detectors 43 and 44, the second input of which is supplied with an intermediate frequency voltage ω _p (t).

At the output of the phase detectors 43 and 44, voltages are generated, respectively:

U ₁₇ (t) = υ ₁₇ cos [(ω ₂ -ω ₂ ) t + φ _к (t)],

U ₁₈ (t) = υ ₁₈ cos [(ω ₂ -ω ₃ ) t-φ _к (t)],

Where $${\upsilon }_{17}=\frac{\mathrm{one}}{2}{\upsilon }_{PR}{\upsilon }_{\mathrm{fourteen}}$$

$${\upsilon }_{\mathrm{eighteen}}=\frac{\mathrm{one}}{2}{\upsilon }_{PR}{\upsilon }_{\mathrm{fifteen}}$$

which is summed in the adder 45

$$U_{\Sigma }\left(t\right)=U_{17}\left(t\right)+U_{\mathrm{eighteen}}\left(t\right)={\upsilon }_{\Sigma }\cos \left[{\omega }_{3}t-\frac{\left({\omega }_2+{\omega }_{\mathrm{one}}\right)t}{2}+\varphi \left(t\right)\right]\cos \frac{\left({\omega }_2-{\omega }_{\mathrm{one}}\right)t}{2}$$

The total voltage U _Σ (t) is supplied to the first (information) input of the phase detector 46, to the second (reference) input of which harmonic oscillation U ₁₆ (t) is supplied from the output of the third narrow-band filter 41.

As a result of synchronous detection, a low-frequency voltage is generated at the output of the phase detector 46

U _H4 (t) = υ _H4 cos [ω ₃ t- (ω ₁ + ω ₂ ) / 2t + φ _к (t)] = υ _H4 cosφ _к (t),

where υ _H4 = 1 / 2υ _Σ υ ₁₆ ; ω ₃ - (ω ₁ + ω ₂ ) / 2 = 0,

since the symbolic frequencies ω ₁ and ω _{2 are} symmetrical with respect to the frequency ω ₃ (Fig. 3).

The low-frequency voltage U _low (t) (figure 4, g), proportional to the modulating code M ₁ (t) (figure 4, a), is fixed by the registration unit 47.

Harmonic oscillations U ₁₄ (t) and U ₁₅ (t) from the output of narrow-band filters 39 and 40 are supplied to the reference inputs of the frequency demodulator 42, the information input of which supplies voltage U _p (t) from the output of the intermediate frequency amplifier 13. As a result of synchronous detection at the output of the frequency demodulator 42, a low-frequency voltage is generated (Fig. 4, h)

U _H5 (t) = υ _H5 cosφ _к2 (t),

where U _H5 = 1 / 2υ _pp υ ₁₇ ,

proportional to the modulating code M ₂ (t). This voltage is detected by the registration unit 47.

To ensure the symmetry of the symbol frequencies ω ₁ and ω ₂ with respect to the frequency ω _h = Ω of the reference oscillator 5, a phase-locked loop system consisting of a reference oscillator 5, a narrow-band filter 41 and a phase detector 48 is used.

The voltage U ₆ (t) of the reference oscillator 5 and harmonic oscillation

U ₁₆ (t) = υ ₁₆ cos (ω ₃ + φ ₃ )

the output of the narrow-band filter 41 is fed to two inputs of the phase detector 48 and compared in phase. If these voltages differ in phase, then a control voltage is generated at the output of the phase detector 48. Moreover, the amplitude and polarity of this voltage depend on the degree of direction of the deviation of the intermediate frequency ω _CR = ω ₃ from the frequency Ω of the reference oscillator 5. This voltage is applied to the control input of the local oscillator 11 and acts on its frequency ω _g so that the symbol frequencies ω ₁ and ω ₂ relative to the frequency Ω of the reference oscillator 5.

The proposed method and device provide accurate and unambiguous determination of the elevation angle of the signal source located on board the aircraft (aircraft, helicopter, airship, probe, etc.). In this case, two measuring bases are used: a small d _{2 is} rough, but unambiguous, and a large 3d _{3 is} accurate, but ambiguous, between which the following inequality is established:

d ₂ / λ <1 / 2≤3d ₂ / λ.

Moreover, an accurate but ambiguous measuring base of 3d _{2 is} formed by an indirect method.

Thus, the proposed method and device in comparison with prototypes and other technical solutions of a similar purpose provide an accurate and unambiguous determination of the azimuth and elevation angle of the radiation source of a complex signal with combined phase and frequency manipulation, located on board the aircraft (aircraft, helicopter, airship, probe etc.), and its synchronous detection.

In this case, the reference voltages necessary for synchronous detection of the received complex signal with combined phase and frequency manipulations are extracted directly from the received signal itself. Thus, the functionality of the known method and device is expanded.

Claims (2)

1. The phase direction finding method, based on the fact that the signals are received, amplified and limited in amplitude, the signals transmitted by two channels are compared in phase, while the signal of one of the channels is pre-phase shifted by 90 °, set in the azimuth plane n receiving antennas around a circle of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receive antennas are arranged around the circle, alternately with frequency Ω, the signal received by the ant the data located in the center of the circle is frequency-converted using the local oscillator frequency, the intermediate frequency voltages are extracted, it is multiplied with signals alternately received by n receiving antennas located around the circle, the phase-modulated voltage is isolated, it is multiplied with the local oscillator voltage, the first low-frequency voltage is isolated with frequency Ω and compare it in phase with the reference voltage, forming an accurate, but ambiguous direction finding scale of the signal radiation source, in the azimuthal plane, at the same time, the phase-modulated voltage is subjected to autocorrelation processing, a second low-frequency voltage with a frequency Ω is isolated, it is compared in phase with the reference voltage, forming a rough but unambiguous direction finding scale for the signal source in the azimuthal plane, the second receiving antenna is installed in the elevation plane at a distance of d ₂ from the first receive antenna, receive a signal on it, amplify and limit it in amplitude, multiply it with an intermediate frequency voltage, emit a harmonic the voltage at the local oscillator frequency, multiply it with the local oscillator voltage, isolate a voltage proportional to the phase difference between the signals received by the first and second receiving antennas, forming a rough, but unequivocal direction finding scale for the signal source in the elevation plane, this voltage is squared, multiplied with the original voltage, forming a product, at the same time the initial voltage, proportional to the phase difference between the signals received by the first and second antennas, is shifted in phase by 9 0 °, square it, multiply it with the generated product using a scaling factor of three, and subtract the resulting product from the generated product, forming an accurate but ambiguous direction finding scale for the signal source in the elevation plane, characterized in that they double the phase of the received signal with combined phase and frequency manipulation at an intermediate frequency, eliminating phase and frequency manipulation and transforming its continuous spectrum into three discrete stavlyayuschie frequencies 2ω _1, 2ω 2ω ₂ and _3, said filtering is carried out discrete components and tracking, phase divided into two discrete components are isolated harmonic voltage at symbol frequencies ω _1, ω ₂ and ω _3, which is selected as follows:
ω ₁ = ω ₃ -1 / 4τ _e - signal frequency corresponding to the symbol "+1";
ω ₂ = ω ₃ + 1 / 4τ ₃ - signal frequency corresponding to the symbol "-1";
ω ₃ = ω _CR = Ω = (ω ₁ + ω ₂ ) / 2 - the average "imaginary" signal frequency,
where τ _e - the duration of the elementary premises,
ω _ol - intermediate frequency,
carry out phase demodulation of the received signal with combined phase and frequency manipulations at an intermediate frequency using harmonic voltages at the first ω ₁ and second ω ₂ symbol frequencies, respectively, select low-frequency voltages at frequencies ω ₃ -ω ₁ and ω ₂ -ω _3, respectively, summarize them , phase demodulation is carried out of the total low-frequency voltage with a voltage at the third harmonic frequency of the symbol ω _3, isolated low-frequency voltage, proportional to the first th modulating code M ₁ (t), used for phase-shift keying, and record it, is performed frequency demodulation of the received signal with combination of the phase and frequency shift keying at an intermediate frequency using harmonic voltages on the first ω ₁ and a second ω ₂ symbol frequencies emit low-frequency voltage proportional to the second modulating code M ₂ (t), used for frequency shift keying, and it is recorded, comparing the phase of the third harmonic voltage symbol frequency ω ₃ GCO nym voltage for Ω the frequency, if said voltages are different from each other in phase, forming a control voltage whose amplitude and polarity of which depends on the extent and direction of deviation of the third symbol frequency ω ₃ of the frequency Ω reference voltage affect them to a frequency ω _g oscillator so so that the symmetry of the frequency Ω of the reference voltage with respect to the symbol frequencies ω ₁ and ω _{2 is} maintained. 2. Phase direction finder containing a first receiving antenna in series, a first receiver, a mixer, the second input of which is connected to the local oscillator output, an intermediate frequency amplifier, a first multiplier, a first bandpass filter, a delay line, a second phase detector, the second input of which is connected to the first bandpass output filter, the first phase shifter 90 °, the first phase detector, the second input of which is connected to the second output of the reference oscillator, and an indicator, the reference oscillator connected in series, the generator a pulse generator, an electronic switch, n inputs of which are connected to the outputs of n receiving antennas arranged around a circle of radius d with the possibility of electronic rotation around the first receiving antenna located in the center of the circle, and a second receiver, the output of which is connected to the second input of the first multiplier, connected in series to the output of the first bandpass filter, a second multiplier, the second input of which is connected to the output of the local oscillator, a second bandpass filter and a third phase detector, the second input of which is connected to the tre the third output of the reference generator, and the output is connected to the second input of the indicator, the second receiving antenna, the third receiver, the third multiplier, the second input of which is connected to the output of the intermediate frequency amplifier, the third pass filter, the fourth phase detector, the second input of which is connected to the local oscillator output the first quadrator, the second input of which is connected to the output of the fourth phase detector, the fourth multiplier, the second input of which is connected to the output of the fourth phase detector, and a transformer whose output is connected to the third input of the indicator, the fourth input of which is connected to the output of the fourth phase detector, serially connected to the output of the fourth phase detector, a second phase shifter 90 °, a second quadrator, the second input of which is connected to the output of the second phase shifter 90 °, and scaling a multiplier, the second input of which is connected to the output of the fourth phase detector, and the output is connected to the second input of the subtractor, while the second receiving antenna is installed in the azimuthal plane a distance d ₂ from the first reception antenna, characterized in that it is provided with a doubler phase, three blocks of the phase-locked loop (PLL), three dividers phase two, three narrowband filters, a frequency demodulator, fifth, sixth, seventh and eighth phase detector, an adder and a registration unit, and a phase doubler, a first PLL, a first phase divider into two, a first narrow-band filter, a frequency demodulator, the second input of which is connected to the second narrow-band filter, and the third input with the intermediate-frequency amplifier output, and the recording unit, the second PLL, the second phase divider into two, the second narrow-band filter, the sixth phase detector, the second input of which is connected to the amplifier output, are connected in series to the output of the phase doubler intermediate frequency, the adder and the seventh phase detector, the output of which is connected to the second input of the registration unit, the third PLL, the third phase divider into two and the third are connected in series to the output of the phase doubler the fifth-band filter, the output of which is connected to the second input of the seventh phase detector, the fifth phase detector is connected to the output of the first narrow-band filter, the second input of which is connected to the output of the intermediate frequency amplifier, and the output is connected to the second input of the adder, the eighth phase detector is connected to the output of the third narrow-band filter , the second input of which is connected to the third output of the reference generator, and the output is connected to the input of the local oscillator.

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