RU2360265C1 - Method of radar detection of mobile targets with phase selection on range and device to this end - Google Patents

Abstract

FIELD: radar-location.

SUBSTANCE: invention pertains to systems for extracting information from probing and reflected signals from a target and can be used in making security systems, in location and navigation. The technical outcome is achieved due to that, a special probing signal is generated and the level of correlation of Doppler frequency signals is analysed. Open-phase fault is eliminated in the device due to smooth frequency tuning, which cuts the operation frequency band of the device and maintains coherence of the probing signal and the reflected signal, and operation of the frequency converter on a matched load in the intervals between pulses, which cuts on time and intensity of transient processes and allows for more accurate pick up of Doppler frequency signals.

EFFECT: increased accuracy of selection on range, minimisation of the number of used frequencies of the probing signal, while maintaining unambiguity within the limits of the intended range of operation.

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Description

The invention relates to the field of radar systems and can be used in security systems, locations and navigation to determine the presence of moving targets.

Methods and devices for a similar purpose are discussed in more detail in the books of Bakulev P.A., Stepina V.M. Methods and devices for moving targets selection. - M.: Radio communication, 1986 and Information technology in radio systems: A manual - 2nd ed. reslave. and add. / Vasin V.A., Vlasov I.B., Egorov Yu.M. et al. - M.: Publishing House. MSTU named after Bauman, 2004.

A known method of radar detection of a maneuvering target in a pulse-Doppler radar described in patent RU 2154837 C1, G01S 13/02, 2000. The method uses complex signals with in-pulse frequency modulation to determine the parameters of a moving target and based on determining the maximum modulus of the correlation sum of samples signal reflected from the target and the reference signal in the parameter space: signal frequency and its derivative. In a device that implements this method, analog-to-digital conversion (ADC) of the received signal is carried out, it is conditionally divided into a number of frequency sub-bands, the accumulation of signals, certain frequencies in the shift registers, fast Fourier transform of the received samples with subsequent processing. The disadvantages of the device are the difficulties of determining and setting the frequency grid, the low accuracy of determining the coordinates due to the nonlinearity of the signal with linear frequency modulation.

There is also a method of detecting and selecting moving targets, which consists in alternately emitting radio pulses at two or more carrier frequencies and dividing the Doppler frequency signals of the respective carrier frequencies in the receiver. The values of the carrier frequencies are chosen so that the phase difference of the Doppler frequencies varies from 0 ° to 90 ° when the target is located and moved in the field of view up to the maximum range. In the radar intrusion detection system described in the application US 2002/0060639 A1, G01S 13/62, 2002, implementing this method, the microwave transmitter alternately emits radio pulses with different carrier frequencies. The received signals are detected, amplified, divided into in-phase and quadrature components and are subjected to ADC, followed by digital phase measurement according to the algorithms for constructing Lissajous figures. The ratio of signals proportional to the sizes of the major and minor axes of the resulting ellipse gives information about the range. For speed selection, the processing device has high and low speed channels, respectively, for fast and slow moving targets. In the microwave sensor described in the application US 2004/0222887 A1, G08B 13/18, 2004, using the same method, radio pulses are emitted with different carrier frequencies and the phase difference of the Doppler frequencies of the respective carrier frequencies is determined. The essence of the operation of the sensor receiving device is the formation of rectangular signals from Doppler frequency signals using a comparator and the subsequent determination of the phase difference between them. The disadvantages of the considered devices are the low accuracy and ambiguity of determining the range inherent in phase ranging methods.

сигналов доплеровской частоты и пропорциональна отношению разности несущих частот ω₀ и ω_n к скорости света с (см. Методы и устройства селекции движущихся целей, с.39), т.е.The closest in fact is a motion detector based on the Doppler principle described in US patent 6380882 B1, G01S 13/56, 2002, using a multi-frequency probe signal and a multi-channel processing device. When using a two-frequency probing signal, the Doppler frequency signals allocated in the receiver corresponding to the carrier frequencies (ω ₁ and ω ₂ ) are supplied to the channels K ₁ and K ₂ and then to the calculation device. The algorithm for calculating the phase difference is as follows: first, the absolute values of the Doppler frequency signals are integrated, getting I ₁ and I _2, respectively, the Doppler frequency signals are also multiplied and the signal product integral is calculated, getting I ₃ , then to calculate the phase difference, I ₃ is divided by the product I ₁ and I ₂ . The calculated values of the phase difference at certain time intervals determine the value of the radial velocity. The direction of movement is determined by the sign of the phase difference. Indirectly, the size of the target is judged by the power value of the received reflected signal. In the last (fourth) version of the motion detector, a multi-frequency sounding signal is emitted and multi-channel processing of Doppler frequency signals is carried out. In channels corresponding to their value of the carrier frequency, amplification, filtering, ADC with subsequent processing based on the Fourier transform, fast Fourier transform or wavelet transform is carried out. In this method, it is possible to eliminate the ambiguity of ranging. However, for the uniqueness of range measurements, it is necessary that the error of a coarser phase measurement be less than the interval of uniqueness of a more accurate phase measurement. Since the error and the interval of uniqueness of phase measurements of the range R to the moving target is determined by the steepness of the phase characteristic

signals of Doppler frequency and is proportional to the ratio of the difference of the carrier frequencies ω ₀ and ω _n to the speed of light s (see Methods and devices for selecting moving targets, p. 39), i.e.

then randomly selected carrier frequencies of radio pulses do not guarantee the unambiguity of phase measurements within the intended range, and an increase in the number of carrier frequencies of the probe signal complicates the device.

The technical result of the proposed method for radar detection of moving targets with phase selection in range is to increase the accuracy of range selection, minimizing the number of frequencies of the probing signal while maintaining ambiguity within the intended range.

The technical result of the proposed device is the improvement of technical characteristics, because:

- the operation of the transmitter and receiver nodes with a continuous signal with a smoothly tunable frequency eliminates phase discontinuity, which reduces the working frequency band of the device and maintains the coherence of the probe and reflected signals;

- the operation of the frequency converter for a coordinated load during pauses between radiation pulses reduces the time and intensity of transient processes, which makes it possible to more accurately isolate Doppler frequency signals and use probing radio pulses of nanosecond duration.

The essence of the proposed method lies in the fact that they form probing radio pulses with different carrier frequencies, then, after emitting and receiving radio pulses, the frequency is converted by multiplying the probing and received radio pulses with simultaneous multiplexing to extract the Doppler frequency signals of the corresponding carrier frequencies, then the frequency and phase difference of the signals are determined Doppler frequency, receiving information about the radial speed and range to the target, according to the proposed image probing periodic bursts of radio pulses are formed, the carrier frequency of which relative to the reference value of the carrier frequency of the first burst radio pulse is changed to a value that doubles with each subsequent burst pulse, then, to detect a moving target at a given range, after converting and multiplexing the signals, the Doppler signal convolution operations are performed the frequency of the corresponding reference carrier frequency with other Doppler frequency signals shifted in phase in accordance and with a given range, then multiply the results of the convolution of the signals, getting the correlation value of the Doppler frequency signals, a comparison of which with the threshold level allows us to conclude that there is a target at a given range.

In the device for implementing the method of radar detection of moving targets with phase selection in range, containing a pulse modulator in the transmitter and serially connected first and second frequency dividers, the outputs of which are connected to the matching circuit, the output of the matching circuit is connected to the second input of the pulse modulator, in series connected frequency converter, multiplexer, Doppler frequency filters, which are parallel bandpass filters, in the antenna node - the circulator and the antenna are connected properly, the input of the circulator connected to the output of the pulse modulator, the output of the circulator connected to the first input of the frequency converter, the second output of the frequency synthesizer connected to the pulse modulator and the second input of the frequency converter, the output of the matching circuit connected to the control input of the multiplexer, according to the proposed the invention is additionally introduced: in the transmitter - a third frequency divider and series-connected control code generator and frequency synthesizer, on the first output of which is formed by a continuous signal with a reference frequency, and the second is a continuous signal with a smoothly tunable frequency; to the receiver, a coordinated load and a processing unit, the first output of the frequency synthesizer connected to the input of the third frequency divider, the output of which is connected to the input of the first frequency divider, the outputs of the frequency dividers are connected to the inputs of the driver of the control code, while the first divider generates a signal specifying the repetition period and the duration of the probe pulses, the second divider generates a signal that sets the repetition period and the duration of the packets of probe pulses, the third divider sets the control signal for smoothly of frequency tuning, one of the outputs of the multiplexer is connected to a matched load, the input impedance of which is equal to the input impedance of the Doppler frequency filters, the outputs of the Doppler frequency filters are connected to a processing unit for calculating the correlation value of Doppler frequency signals, comparing the obtained value with a threshold level and obtaining output about the presence of a target at a given range.

Figure 1 presents the main and side lobes of the overall phase mismatch function.

Figure 2 presents the General function of the mismatch in phase.

Figure 3 presents diagrams explaining the method of processing Doppler signals.

Figure 4 and 5 presents the dependence of changes in the carrier frequency of the probe radio pulses for the implementation of the proposed method.

Figure 6 presents a functional diagram of a device that implements the proposed method.

7 is a timing chart explaining the operation of a device that implements the proposed method.

A device for radar detection of moving targets (Fig.6) consists of a transmitter 1, a receiver 2 and an antenna assembly 3. The transmitter 1 contains: PKU 4 — a control code generator, MF 5 — frequency synthesizer, IM 6 — pulse modulator, FM 7, 8 , 9 - frequency dividers, And 10 - coincidence scheme. Receiver 2 contains: IF 11 - frequency converter, MP 12 - multiplexer, PDF 13 - Doppler frequency filters, UO 14 - processing unit, SN 15 - matched load. Antenna node 3 contains: C 16 - circulator, A 17 - antenna.

To justify the receipt of the claimed technical result by the proposed method, it is necessary to determine the overall phase mismatch function (OFP), which is proportional to the correlation value of the Doppler frequency signals. The correlation values of the Doppler frequency signals are greater, the closer the moving target to a given range.

Further in the description, the term “phase difference" refers to the phase difference due to propagation delay, and the term "phase shift" refers to a hardware phase shift.

Let the carrier frequencies of the radio pulses ω ₀ , ω ₁ , ω ₂ , ..., ω _{N be given} . Information on the range to the moving target is contained in the phase difference Θ ₁ , Θ ₂ , ..., Θ _N Doppler signals S ₁ , S ₂ , ..., S _N corresponding to the carrier frequencies ω ₁ , ω ₂ , ..., ω _N relative to the Doppler signal S ₀ corresponding to the reference carrier frequency ω ₀ . The phase difference is determined in accordance with (see Methods and devices for moving targets selection, p. 39)

where R is the distance to the moving target;

Δω _n is the difference of the respective carrier frequencies.

Representing S _{n as a} sinusoidal oscillation with frequency ω _D and amplitude A ₀ , we determine the phase mismatch function χ _n (ΔΘ) for signals S ₀ and S _n proportional to the convolution value of signals S ₀ and S _n , as

where Θ _n is the phase difference of the signals S ₀ and S _n ;

Θ ' _n is the hardware phase shift S _n relative to S ₀ ;

ΔΘ _n - phase mismatch equal to

where R, R'- the actual range to the moving target and the range to which the device is configured, respectively.

Performing integration (3), we obtain χ _n (ΔΘ) for signals S ₀ and S _n

or subject to (4)

Equations (5), (6) show that the phase mismatch function (in range) for the signals S ₀ and S _n has a cosine dependence with a period inversely proportional to the difference of the corresponding carrier frequencies.

It is optimal to set the shape of the main and side lobes of the FGP in the form of functions

where Ψ ^DOS - the main lobe of the OFR;

- отрицательные побочные лепестки, удаленные от основного на нечетное число раз π, в соответствии с индексом k'=±1, ±3, ±5…;

- negative side lobes, removed from the main one an odd number of times π, in accordance with the index k '= ± 1, ± 3, ± 5 ...;

- положительные побочные лепестки, удаленные от основного на четное число раз π, в соответствии с индексом k''=±2, ±4, ±6…;

- positive side lobes removed from the main one an even number of times π, in accordance with the index k '' = ± 2, ± 4, ± 6 ...;

G - coefficient characterizes the ratio of the OFR repetition period to the width of the petal to the zero level, and the last equality in (8), (9) holds for a whole value of G.

Figure 1 presents the main and side lobes of the OFR for the case of G = 8.

OFs are determined by the sum of the main and side lobes i.e.

where the index k = ± 1, ± 2, ± 3 ....

можно представить какEquation (10) for small values of the argument

can be imagined as

Figure 2 presents the OFR, constructed according to (11), for the case of G = 8.

Further, if we take G = 2 ^N , where N is a positive integer, then sequentially applying the trigonometric formula of the double angle, we can express expression (11) as

Expression (12) shows that, taking into account (6), one can express FGD in range as the product of the mismatch functions χ _n (ΔR), if Δω _{n is} defined as (2 ^n-1 · Δω). Dropping the factor A ₀ for normalizing the RBF, we obtain

where Δω is the minimum change in the carrier frequency.

Figure 3 presents diagrams explaining the method of processing Doppler signals for the case of G = 4. On figa presents Doppler signals S ₀ , S ₁ , and S _{2 a} moving target, which have a phase difference relative to each other depending on the distance to the target in accordance with (2). On figb presents the result of multiplying S ₀ S ₁ and S ₀ S ₂ in accordance with the integrand expression of the convolution formula (3), while the hardware phase shift was not performed. On figv presents the result of integration of the multiplied Doppler signals in accordance with (3), with an integration period equal to three periods of the Doppler frequency. On Figg presents the result of the multiplication of signals obtained as a result of previous operations in accordance with (13). The obtained dependence of the correlation value of the Doppler frequency signals on the range is proportional to (11) for the case G = 4.

Figure 4 and 5 presents the dependence of changes in the carrier frequency of the probe radio pulses for the implementation of the proposed method. At times

t ₁ , t ₂ , t ₃ , t ₄ , ..., t _{(N + 1)} , t ₁ + T _P , t ₂ + T _P , t ₃ + T _P , t ₄ + T _P ...,

where T _P - the period of the probe bursts of radio pulses, formed radio pulses with a carrier frequency (figure 3)

or (figure 4)

where ω ₀ is the reference carrier frequency corresponding to the first radio pulses in packs;

Δω is the minimum change in the carrier frequency;

(2 ^N-1 Δω) is the maximum change in the carrier frequency;

(N + 1) - the number of radio pulses in the packet and, accordingly, the number of used carrier frequencies in the probe signal.

OFR (13) within the range of uniqueness (intended range) has a dependence similar in shape to the function sin (x) / x, having the narrowest width of the main lobe compared with other autocorrelation functions of the compact signals, which explains the claimed technical result of increasing selection accuracy.

The following are the calculation formulas for specific implementations of the method.

The OFR uniqueness interval can be obtained by substituting in (2) the values of the phase difference 2π and the minimum change in the carrier frequency Δω, i.e.

The resolved distance of the FGD by distance at the zero level can be obtained by substituting in (2) the values of the phase difference π and the maximum change in the carrier frequency (2 ^N-1 Δω), i.e.

A coefficient characterizing the ratio of the OFR repetition period to the width of the lobe to the zero level can be obtained by taking the ratio R _ODN and ΔR, i.e.

To describe the operation of the device, we use a mathematical apparatus taken from the book by A. Denisenko Signals. Theoretical radio engineering. Reference manual. - M .: Hot line - Telecom, 2005.

MF 5 generates, at the first benefit, a signal with a reference frequency ω ₀ , which is fed to the input of the PM 7. From the output of the MF 7, a signal U is supplied to the PCF 4, which sets the control signal for smooth frequency tuning, and to the input of the MF 8 (Fig. DF 8 generates a signal U _And , setting the repetition period and the duration of the probe pulses (Fig.7b), which can be represented as

where rect () is the function of the rectangular momentum equal to

n is the index defining the pulse number;

T _And - the period of the pulses;

τ is the pulse duration.

The output of the PM 8 is connected to the PMF 4 and I 10. The output of the PM 8 is also connected to the PM 9, forming a signal U _P , which sets the repetition period and duration of the packets of probe pulses (Fig.7c) and can be represented as

where m is the number of the pulse forming the pack;

T _P - the repetition period of the pulses forming the packs;

(N + 1) · T _AND is the duration of the pulse forming the packet.

TM output 9 is connected with PKU 4 and AND 10. AND 10 generates a signal U _mgmt (fig.7g) being the control signal MI for 6 and MPU 12, which may be represented as

PKU 4 controls the midrange 5 to form on the second output a continuous signal with a smoothly tunable frequency (Fig.7d), fed to the MI 6 and to the second input of the inverter 11, which is the input of the local oscillator signal.

PKU generates a control signal, which is a dependence of the frequency change (Fig.7d) in binary code and is generated by the following algorithm: if U _П = 1 and U _И = 0, then along the edge of the signal U, the value of the control code increases; if U _П = 1 and U _И = 1, then the value of the control code does not change; if U _П = 0, then along the edge of the signal U, the value of the control code decreases.

To describe the law of frequency change, we define the unit jump function

The law of variation of the carrier frequency in the probing radio pulses can be represented as

where the “+” sign is used to describe the option presented in FIG. 4, the “-” sign is for the option presented in FIG. 5.

To describe the law of smoothly tunable frequency, the moments between pulses in a packet and between pulse packets are represented by a signal U ' _Upr , which is inverted with respect to U _Upr .

where U ₁ (m, n, t), U ₂ (m, t) are the control signals at the moments between pulses in the packet and between the pulse packets, respectively, determined

then the law of smoothly tunable frequency can be represented

where ω ₁ (m, n, t), ω ₂ (m, t) is the law of smoothly tunable frequency at moments between pulses in a packet and between pulse packets, respectively, determined

where the “+” sign is used to implement the option presented in FIG. 3, the “-” sign is used to implement the option presented in FIG. 4.

Thus, the law of frequency change (Fig.7d) at the second output of the midrange 5 can be represented as

IM 6 forms a probe radio pulse, which is transmitted through space through C 16 and A 17. Next, the received A 17 and the branched C 16 signal are supplied to the first signal input of the inverter 11. The antenna node may contain separate transmitting and receiving antennas, respectively connected to the MI 6 and the IF 11. The output of the IF 11 is connected to the MP 12. And 10 controls the MP 12 for separation of Doppler frequency signals of the respective carrier frequencies. The Doppler frequency signals are fed to the FDCH 13, which are parallel bandpass filters of the same type, limiting the range of possible frequencies of Doppler frequency signals and amplifying them. During pauses between pulses and bursts of pulses, the MP 12 connects the output of the inverter 11 to the CH 15, the input resistance of which is equal to the input impedance of the PDF 13. From the output of the PDF 13, the Doppler frequency signals are fed to UO 14, which, in accordance with the proposed method, calculates the correlation value of the Doppler signals frequency, a comparison of which with a threshold level allows us to conclude that there is a target at a given range.

Claims (2)

1. The method of radar detection of moving targets with phase selection in range, which consists in generating probing radio pulses with different carrier frequencies, after emitting and receiving radio pulses, the frequency is converted by multiplying the probing and received radio pulses with simultaneous multiplexing to extract Doppler frequency signals of the respective carriers frequency, then determine the frequency and phase difference of the Doppler frequency signals, receiving information about the radial speed and range to the target, characterized in that they form probing periodic packets of radio pulses, the carrier frequency of which relative to the reference value of the carrier frequency of the first radio pulse of the packet is changed to a value that doubles with each subsequent pulse of the packet, then to detect a moving target at a given range after conversion and the multiplexing signals carry out the convolution of the Doppler frequency signal corresponding to the reference carrier frequency with other Doppler signals frequency, phase shifted in accordance with a predetermined range, and then multiply the results of the convolutions signals to obtain a correlation value of the Doppler frequency signals, comparison with a threshold level which allows to infer the presence of target by a predetermined distance. 2. A device for radar detection of moving targets with phase selection in range, comprising a frequency synthesizer, a pulse modulator and series-connected first and second frequency dividers, the outputs of which are connected to the matching circuit, the output of the matching circuit is connected to the second input of the pulse modulator, in the receiver serially connected frequency converter, multiplexer, Doppler frequency filters, which are parallel bandpass filters, in the antenna node are followed the circulator and the antenna are connected directly, the input of the circulator connected to the output of the pulse modulator, the output of the circulator connected to the first input of the frequency converter, the second output of the frequency synthesizer connected to the first input of the pulse modulator and the second input of the frequency converter, the output of the matching circuit connected to the control input of the multiplexer, characterized in that it is additionally introduced: a third frequency divider and a control code generator connected to the frequency synthesizer at the first output to of which a continuous signal with a reference frequency is generated, and on the second a continuous signal with a smoothly tunable frequency, a matched load and a processing unit to the receiver, the first output of the frequency synthesizer connected to the input of the third frequency divider, the output of which is connected to the input of the first frequency divider, outputs frequency dividers are connected to the inputs of the driver of the control code, while the first divider generates a signal that sets the repetition period and the duration of the probe pulses, the second divider generates a signal, sets the repetition period and duration of the bursts of probe pulses, the third divider sets the control signal for smooth frequency tuning, one of the outputs of the multiplexer is connected to a matched load, the input resistance of which is equal to the input resistance of the Doppler frequency filters, the outputs of the Doppler frequency filters are connected to the processing unit for calculating the correlation values of the Doppler frequency signals, comparing the obtained value with a threshold level and obtaining a conclusion about the presence of a target at a given range.

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