RU2491572C1 - Method of providing constant range resolution in pulse radar station with quasirandom phase modulation - Google Patents

Abstract

FIELD: radio engineering, communication.

SUBSTANCE: emitted pulse consists of multiple time sections with serially varying duration; inside each time section, a quasirandom phase-modulated sequence of harmonic signal readings is generated and then converted to voltage which is passed through a low-pass filter which forms the modulation bandwidth ΔF_m; from the output of filter, the signal is transferred to carrier frequency, power-amplified and transmitted to the antenna for radiation; reflected pulses received by the antenna are amplified, filtered at carrier frequency by a band-pass filter with transmission band ΔF_m, time-sampled with frequency F_D≥2ΔF_m and level-quantised by an analogue-to-digital converter; the formed readings are transferred to zero frequency by digital quadrature heterodyning of a complex sinusoid with frequency ΔF_m/2, and then filtered with a digital quadrature low-pass filter, readings at the output of which are formed with repetition frequency F_S≥ΔF_m and then processed with a compression filter, the pulse characteristic of which on duration, on the phase modulation function and on the expected Doppler frequency shift is matched with the reflected pulse, wherein the delay obtained at the output of the filter of maximum signal amplitude relative the beginning of the reception cycle corresponds to distance to the detected target, and frequency of the pulse characteristic at which maximum amplitude is obtained corresponds to Doppler shift of the frequency of the target, uniquely associated with its radial velocity.

EFFECT: measuring distance to targets with constant resolution in the entire range of distance and simultaneous measurement of radial velocities of targets when emitting one modulated pulse and when receiving reflected pulses, duration of which can be shorter than duration of the probing pulse.

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Description

The invention relates to radar technology and can be used to detect and measure distances to various kinds of stationary objects, as well as to measure the radial speed of moving objects. The method can be used in radar and sonar, in the field of geology, archeology, land management and other fields of activity where it is necessary to measure distances to inaccessible objects.

Known methods for measuring the distance to the target, consisting in the emission of a probe pulse (ZI) with linear frequency modulation of the carrier frequency (LFM) and processing of reflected pulses (OI) either by the correlation-filter method, or by compressing the duration of the OI using a matched filter.

Such methods and devices are described in the technical and patent literature, for example:

Theoretical Foundations of Radar, textbook, ed. Shirman Ya. D, p. 120, 128, 135, "Air Defense Forces of the country", 1968;

Handbook of Radar, vol. 3, ed. M. Skolnik, pp. 400, 403, 383, M. "Soviet Radio", 1979;

FM radar apparatus: US Pat. 5905458, IPC ⁶ G01S 13/42 / Ashihara J .; Honda Giken Kogyo K, K - No. 08/974013; Claim 11/19/97; Publ. 18.5.99; Prior. 11/19/96, No. 8-308440 (Japan): NPK 34270.

These methods due to the use of frequency modulation provide detection of targets at large distances with high resolution in range when the delay time of the OI is longer than the duration of the ZI, i.e. when the OI does not overlap in time with the ZI.

The increase in the duration of the ZI is a means of increasing its energy in conditions when the maximum peak power of the transmitter is reached. The duration of the ZI can be brought up to a value equal to twice the propagation time of the signal to the target at the maximum distance. In order to receive the entire signal from the maximum range, the duration of the reception interval should be no less than the duration of the ZI. In this case, pulses reflected from targets located at shorter distances will begin to arrive even during radiation.

In single-position radars, to protect the receiver from powerful radiation, its input circuits are blocked for the duration of the radiation. As a result of this, a shortening of the duration of the OI occurs at the output of the receiver. For this reason, the signal base decreases, equal to the product of the spectral width and the signal duration. Accordingly, when filtering the OI, the degree of compression of its duration decreases and as a result of this, the resolution and accuracy of the range measurement decrease, which is a drawback of these methods.

In addition, the current target range measurement contains an error due to the Doppler shift of the carrier frequency of the OI.

This error is eliminated by implementing a method that includes sequentially emitting a signal with a LFM carrier frequency with a certain slope and without LFM, measuring the Doppler frequency shift of the signal from the target and calculating the distance to the target by dividing the Doppler shift by twice the slope of the carrier frequency. (See M. Skolnik, p. 383, M. "Soviet Radio", 1979).

However, this method does not solve the problem of reducing the resolution and accuracy of range measurement when superimposing time intervals for the formation of the probing and the arrival times of the reflected pulses.

A known method of sequential emission of a signal with two operating pulse repetition frequencies, determining the temporal position of the pulses reflected from the target at each repetition frequency and determining the true range. This method provides a range measurement in the survey mode in a radar with an average repetition rate, (see. "On-board radar systems" edited by D. Powysil., R. Roven, P. Waterman, Military Publishing House of the USSR Ministry of Defense, Moscow, 1964, pp. .317-320).

The disadvantage of this method is the ambiguity of the range measurement, which is greater, the longer the repetition period. An increase in the repetition period leads to a decrease in the average power of the ZI, which results in a decrease in the range of the radar. Like the previous methods, this method does not provide a constant resolution and accuracy of range measurement when the sounding interval and the arrival time of the reflected pulses are overlapped.

A known method, consisting in a combination of the operations of two methods, linear-frequency modulation and two-frequency. The method consists in emitting pulsed signals with a repetition frequency F1, receiving reflected pulsed signals and measuring their Doppler frequency f1, then emitting signals with linear frequency modulation of the carrier frequency with slope S, receiving the reflected LFM signal, measuring the frequency difference f2 between the emitted and received LFM signals and calculating the range according to a certain formula, measuring the delay of reflected pulse signals t1 at a repetition frequency F1, additional emission of pulse signals with a repetition frequency eniya F2, calculated according to the formula, depending on the possible errors of the measured distance values, receiving reflected signals and measuring their delay t2, a true distance is determined at indicated in the claims. (See "The method of measuring range." RF patent, No. 2145092, G01S 13/02, M, 01/27/2000, authors Babichev V.A .; Rives L.S .; Riman A.I .; Orphan O.A. ; Dubinsky M.L .; Grinberg V.B .; Sinitsyna O.S.).

This method provides high accuracy of range measurement when used in a radar with a high pulse repetition rate. However, it cannot be used in radars emitting long pulses with a low repetition rate. The method requires the emission of a number of pulses, i.e. is not monopulse, which leads to an increase in the time of viewing the space.

As well as the above methods, this method does not solve the problem of reducing the resolution and accuracy of range measurement when the sounding interval and the time of arrival of reflected pulses are overlapped.

Therefore, of all the known methods for providing the required range resolution for detecting targets at large and shorter distances, the first method mentioned above is most suitable as a prototype (see Theoretical fundamentals of radar, a textbook edited by Shirman Y.D., p. 120, 128, 135, "Air Defense Forces of the country", 1968). This method is basic for everyone else, although in its description there are no such components necessary for practical implementation as an amplifier, modulator, mixer, carrier frequency generator, etc.

When using this method, in order to maintain the same resolution and accuracy in the entire range of the range, it would be possible to radiate a ZI with a fixed deviation and with a duration corresponding to each range channel (resolution element). In this case, the number of radiation-to-receive intervals would become equal to the number of range channels, the value of which can reach tens of thousands. Accordingly, the duration of the space survey period would increase by the same number, which makes the use of this method unrealistic.

You can implement the option in which the entire range of the radar range is previously divided into a series of three to five sections of the range. To view the sections, intervals of alternating emission and reception of pulses of the corresponding duration and with the same frequency deviation are organized. For the farthest section, ZI of maximum duration is used. For other sites, shorter ZIs are used. The duration of each ZI is taken, for example, equal to ½ of the duration of the previous ZI. In this case, a predetermined resolution value is provided in the middle of each section, and to the edges of the sections it decreases by 1/3 (in this example), and in the general case, to a value determined by the number and arrangement of sections. With this method of operation in the radar, the unevenness of the resolution in range is maintained, although reduced. The introduction, in addition to the long “emission-reception” interval, of additional shorter intervals, leads to an increase in the viewing period of the range of not less than 1.5 times, which in turn leads to a proportional increase in the viewing time of the radar’s responsibility sector, which is an undesirable and sometimes unacceptable limiting factor.

In addition, filtering, consistent with the probing LFM signal, provides a range measurement with the error introduced by the Doppler shift of the carrier frequency. The elimination of this error is provided by several measurements, calculating the speed and clarifying the distances in the process of constructing the trajectory. At the same time, the process is delayed by the processing path.

The aim of the invention are:

- providing a constant resolution of measuring distances to targets in the entire range when emitting a single modulated pulse and processing reflected pulses, the duration of which is equal to or less than the duration of the probe pulse;

- providing simultaneous measurement of distances to targets and radial velocities of targets

The implementation of these goals is achieved by using the method below.

In the method, which consists in emitting one LFM pulse and processing the reflected pulses (OI) with a matched compression filter, in order to achieve the goals of the present invention, the emitted pulse is generated in the form of a digital sequence of complex signal samples in several time sections, successively decreasing one after another duration, inside each section, the signal is frequency-modulated with a change in the modulation frequency from zero to the deviation frequency Fdev, samples the signal is calculated with a sampling frequency Ff> 2Fdev, the calculated samples in each section are rearranged in time within the section, thus forming a vector of a quasi-random phase-modulated sequence of samples; with the beginning of the emission cycle, this sequence is converted by a digital-to-analog converter into a sequence of voltage pulses, which is passed through a low-pass filter, limiting the frequency band of the signal to the band ΔFм≥Fdev, from the output of the low-pass filter, the signal is transferred to the carrier frequency f0, amplified by power and through the antenna the switch is transmitted to the antenna for radiation; the pulses reflected from the targets received by the antenna in the reception cycle are amplified, filtered at the carrier frequency f0 by a band-pass filter with a passband ΔFm, time discretized with a frequency Fd≥2ΔFm with the condition f0 = nFd + ΔFm / 2, n = 1, 2, 3, ..., and are quantized by the level by an analog-to-digital converter, then the actual digital samples of the reflected signals are transferred to the zero frequency by digital quadrature heterodyning, the complex results of the heterodyning are passed through a digital complex filter of the lower frequencies with a passband ΔFm, a sequence of complex samples is formed at the output of the low-pass filter with a repetition rate F _c ≥ΔFm, which are then processed by a compression filter whose impulse response in terms of duration, phase modulation function and expected Doppler frequency shifts is consistent with the reflected pulse; the delay of the maximum signal amplitude obtained at the filter output corresponds to the distance to the detected target, and the frequency of the impulse response at which the maximum amplitude is obtained corresponds to the Doppler frequency shift of the target, which is uniquely associated with its radial velocity.

For a visual representation of the possibility of ensuring a constant resolution over the entire range when emitting one composite probe pulse, Fig. 1 shows such a ZI, as well as the pulses received at the input of the compression filter reflected from targets at different distances. Dotted lines show the time intervals of the plots. The case of equality of the intervals of radiation T _ZI and reception T _PR . It can be seen from the figure that at any range, the width of the spectrum of the optical radiation is equal to the range of modulation frequencies F _m ≥F _dev , which ensures the preservation of a constant resolution over the entire range. As an example of a possible implementation of this method, figure 2 shows a simplified block diagram of the radar.

In figure 2 is indicated:

1 - digital generator of a composite chirp pulse;

2 - RAM quasi random samples ZI;

3 - quasi random address generator;

4 - digital-to-analog converter (DAC);

5 - analog low-pass filter (low-pass filter);

6 - quadrature modulator;

7 - carrier frequency generator;

8 - power amplifier;

9 - antenna switch;

10 - analog-to-digital receiver;

11 - high frequency amplifier;

12 - band-pass filter;

13 - analog-to-digital Converter (ADC);

14 - multiplier;

15 - RAM coefficients of the digital local oscillator;

16 - digital integrated low-pass filter (DSP);

17 - RAM coefficients DSC;

18 - matched pulse compression filter with quasi-random phase modulation;

19 - buffer RAM input samples;

20 - address counter;

21 - a multiplier of complex numbers with an accumulating adder;

22 - RAM weight coefficients of the compression filter;

23 - detector;

24 - buffer RAM coefficients of phase modulation;

25 - generator of a complex sine wave of Doppler frequency;

26 - multiplier of complex numbers;

27 - block synchronization and control.

The operation of the radar is as follows.

In block 1, before the start of radiation, complex digital multi-bit codes of frequency-modulated samples of the ZI are described, described by expression (I):

$$\text{z}\left(\text{n}\right)=\text{e}\text{xp}\left\{\text{j2π}{\gamma }_{\text{m}}{\left[\left({\text{nN}}_{\text{hm}}\right){\text{T}}_{\text{f}}\right]}^{\text{2}}\text{/ 2}\right\}\text{,}\text{n}=\text{0}÷{\text{N}}_{\text{ZI}}\text{-one}\text{,}{\gamma }_{\text{m}}=F_{de\mathrm{at}}/T_m\left(\text{one}\right)$$

where z (n) is the value of ZI at the moment of time nT _f , counted from the beginning to the end of the radiation;

exp {...} is the complex sine wave, exp {...} = cos {...} +) jsin {...};

n is the number of the cycle of formation of ZI from the beginning of radiation;

T _f - the cycle of signal formation, T _f = 1 / F _f ;

m is the signal section number, m = 1 ÷ M;

T _m - the duration of the m-th section of the signal;

F _dev - the value of the deviation of the modulation frequency;

γ _m is the rate of change of frequency on the mth portion of the signal;

N _ZI - the number of cycles of discretization (formation) of ZI. This is the number of samples ZI, N _ZI = T _ZI F _f ;

- N _nm - the measure number corresponding to the beginning of the m-th section of ZI.

The samples calculated in block 1 within each section of the signal, using the generator 3 quasi random natural numbers (addresses) are written RAM 2, so that the samples from the output of the chirp generator are randomly rearranged within the chirp sections. After that, the entire array of samples of the ZI is recorded in the buffer RAM 24 of the compression filter 18 for the subsequent calculation of the weight coefficients of the filter. On the radiation interval at time points counted from the start of the radiation and set by the clock pulses of the synchronization and control unit 27, sequential reading of the ZI samples from RAM 2 is performed. Due to the quasi-random arrangement of the samples in RAM 2, a sequential reading of the addresses at the output of the RAM 2 generates a complex quasi-random sequence samples with different phase values, which is fed to the inputs of the two-channel block 4 of the digital-to-analog converter. In block 4, the codes of the samples of both quadratures are converted to voltages, which are filtered by the dual analog low-pass filter in block 5. The low-pass filter allocates a modulation frequency band ΔF _m ≥F _dev . The quadrature modulation signals from block 5 are fed to the first two inputs of the quadrature modulator 6, the two second inputs of which are supplied from the generator 7 by the sine and cosine components of the harmonic signal of the carrier frequency f _0, respectively. At the modulator output, the carrier signal will have quasi random phase modulation (FM). From the output of the modulator, the signal is fed to the power amplifier 8 and through the antenna switch 9 is fed to the radiation in the antenna. The antenna switch 9 connects the antenna to the output of the power amplifier 8 for the duration of the radiation and connects the antenna to the input of the receiver 10 at the time of reception. The signals received by the antenna reflected from the targets through the switch 9 are sent to an analog-to-digital receiver 10. In the receiver they are amplified by an amplifier 11, filtered by a band-pass filter 12, the passband of which is equal to the modulation frequency band ΔF _m , time-sliced with a frequency F _d ≥2ΔF _m ensuring the fulfillment of the condition f ₀ = nF _d + ΔF _m / 2, n = 1, 2, 3, ..., and are quantized by the level in the ADC 13. For clarity, figure 3 shows the nature of the conversion of the signal spectra in the receiver. The obtained actual signal samples are multiplied in block 14 by samples of a complex sinusoid of frequency ΔF _m / 2, arriving at a clock cycle T _d = 1 / F _d from RAM 15, which performs the function of a digital local oscillator. The multiplication results are quadrature components, they are filtered in block 16 by a digital complex low-pass filter with a passband ΔF _m . The coefficients of the impulse response of the DPSF are stored in RAM 17 and fed to the second inputs of block 16. At the output of the DPSF, samples are generated with a frequency F _c ≥ΔF _m in accordance with the DPSF band ΔF _m and the complex shape of the samples. These samples are transmitted to the compression filter 18, where they are written to the buffer RAM 19, from where they are read alternately for each range and Doppler frequency channels using the control codes from the address counter 20 and are sent to the complex number multiplier 21 with an accumulating adder that performs the basic operation of matched filtering - coherent signal accumulation. Thus, the processing of the received signal in different Doppler channels is achieved by sequentially calculating the convolution function in the filter 18 for different values of the Doppler frequency generated by block 25. The number of Doppler channels is determined by the desired accuracy of determining the radial velocity of the target. The minimum and maximum expected Doppler shift are determined by the expected speed range of the detected targets. Processing of the received signal in different range channels is carried out for different delay values of the received signal relative to the beginning of the reception cycle in the range from 0 to T _ZI . The matched filter generates the output sample of the compressed reflected pulse for each range channel and for each Doppler frequency by calculating the convolution of the input samples of the OI with the filter weight coefficients, formed taking into account the duration and type of modulation function of the signal coming from the expected range limit, as well as taking into account the expected Doppler frequency offsets. The vectors of the weight coefficients of the signal compression filter in range and for all expected Doppler shifts of size N _ZI coefficients each are calculated on the radiation interval by multiplying in block 26 a quasi random sequence of complex samples of the phase modulation function of the ZI read from buffer RAM 24 by the complex sinusoids of the corresponding Doppler frequencies, generated by the generator 25. Multiplication results - an array of weight coefficients of the compression filter in range and for all Doppler frequencies, writes There are 22 compression filter weights in RAM.

At the output of the matched filter, samples of the convolution function are formed:

$$y_f\left(n\right)={\displaystyle \sum _{k=0}^{Ns\mathrm{and}-\mathrm{one}}x(n-N_R+k)z*(k)\exp (-\mathrm{<figure-callout\; id="2"\; label="j"\; filenames="00000007.png,00000008.png"\; state="\{\{state\}\}">j</figure-callout>}2\pi f_{d\mathrm{about}Pl}k{T}_c)},n={\mathrm{<figure-callout\; id="3"\; label="N"\; filenames="00000008.png"\; state="\{\{state\}\}">N</figure-callout>}}_{3\mathrm{AND}}÷2{\mathrm{<figure-callout\; id="3"\; label="N"\; filenames="00000008.png"\; state="\{\{state\}\}">N</figure-callout>}}_{3\mathrm{AND}}\text{-}\mathrm{one},\left(2\right)$$

where n is the cycle number of the samples of the signal T _c at the input / output of the compression filter, measured from the beginning of the radiation interval;

T _c is the clock cycle of the signal samples at the input / output of the compression filter, T _c = 1 / F _c . The frequency F _c must satisfy the condition F _c ≥F _m ;

z * (k) are the complex conjugate readings of the phase modulation function of the ZI;

N _ZI - the number of samples in the interval of duration ZI Tzi;

f _dopl - the value of the expected Doppler frequency shift of the OI;

N _R - the number of ticks T _c from the beginning of the reception cycle, corresponding to the delay time of the OI.

The implementation of formula (2) is easier to present in the figure in Fig. 4, where two arrays are conventionally shown as rectangles: the 1st rectangle is a register with an array of input samples, the 2nd rectangle is a register with an array of samples of the impulse response (weight function) of the filter. The 1st rectangle is fixed, the 2nd rectangle moves from left to right (or vice versa) with each measure T _c .

Before calculating the next output sample, the rectangle is set to the place corresponding to the time of arrival of the reflected pulse from a given distance. Each output sample is obtained by calculating the sum of the products of the samples in the overlapping parts of the rectangles. Thus, only those weighting coefficients that correspond to the signal samples coming from the corresponding range, i.e. the filter is adaptive to the processed signals from different ranges. The processing of data read from the buffer RAM 19 to block 21 consists in multiplying them by the weight coefficients coming from the RAM 22 and summing the products over the interval determined by the order (number of samples) of the weight function of the compression filter.

After summing the products and completing the accumulation cycle, the quadrature components are transmitted to the detection unit 23, where the amplitudes of the output signal in each range channel are calculated by the formula: $$U_f\left(n\right)=\left|y_f\left(n\right)\right|\left(3\right)$$

The analysis of the amplitude relief on the plane "delay-Doppler frequency" allows you to determine the delay of the OI from the maximum amplitude, i.e. distance to the target, and Doppler shift, i.e. radial velocity of the target. In the graphs of amplitude reliefs and frequency characteristics below

$$S_f\left(n\right)=\mathrm{twenty}Lg{U}_f\left(n\right)/U_{f\max }\left(\mathrm{four}\right)$$

In confirmation of the achievement of the goals when using this method, modeling was carried out using the Matlab system. In the simulation the following initial parameters have been taken: the length of the GI and the interval T _connection receiving the OI = 5 ms; the value of the frequency deviation F _dev = 200 kHz; the number of sites in the ZI of different duration M = 11; the duration of the first section T ₁ = 0.5 ms; the duration of the remaining sections was initially determined by the expression T _m = T ₁ λ ^m-1 with subsequent rounding, a multiple of the cycle of readings T _c . The value of λ = 0.9807; the sampling frequency of the OI and the repetition rate of complex signals at the output of the receiver F _d = F _c = 500 kHz (cycle T _c = 2 μs); the number of range channels N _R = T _zi F _c = 2500.

Due to the quasi-random phase modulation, the spectrum of which during modeling was limited to Vi of the sampling frequency (250 kHz), the compression ratio of the signal (signal base) was B = T _zi F _d = 1250.

In a matched filter, the reflected pulse with a duration T _oi = 5 ms is compressed to a duration T _cf = T _zi / B = 4 μs. This value may be an estimate of resolution over the entire range. The simulation results are presented by graphs of the amplitude reliefs of the signals at the outputs of matched filters.

Figure 5 shows the amplitude profile of the compressed signal at the output of the filter with a zero tuning frequency (fph = 0 Hz) reflected from a stationary target near the maximum range (Rmax).

Figure 6 shows the amplitude relief of the signal in the same filter from a fixed target located near the minimum range.

Figure 7 shows the output of the filter with f = 0 Hz the relief of two compressed OI from stationary targets located near the minimum and maximum ranges. The amplitudes of both OIs at the input of the compression filters were set equal. At the output of the short-range filter, the signal amplitude, as expected, is approximately 25 dB less than the amplitude of the signal with a greater range due to a decrease in the pulse duration and, accordingly, a decrease in the order of the filter.

On Fig in an enlarged scale depicts the relief of the OI from a stationary target near the maximum range. The base pulse duration at -14 dB is 2 samples of the sampling frequency, i.e. 4 μs. The reference number of the range channel (peak amplitude) is 2250.

Figure 9 on an enlarged scale presents the relief of the OI from a fixed target near the minimum range. The base pulse width of -14 dB is also 2 samples of the sampling frequency, i.e. 4 μs. The peak amplitude reference number is 123.

Figures 10 and 11 at different scales depict reliefs of compressed OI from two stationary targets located near the maximum range and spaced apart from each other in time by 2 counts. Peaks of amplitudes are clearly visible, the distance between which is 4 μs. This is the value of resolution over time (range).

12 and 13 depict reliefs of compressed OI from two stationary targets located near the minimum range and spaced apart from each other by time also by 2 counts. It can be seen that the distance between the two peak amplitudes is 4 μs. That is, at the minimum range, the resolution in range is the same as at the maximum.

On Fig shows a characteristic of the frequency selectivity of the compression filter for a signal reflected from the target with a maximum range with zero tuning frequency in the range of positive Doppler frequencies from 0 to 1 kHz. The full filter bandwidth at -3 dB is 200 Hz.

On Fig depicts the same characteristic of frequency selectivity in the frequency range from 0 to 20 kHz.

On Fig shows a characteristic of the frequency selectivity of the compression filter for a signal reflected from the target at the minimum range with zero tuning frequency in the range of positive Doppler frequencies from 0 to 5 kHz. The full filter bandwidth at -3 dB is 4 kHz.

17 shows the same characteristic of frequency selectivity in the frequency range from 0 to 20 kHz.

In Fig.18 shows the amplitude relief in the filter with a tuning frequency f _f = 5 kHz of the reflected pulse from a moving target located near Rmax.

On Fig and 20 at different scales shows the characteristic of the frequency selectivity of the compression filter with the setting f = 5 kHz for a moving target at maximum range. It can be seen that, as in Fig. 14, the passband at a level of -3 dB is 200 Hz.

On Fig presents the amplitude relief in the filter with GF = 5 kHz reflected pulse from a moving target, located near Rmin.

On Fig shows a characteristic of the frequency selectivity of the compression filter with the setting GF = 5 kHz for a moving target located at the minimum range. It can be seen that, as in FIG. 14, the passband at the -3 dB level is 4 kHz.

On Fig shows the amplitude relief of the OI in the filter with a tuning frequency of 5 kHz from a moving target near the maximum range. It can be seen that the signal was allocated in the same range channel with the reference number of the peak amplitude 2250, as was the case with the filter with Гф = 0 Hz (see Fig. 8).

On Fig shows the amplitude relief of the OI in the filter with a tuning frequency of 5 kHz from a moving target near the minimum range with the reference number of the peak amplitude 123, as was the case with the filter with Gf = 0 Hz (see Fig. 9).

Thus, the proposed method, which consists in the formation of a probing composite quasirandom phase-modulated monopulse and processing reflected pulses, the duration of which is less than or equal to the duration of the probing pulse, matching compression filters will ensure the detection of reflected signals in the entire range of ranges and Doppler frequencies and measure distances to objects with a constant structurally preset resolution without Doppler error, regardless of the location of targets at lnosti and provide simultaneous measurement of Doppler frequency offsets.

Literature

1. Theoretical foundations of radar, textbook, ed. Shirman Y.D., p. 120, 128, 135, "Air Defense Forces of the country", 1968;

2. Guide to radar, t.3, ed. M. Skolnik, pp. 400, 403, 383, M. "Soviet Radio", 1979;

3. FM radar apparatus: US Pat. 5905458, IPC ⁶ G01S 13/42 / Ashihara J .; Honda Giken Kogyo K, K - No. 08/974013; Claim 11/19/97; Publ. 18.5.99; Prior. 11/19/96, No. 8-308440 (Japan): NPK 34270.

4. "Airborne Radar Systems", ed. D. Powysil, R. Roven, P. Waterman, Military Publishing House of the USSR Ministry of Defense, Moscow, 1964, pp. 317-320.

5. "Method of measuring range." RF patent, No. 2145092, G01S 13/02, M, 01/27/2000, authors Babichev V.A .; Rives L.S .; Riemann A.I .; Orphan O.A .; Dubinsky M.L .; Greenberg V.B .; Sinitsyna O.S.

Claims (1)

A method for measuring distances and radial velocities of targets with constant range resolving power in radar devices over the entire range of the range, which consists in emitting a single long pulse with a quasi-random phase modulation in the radiation cycle and processing the pulses reflected from the targets in the receiving cycle with an agreed compression filter, characterized in that the emitted pulse is formed in the form of a digital sequence of complex samples of the signal at several time sections following one another and another with successively decreasing duration, within each signal portion is frequency-modulated with the change of the modulation frequency of zero to _nine F frequency deviation, signal samples are calculated with a frequency _f F deskretizatsii ≥2F _ninth calculated counts in each area are permuted over time within a site , thus forming a vector of a quasi-random phase-modulated sequence of samples; with the beginning of the emission cycle, this sequence is converted by a digital-to-analog converter into a sequence of voltage pulses, which is passed through a low-pass filter that limits the frequency band of the signal to a band of F _f ≥2F _dev , from the output of the low-pass filter, the signal is transferred to the carrier frequency f ₀ , amplified by power, and through the antenna switch is transmitted to the antenna for radiation; the pulses reflected from the targets received by the antenna in the reception cycle are amplified, filtered at the carrier frequency f _{0 with a} band-pass filter with a passband ΔF _m , time discretized with a frequency F _d ≥2ΔF _m with the condition f ₀ = nF _d + ΔF _m / 2, n = 1, 2, 3, ..., and are quantized by the level of an analog-to-digital converter, then the actual digital samples of the reflected signals are transferred to the zero frequency by digital quadrature heterodyning, the complex results of the heterodyning are passed through a digital complex filter izhnih frequencies with a bandwidth ΔF _m, the output of the lowpass filter is formed by a sequence of complex samples with a repetition frequency F _c ≥ΔF _m, which are then processed by the compression filter, whose impulse response of duration, the phase modulation function and the expected Doppler frequency shifts compatible with reflected pulse; in this case, the delay of the maximum signal amplitude obtained at the filter output relative to the beginning of the reception cycle corresponds to the distance to the detected target, and the frequency of the impulse response at which the maximum amplitude is obtained corresponds to the Doppler frequency shift of the target, which is uniquely associated with its radial velocity.

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