RU2426143C1 - Method of phase direction finding and phase direction finder to this end - Google Patents

Abstract

FIELD: physics.

SUBSTANCE: proposed phase direction finder comprises receiving antennas, two receivers, reference oscillator, pulse generator, electronic switch, three 90°-phase shifters, three phase detectors, indicator, local oscillator, mixer, intermediate frequency amplifier. Two multipliers, two band filters, delay line, eight square-law function generators, two scaling multipliers, two subtractors, two adders interconnected in some way.

EFFECT: higher sensitivity.

2 cl, 2 dwg

Description

The proposed method and device relates to the field of electronics and can be used to determine the angular coordinates of the radiation source of complex signals.

Known phase direction finding methods and phase direction finders (RF patents Nos. 2,003.131, 2.006.872, 2.010.258, 2.012.010, 2.134.429, 2.155.352, 2.175.770, 2.290.658, 2.311.656; German patents No. 2.127.087, 2.710.955; UK patents No. 1.395.599, 1.598.325; I. Kinkulkin et al. “Phase method for determining coordinates.” M .: Sov. Radio, 1979; “Space Radio Engineering Complexes ". Edited by S. I. Bychkov. M.: Sov. Radio, 1967, p.130-138, and others).

Of the known methods and devices closest to the proposed are the "Phase direction finding method and phase direction finder for its implementation" (RF patent No. 2.290.658, G01S 3/46, 2005), which are selected as prototypes.

These technical solutions eliminate the contradiction between the requirements for measurement accuracy and the uniqueness of reading the angular coordinate α.

A disadvantage of the known methods and devices is the low sensitivity and accuracy when measuring small angular coordinates α of radiation sources of complex signals.

An object of the invention is to increase the sensitivity and accuracy when measuring small angular coordinates of the radiation sources of complex signals.

The problem is solved in that the phase direction finding method, based in accordance with the closest analogue on receiving signals, amplifying and limiting them in amplitude, comparing signals transmitted through two channels, in phase, while the signal of one of the channels is pre-shifted in phase by 90 ° , set in the azimuthal plane n of the receiving antennas around a circle of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receiving antennas are placed along circles, alternately with frequency Ω, the signal received by the antenna located in the center of the circle is converted in frequency, the intermediate frequency voltage is isolated, it is multiplied with signals alternately received by n receiving antennas located around the circle, the phase-modulated voltage is isolated, it is multiplied with the local oscillator voltage , select a low-frequency voltage with a frequency Ω and compare it in phase with the reference voltage, forming an accurate, but ambiguous direction finding scale of the signal source, simultaneously This phase-modulated voltage is subjected to autocorrelation processing, a low-frequency voltage with a frequency Ω is extracted, it is compared in phase with the reference voltage, forming a rough but unambiguous direction finding scale for the signal source, differs from the closest analogue in that the voltage proportional to the sine of the measured angular coordinate is shifted in phase by 90 °, the phase-shifted voltage by 90 ° and the initial voltage are sequentially twice squared, the second-degree phase-shifted voltage by 90 ° and the outcome the second voltage of the second degree is multiplied with each other using a scaling factor of six, the resulting voltage is subtracted from the phase-shifted voltage of the fourth degree and summed with the initial voltage of the fourth degree, the voltage proportional to the cosine of the measured angular coordinate is shifted in phase by 90 °, phase shifted by 90 ° and the initial voltage is sequentially squared twice, the initial voltage of the second degree and phase shifted by 90 ° voltage of the second degree is multiplied They are interconnected using a scaling factor of six, the obtained voltage is subtracted from the initial voltage of the fourth degree and summed with the fourth-degree voltage shifted in phase by 90 °.

The problem is solved in that a phase direction finder, comprising, in accordance with the closest analogue, a first receiving antenna, a first receiver, a mixer, a second input of which is connected to the local oscillator output, an intermediate frequency amplifier, a first multiplier, a first band-pass filter, a delay line, a second phase detector, the second input of which is connected to the output of the first band-pass filter, the first phase shifter 90 ° and the first phase detector, the second input of which is connected to the second output of the reference generator, in series A reference generator, a pulse generator, an electronic switch, n inputs of which are connected to the outputs of n antennas arranged around a circle of radius d with the possibility of electronic rotation around the first receiving antenna located in the center of the circle, and a second receiver, the output of which is connected to the second input of the first multiplier connected in series to the output of the first bandpass filter is a second multiplier, the second input of which is connected to the local oscillator output, a second bandpass filter and a third phase detector, the second in One of which is connected to the third output of the reference generator, as well as an indicator, differs from the closest analogue in that it is equipped with a second and third phase shifters 90 °, eight quadrants, two scaling multipliers, two subtractors and two adders, and to the output of the first phase detector in series the second 90 ° phase shifter, the first quadrator, the second quadrator, the first subtractor are connected, the second input of which is connected through the first scaling multiplier to the outputs of the first and third quadrator c, and the first adder, the output of which is connected to the first input of the indicator, the third quadrator and the fourth quadrator are connected in series to the output of the first phase detector, the output of which is connected to the second output of the first adder, the fifth quadrator, sixth quadrator, and the second are connected in series to the output of the third phase detector a subtractor, the second input of which is connected through the second scaling factor to the outputs of the fifth and seventh quadrators, and a second adder, the output of which is connected to the second input of the indicator, the third phase shifter 90 °, the seventh quadrator and the eighth quadrator, the output of which is connected to the second input of the second adder, are sequentially connected to the output of the third phase detector.

The structural diagram of the phase direction finder that implements the proposed phase direction finding method, is presented in figure 1. The relative position of the receiving antennas 1, 2.i (i = 1, 2, ..., n) and the radio emission source of the IRI is shown in FIG. 2.

The phase direction finder comprises in series a first receiving antenna 1, a first receiver 3, a mixer 12, a second input of which is connected to the output of the local oscillator 11, an intermediate frequency amplifier 13, a first multiplier 14, a first band-pass filter 15, a delay line 16, a second phase detector 17, and a second the input of which is connected to the output of the first band-pass filter 15, the first phase shifter 8 by 90 °, the first phase detector 9, the second input of which is connected to the second output of the reference generator 5. To the first output of the reference generator 5 of the sequence A pulse generator 6, an electronic switch 7, n inputs of which are connected to the outputs of n receiving antennas 2.i (i = 1, 2, ..., n) located on a circle of radius d, and a second receiver 4, the output of which is connected to the second the input of the first multiplier 14. The second multiplier 18, the second input of which is connected to the output of the local oscillator 11, the second pass filter 19 and the third phase detector 20, the second input of which is connected to the third output of the reference oscillator 5, is connected in series to the output of the first bandpass filter 5. To the output of the first phases A second detector phase shifter 21 is connected in series with 90 °, the first quadrator 22, the second quadrator 23, the first subtractor 27, the second input of which is connected through the first scaling multiplier 26 to the outputs of the first 22 and third 24 quadrants, and the first adder 28, the output of which is connected with the first input of the indicator 10. To the output of the first phase detector 9, a third quadrator 24 and a fourth quadrator 25 are connected in series, the output of which is connected to the second input of the first adder 28. The output of the third phase detector 20 pos the fifth quadrator 30, the sixth quadrator 31, the second subtractor 35, the second input of which through the second scaling multiplier 34 is connected to the outputs of the fifth 30 and seventh 32 quadrants, and the second adder 36, the output of which is connected to the second input of the indicator 10 are connected to the output of the third phase the detector 20 is connected in series with the third phase shifter 29 by 90 °, the seventh quadrator 32 and the eighth quadrator 33, the output of which is connected to the second input of the second adder 36.

The proposed phase direction finding method is implemented as follows.

Received complex signals, for example with phase shift keying (PSK):

U ₁ (t) = υ _с · Cos [(ω _c ± Δω) t + φ _k (t) + φ _c ],

U ₂ (t) = υ _c · Cos [(ω _c ± Δω) t + φ _k (t) + φ _c + 2π · d / λ · Cos (Ωt-α)], 0≤t≤T _c ,

where υ _c , ω _s , φ _c , T _c - amplitude, carrier frequency, initial phase and signal duration;

± Δω - instability of the carrier frequency of the signal due to various destabilizing factors;

φ _k (t) = {0, π} is the manipulated component of the phase that displays the law of phase manipulation in accordance with the modulating code, and φ _k (t) = const for k · τ _e <t <(k + 1) · τ _e and can change abruptly at t = k · τ _e , i.e. at the borders between elementary premises (k = 1, 2, ..., N-1);

τ _e , N - the duration and number of chips that make up the signal with a duration of T _c (T _c = N · τ _e );

d is the radius of the circle on which the receiving antennas are located 2.i (i = 1, 2, ..., n) (measuring base);

Ω is the speed of electronic rotation of the receiving antennas 2.i (i = 1, 2, ..., n) around the receiving antenna 1;

α - bearing (azimuth) to the signal source,

from the outputs of the receiving antennas 1, 2.i (i = 1, 2, ..., n), directly and through the electronic switch 7 are fed to the inputs of the receivers 3 and 4, and then to the first inputs of the mixer 12 and the multiplier 14, respectively. The second input of the mixer 12 from the output of the local oscillator 11 receives voltage

U _g (t) = υ _g Cos (ω _g t + φ _g ).

At the output of the mixer 12, voltages of combination frequencies are generated. The amplifier 13 is allocated the voltage of the intermediate (differential) frequency

U _pr (t) = υ _pr · Cos [(ω _pr ± Δω) t + φ _k (t) + φ _pr ], 0≤t≤T _c ,

where υ _pr = 1/2 · υ _c · υ _g ;

ω _CR = ω _with -ω _g is the intermediate frequency;

φ _CR = φ _s -φ _g ,

which is fed to the second input of the multiplier 14. At the output of the multiplier 14 is formed phase-modulated (FM) oscillation at a frequency ω _{g of the} local oscillator 11

U ₃ (t) = υ ₃ · Cos [ω _g t + φ _g + 2π · d / λ · Cos (Ωt-α)], 0≤t≤T _c ,

where υ ₃ = 1/2 · υ _c · υ _pr ,

which is allocated by the band-pass filter 15 and supplied to the first inputs of the phase detector 17, delay line 16 and multiplier 18. The second input of the latter is supplied with the voltage U _r (t) of the local oscillator 11. At the output of the multiplier 18, a harmonic voltage

U ₄ (t) = υ ₄ · Cos [2π · d / λ · Cos (Ωt-α)], 0≤t≤T _c ,

where υ ₄ = 1/2 · υ ₃ · υ _g ,

which is allocated by the band-pass filter 19 and supplied to the first input of the phase detector 20. The reference voltage is applied to the second input of the phase detector 20 from the third output of the reference generator 5

U ₀ (t) = υ ₀ CosΩt.

At the output of the phase detector 20, a voltage is generated

U _н1 (α) = υ _н1 · Cosα,

where υ _н1 = 1/2 · υ ₄ · υ ₀ .

This voltage is fed to the input of the fifth quadrator 30, which is a multiplier, the two voltage of which is supplied with the same voltage U _n1 (α).

The output of the squared 30 voltage is generated

U ₅ (α) = υ _н1 ² · Cos ² α,

which is fed to the input of the sixth quadrator 31. A voltage is generated at the output of the latter

U ₆ (α) = υ _н1 ⁴ · Cos ⁴ α.

At the same time, the voltage U _n1 (α) from the output of the phase detector 20 is supplied to the input of the phase shifter 29 by 90 °, the output of which is formed

U ₇ (α) = υ _н1 · Cos (α + 90 °) = - υ _н1 · Sinα.

This voltage is supplied to the input of the seventh quadrator 32, at the output of which a voltage is generated

U ₈ (α) = υ _н1 ² · Sin ² α.

This voltage is fed to the input of the eighth quadrator 33, at the output of which a voltage is generated

U ₉ (α) = υ _н1 ⁴ · Sin ⁴ α.

The voltages U ₅ (α) and U ₈ (α) from the outputs of the quadrants 30 and 32 are supplied to the two inputs of the scaling multiplier 34, the scaling factor of which is chosen equal to 6 (K _m = 6). The output of the scaling multiplier 34 is formed voltage

U ₁₀ (α) = 6 · U ₅ (α) · U ₈ (α) = 6 · υ _н1 ² · Cos ² α · Sin ² α.

The voltages U ₆ (α) and U ₁₀ (α) from the outputs of the quadrator 31 and the scaling multiplier 34, respectively, are supplied to the two inputs of the subtractor 35, at the output of which a voltage is generated

U ₁₁ (α) = U ₆ (α) -U ₁₀ (α) = υ _н1 ⁴ · Cos ⁴ α-6 · υ _н1 ² · Cos ² α · Sin ² α.

The voltage U ₉ (α) and U ₁₁ (α) from the outputs of the quadrator 33 and the subtractor 35, respectively, are supplied to the two inputs of the adder 36, the output of which is formed voltage

U ₁₂ (α) = υ _н1 ⁴ · Cos ⁴ α-6 · υ _н1 ² · Cos ² α · Sin ² α + υ _н1 ⁴ · Sin ² α = υ _н1 ⁴ · Cosα.

This voltage is supplied to the second input of the indicator 10. Thus, an accurate, but ambiguous reference scale of the angular coordinate α is formed.

At the same time, the phase-modulated oscillation U ₃ (t) is subjected to autocorrelation processing using an autocorrelator consisting of a delay line 16 and a phase detector 17.

In the phase-modulated voltage U ₃ (t), the quantity mφ = 2π · d / λ, called the phase modulation index, characterizes the maximum value of the phase deviation from the zero value that occurs during the electronic rotation of the receiving antennas 2.i (i = 1, 2, ..., n ) around the receiving antenna 1 (figure 2).

Receiving antennas 2.i (i = 1, 2, ..., n) are switched alternately with a frequency Ω using an electronic switch 7 controlled by an n-phase pulse generator 6. The control pulses are generated by the 6 pulse generator from the harmonic voltage generated by the reference generator 5

U ₀ (t) = υ ₀ · CosΩt.

However, for d / λ> 1/2, the ambiguity of the reference angle α occurs. The elimination of this ambiguity by reducing the d / λ ratio usually does not justify itself, since the main advantage of the wide-base direction finder is lost. In addition, in the range of meter and especially decimeter waves, it is often not possible to take small values of d / λ due to design considerations.

In connection with the stated reason, the problem arises of decreasing the phase modulation index without decreasing the relative size of the measuring base d / λ. This is achieved by autocorrelation processing of the phase-modulated voltage U ₃ (t) using the delay line 16 and the phase detector 17. Moreover, the delay time τ of the delay line 16 is chosen so as to reduce the phase modulation index to

mφ ₁ = 2π · d ₁ / λ,

where d ₁ <d,

where the inequality holds

d ₁ / λ <1/2,

providing unambiguous direction finding of the signal source. The output of the phase detector 17 produces a harmonic voltage

₁₃ U (t) = υ ₁₃ · Cos (Ωt-α), 0≤t≤T c,

where υ ₁₃ = 1/2 · υ ₃ ² ,

which through the phase shifter 8 through 90 ° enters the first input of the phase detector 9, the second input of which from the second output of the reference generator 5 is supplied with a reference voltage U ₀ (t). The output of the phase detector 9 produces a voltage

U _n2 (α) = υ _n2 · Sinα,

where υ _Н2 = 1/2 · υ ₁₃ · υ ₀ ,

which is fed to the input of the second phase shifter 21 by 90 °, the output of which is generated voltage

U ₁₄ (α) = υ _Н2 · Sin (α + 90 °) = υ _Н2 · Cosα.

This voltage is supplied to the input of the first quadrator 22, at the output of which a voltage is generated

₁₅ U (α) = υ _2n ^{2 2} · Cos α.

This voltage is supplied to the input of the second quadrator 23, at the output of which a voltage is generated

₁₆ U (α) = υ _H2 ^{4 4} · Cos α.

At the same time, the voltage U _n2 (α) from the output of the phase detector 9 is supplied to the input of the third quadrator 24, at the output of which a voltage is generated

U ₁₇ (α) = υ _н2 ² · Sin ² α.

This voltage is supplied to the input of the fourth quadrator 25, at the output of which a voltage is generated

U ₁₈ (α) = υ _н2 ⁴ · Sin ⁴ α.

The voltages U ₁₅ (α) and U ₁₇ (α) from the outputs of the quadrants 22 and 24, respectively, are supplied to the two inputs of the scaling multiplier 26, the scaling coefficient K _{m of} which is chosen equal to 6 (K _m = 6). The output of the scaling multiplier 26 is formed voltage

U ₁₉ (α) = 6 · U ₁₅ (α) · U ₁₇ (α) = 6 · υ _н2 ² · Cos ² α · Sin ² α.

The voltages U ₁₆ (α) and U ₁₉ (α) from the outputs of the quadrator 23 and the scaling multiplier 26, respectively, are supplied to the two inputs of the subtractor 27, at the output of which a voltage is generated

U ₂₀ (α) = U ₁₆ (α) -U ₁₉ (α) = υ _Н2 ⁴ · Cos ⁴ α-6 · υ _Н2 ² · Cos ² α · Sin ² α.

The voltages U ₁₈ (α) and U ₂₀ (α) from the outputs of the quadrator 25 and subtractor 27, respectively, are supplied to the two inputs of the adder 28, the output of which is formed voltage

U ₂₁ (α) = υ _Н2 ⁴ · Cos ⁴ α-6 · υ _Н2 ² · Cos ² α · Sin ² α + υ _Н2 ⁴ · Sin ⁴ α = υ _Н2 ⁴ · Cos ⁴ α.

This voltage is supplied to the first input of the indicator 10. This forms a rough, but unequivocal scale of reading the angular coordinate α.

Improving the accuracy of direction finding of the radiation source of complex signals is provided by increasing the measuring base d. A the resulting ambiguity in reading the angular coordinate α is eliminated by using n receiving antennas 2.i (i = 1, 2, ..., n), which are installed in the azimuthal plane along a circle of radius d (measuring base) with the possibility of their electronic rotation with angular velocity Ω around the receiving antenna 1, located in the center of the circle, and autocorrelation processing of the received complex signals.

Moreover, the proposed technical solutions are invariant to instability of the carrier frequency of the received signals, due to their modulation (manipulation) and the width of the spectrum, and the exact and unambiguous measurement of the angular coordinate α is carried out at the frequency Ω of the reference oscillator.

Due to the convolution of the spectrum of a complex FMN signal, it is converted into a narrow-band phase-modulated (FM) voltage, which makes it possible to isolate it with a band-pass filter, filtering out a significant part of noise and interference, i.e. to increase the real sensitivity of the frequency-phase direction finder with a relatively low signal to noise ratio.

Thus, the proposed method and device in comparison with prototypes and other technical solutions of a similar purpose provide increased sensitivity and accuracy when measuring small angular coordinates of radiation sources of complex signals. This is achieved by “reinforcing” the small angular coordinates four times.

Claims (2)

1. The phase direction finding method, based on receiving signals, amplifying and limiting them in amplitude, comparing signals transmitted through two channels in phase, while the signal of one of the channels is pre-phase shifted by 90 ° in phase, set n receiving antennas in the azimuth plane circles of radius d with the possibility of their electronic rotation with an angular velocity Ω around a receiving antenna located in the center of the circle, receive antennas are arranged around the circle, alternately with frequency Ω, the signal received by the antenna is placed at the center of the circle, it is converted in frequency, the intermediate frequency voltage is isolated, it is multiplied with signals alternately received by n receiving antennas located around the circle, the phase-modulated voltage is isolated, it is multiplied with the local oscillator voltage, the low-frequency voltage is isolated with the frequency Ω and the phase is compared in phase with a reference voltage, forming an accurate but ambiguous direction finding scale of the signal radiation source, simultaneously phase-modulated voltage is subjected to autocorrelation processing They isolate a low-frequency voltage with a frequency Ω, compare it in phase with the reference voltage, forming a rough but unambiguous direction finding scale for the signal source, characterized in that the voltage proportional to the sine of the measured angular coordinate is phase-shifted 90 °, phase-shifted by 90 ° the voltage and the initial voltage are sequentially squared twice, phase-shifted voltage of the second degree 90 ° and the initial voltage of the second degree are multiplied among themselves using a scaling factor a factor of six, the resulting voltage is subtracted from the fourth-degree phase-shifted voltage by 90 ° and summed with the initial fourth-degree voltage, the voltage proportional to the cosine of the measured angular coordinate is shifted by 90 ° in phase, 90 ° shifted in phase and the original the voltage is sequentially squared twice, the initial voltage of the second degree and phase-shifted voltage of 90 ° are multiplied with each other using a scaling factor of six, obtained by posal subtracted from the source voltage of the fourth degree, and it is summed with the phase-shifted by 90 ° fourth power voltage. 2. Phase direction finder containing a first receiving antenna in series, a first receiver, a mixer, the second input of which is connected to the local oscillator output, an intermediate frequency amplifier, a first multiplier, a first bandpass filter, a delay line, a second phase detector, the second input of which is connected to the output of the first band-pass filter, the first phase shifter 90 ° and the first phase detector, the second input of which is connected to the second output of the reference oscillator, series-connected reference oscillator, pulse generator o, an electronic switch, n inputs of which are connected to the outputs of n antennas arranged around a circle of radius d with the possibility of electronic rotation around the first receiving antenna located in the center of the circle, and a second receiver, the output of which is connected to the second input of the first multiplier, connected in series to the output the first bandpass filter, the second multiplier, the second input of which is connected to the output of the local oscillator, the second bandpass filter and the third phase detector, the second input of which is connected to the third output of the supports second generator, as well as an indicator, characterized in that it is equipped with a second and third phase shifters 90 °, eight quadrants, two scaling multipliers, two subtractors and two adders, and the second phase shifter 90 °, the first quadrator, are connected in series to the output of the first phase detector , the second quadrator, the first subtractor, the second input of which is connected through the first scaling factor to the outputs of the first and third quadrators, and the first adder, the output of which is connected to the first input ind the third phase detector and the fourth quadrator, the output of which is connected to the second input of the first adder, the fifth quadrator, the sixth quadrator, and the second subtractor, the second input of which is connected to the outputs through the second scaling multiplier, are sequentially connected to the output of the first phase detector the fifth and seventh quadrators, and the second adder, the output of which is connected to the second input of the indicator, are connected to the output of the third phase detector in series the third phase shifter 90 °, the seventh quadrator and the eighth quadrator, the output of which is connected to the second input of the second adder.

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