RU2419811C2 - Radio transducer - Google Patents

Abstract

FIELD: physics.

SUBSTANCE: in compliance with proposed method, modulation pulse repetition rate is varied from period to period of linear frequency modulation to allow attenuating noise signal by connecting, between first mixer, unit of filters with variable adjustment frequency equal to the sum of repetition frequency of modulation pulses and range finding frequency, while, after, second mixer, to allow forcing noise signal out beyond frequency detector aperture range.

EFFECT: higher noise immunity of radio transducer to noise signal with FM frequency matching range finding frequency corresponding to preset range of actuating instruction output range.

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Description

The present invention relates to short-range radar devices using a continuous radiated radio signal with linear frequency modulation according to a sawtooth law and phase modulation within (0 ÷ ΔF) with a periodic sequence of modulating pulses of the meander type, and can be used as a non-contact target sensor for undermining the warhead of a weapon.

The closest to the proposed device of the identified analogues is the "Radar with frequency modulation", given in (1, p. 38). The radar contains a transmitting antenna, a first switch, the output of which is connected to the input of the transmitting antenna, a directional coupler (BUT), the first output of which is connected to the first input of the first switch, a sawtooth voltage generator (GPN) connected in series, a voltage controlled generator (VCO), the output which is connected to the input of the HO, the receiving antenna, the second switch, the first input of which is connected to the output of the receiving antenna, the first mixer, the first input of which is connected to the output of the second switch, and the second input d is connected to the second output of the BUT, the second mixer, the first input of which is connected to the output of the first mixer, the processing unit (BO), the input of which is connected to the output of the second mixer, the control signal unit (BUS), the first output of which is connected to the second input of the second mixer, the second output is connected to the second input of the first switch, and the third output is connected to the second input of the second switch. A radar radiates into space through an antenna transmitting and the first switch a pulse-modulated radio signal with linear frequency modulation (LFM) according to a sawtooth law. From the output of the receiving antenna, the radio signal reflected from the target (object) is fed to the first input of the second switch, which opens at the time the reflected radio signal arrives and closes at the time the first switch is unlocked. The out-of-phase operation of the switches provides the required isolation of the transmission-reception modes. From the output of the second switch, the radio signal is fed to the first input of the first mixer, the second input of which receives the radio signal with the LFM according to a sawtooth law from the second output of the NO. As a result of the conversion, video pulses without a constant component with a repetition rate of modulating pulses and with amplitude modulation by a sinusoidal signal with a difference frequency are extracted at the output of the first mixer. The shape and spectrum of the converted signal are similar to the shape and spectrum of the signal in Fig. 2.25v (2, p. 136) with the difference that the pulse envelope is determined not by the Doppler frequency, but by a difference frequency proportional to the distance to the target. The spectrum of the converted signal contains a component with the frequency of the beat signal and components that are left and right on the frequency of the beat signal from frequencies that are multiples of the repetition frequency of the modulating pulses. At short ranges, the beat signal frequency is in the region of high intensity spurious amplitude modulation (PAM) signal with frequencies that are multiples of the chirp frequency. At the same time, it is impossible to provide the required radar sensitivity and, therefore, select an object with a given probability of correct detection. In order to exclude the influence inside the receiving interference caused by the PAM signal, the repetition frequency of the modulating pulses is chosen much higher than the frequency region where the PAM signal has an unacceptably high intensity. After mixing in the second mixer the reference pulse signal with the output signal of the first mixer, a useful signal with a difference frequency proportional to the distance to the target is extracted at the output of the second mixer. The disadvantage of this device is the low noise immunity from the interference radio signal with amplitude modulation frequency, at which the output signal of the second mixer produces an interference signal with the frequency setting of the frequency detector (BH) in the BO. The interference radio signal at the output of the receiving antenna is written as:

where E _C is the amplitude of the interference signal;

M is the coefficient of amplitude modulation of interference;

Ω _C is the angular frequency (3, p. 41) of the amplitude modulation of the noise, hereinafter the frequency;

t is time.

The interference signal at the output of the second switch acquires additional pulse modulation in accordance with the periodic sequence of modulating pulses η (t), and we write it in the form:

Where

n is the number of the repetition period of the modulating pulses;

T _M - the repetition period of the modulating pulses.

Using relation (2.IX) given in (3, p. 48), for simplicity of presentation, we represent the sequence of modulating pulses η (t) by the first and third harmonics of the Fourier series:

where Ω _M = 2π / T _M is the frequency of the first harmonic of the modulating pulses;

π = 3.14.

Given relation (4), the interference signal at the output of the second switch is written in the form:

According to the expression (13.42) given in (3, p. 482), the component contains in the current of the mixing diode, which we write in the form:

After the conversion, taking into account the filtering of high-frequency components, the voltage at the load resistance R at the output of the first mixer is written in the form:

It can be seen from the obtained expression that at the mixer output there are components with frequencies kΩ _M , kΩ _C , k (Ω _M ± Ω _C ), where k = 1, 2 .... The signal from the output of the first mixer is fed to the first input of the second mixer, to the second input of which a periodic sequence of pulses without a constant component, represented by harmonics of the Fourier series (4), is supplied as a reference signal. The alternating voltage of the converted interference signal at the output of the second mixer, taking into account the filtering of side components after the transformations, is written as:

It can be seen from the expression obtained that if the interference frequency Ω _C or 2Ω _C coincides with the frequency setting of the BH in the BO, then the actuator will be malfunctioned. At the output of a double balanced frequency converter, the circuit of which is shown in Fig. 13.34 in (3, p. 490), the components from the input radio signal with amplitude modulation are significantly weakened, but in practice, due to the spread of the parameters of the mixing diodes, it is impossible to completely attenuate the signal interference, which means to eliminate the false operation of the radio sensor, which is unacceptable under the conditions of use.

The aim of the invention is the provision of noise immunity of the radio sensor from the interference signal with amplitude modulation. This goal is achieved by the fact that in the device containing the transmitting antenna, the receiving antenna, the first mixer, the second mixer, GPN, VCO, the input of which is connected to the first output of the GPN, BUT, the input of which is connected to the output of the VCO, and the second output is connected to the second input a first mixer, a BUS, the first output of which is connected to the second input of the second mixer, BO, the input of which is connected to the output of the second mixer, an additional phase modulator (FM) is introduced, the first input of which is connected to the first output of the BUT, the second input is connected to the first a bus control unit, and the output is connected to the input of the transmitting antenna, a bandpass filter (PF), the input of which is connected to the output of the receiving antenna, and the output is connected to the first input of the first mixer, the filter unit (BF), the first input of which is connected to the output of the first mixer, the second the input is connected to the second output of the bus, and the output is connected to the first input of the second mixer, and the bus contains a clock generator (GTI), a first frequency divider (DC), the first input of which is connected to the output of the GTI, the second DC, the input of which is connected to the output of the first Dh, and output connected to the first output of the BUS, a coincidence element, the first input of which is connected to the first input, and the second input to the second input of the BUS, a counter whose input is connected to the output of the coincidence element, and the output is connected to the second input of the first RF and second output of the BUS, BO contains connected in series to an amplifier whose input is connected to the first input of the BO, a limiter, BH, the first amplitude comparator (AK), the output of which is connected to the first output of the BO, connected in series to a low-pass filter (LPF), the input of which is connected to the output of D, the second AK, the output of which is connected to the second output of the BO, the BF contains a decoder, the input of which is connected to the second input of the BF, the first, second, third and fourth keys, the first inputs of which are connected to the first input of the BF, and the second inputs are connected respectively to the first , the second, third and fourth outputs of the decoder, the first, second, third and fourth band-pass filters (PF), the inputs of which are connected respectively to the outputs of the first, second, third and fourth keys, the adder, the first, second, third and fourth inputs of which are connected to tvetstvenno with outputs of the first, second, third and fourth PD, and an output connected to the output BF. The proposed device emits a radio signal with a chirp according to a sawtooth law and phase modulation with a discrete phase change within (0 ÷ ΔФ) with a repetition period T _M and the duration of the modulating pulses T _M / 2 with subsequent conversion in the first mixer of the received radio signal and the allocation of its output a sinusoidal signal with a frequency of phase modulation, modulated in amplitude by a sinusoidal signal with a frequency proportional to the distance to the target, highlighting the output of the BF signal of the total frequency with subsequent conversion azovaniem second mixer and the allocation of a sinusoidal signal processing unit with a frequency proportional to the target range, followed by the detection frequency and outputting the executive team (IR). The noise immunity of the radio sensor is ensured by switching the frequency of the phase modulation from period to period of the chirp, connecting the corresponding PF to the BF, tuning the frequency of the interference signal at the output of the second mixer outside the BH selection band.

Comparative analysis with the prototype shows that the inventive device is characterized by the presence of new blocks, the connections between the blocks, as well as the execution of the BUS and BO. Thus, we can conclude that the claimed device meets the criteria of "novelty" and "significant differences".

Figure 1 shows a block diagram of a radio sensor; figure 2 - diagrams explaining the operation of the radio sensor; on figa - normalized amplitude - frequency response (AFC) of the amplifier and the BF at the lower and upper settings; on figb - dependence on the frequency of the voltage at the output of the BH.

The radio sensor contains the GPS 10, VCO 8, the input of which is connected to the first output of the GPN10, BUT 6, the input of which is connected to the output of the VCO8, FM5, the first input of which is connected to the first output of the HO6, transmitting antenna 1, the input of which is connected to the output of FM5, the receiving antenna 2, a band-pass filter 4, the input of which is connected to the output of the receiving antenna 2, the first mixer 7, the first input of which is connected to the output of PF4, and the second input is connected to the second output of HO6, the second mixer 11, BUS3, the first output of which is connected to the second input of FM5 and the second input of the second mixer 11, and the second input is connected to the second output of the GPN 10, BF 9, the first input of which is connected to the output of the first mixer 7, the second input is connected to the second output of the BUS 3, and the output is connected to the first input of the second mixer 11, BO 12, the input of which is connected to the output of the second mixer 11, the first output is connected to the first input of the BUS 3, and the BUS3 contains the GTI13, the first DCH15, the first input of which is connected to the output of the GTI13, the second DCH14, the input of which is connected to the output of the first DCH15, and the output is connected to the first output of the BUS3, counter 16, the output of which is connected to the second input of the first about DC 15 and the second output of the BUS 3, the coincidence element 17, the first input of which is connected to the first input, the second input to the second input of the BUS3, and the output is connected to the input of the counter 16, BO12 contains a series-connected amplifier 33, the input of which is connected to the input of BO12, limiter 32, BH31, the first AK28, the output of which is connected to the first output of BO12, connected in series by the low-pass filter 29, whose input is connected to the output of BH 31, the second AK 30, the output of which is connected to the second output of BO12.

The radio sensor operates in the following sequence. The sawtooth voltage (figa) from the first output of the GPN 10 as a modulating signal is fed to the input of the VCO8. The phase of the LFM of the radio signal at the output of the VCO8 within the linear frequency change is written as:

where ω _N is the lower frequency value of the radio signal;

Δω is the frequency deviation of the radio signal;

T is the duration of the linear section of the sawtooth voltage;

F ₀ - the initial phase of the radio signal.

From the output of the VCO8, the radio signal from the LFM goes to the input HO6, from the first output of which the radio signal goes to the first input FM5, similar to that given in (4, p. 128). The second input FM5 from the first output of the BUS3 receives a periodic sequence of modulating pulses, described by expression (3), while a discrete change in the phase of the radio signal is carried out within (0 ÷ ΔФ). From the output of FM5, the radio signal is fed to the input of the transmitting antenna 1. We write the phase of the emitted radio signal in the form:

where ΔФ is the phase change.

In the future, for simplicity of presentation, we will assume the transmission coefficients of PF 4, the first mixer 7, BF 9, the second mixer 11, amplifier 33 as 1. The radio signal from the output of the receiving antenna 2 through PF 4 is fed to the first input of the first mixer 7. PF 4 is designed to protect the receiving path of the radio sensor from powerful out-of-band interference. The phase of the radio signal at the input of the first mixer 7 is written in the form:

where τ is the delay of the received radio signal.

At the second input of the first mixer 7 receives a heterodyne radio signal from the second output of HO 6 with the phase determined by the relation (9). The phase of the converted signal at the output of the first mixer 7 is written in the form:

Using relations (4, 12), the signal voltage at the output of the first mixer 7 is written in the form:

where A ₀ is the amplitude of the signal at the output of the first mixer 7;

the output of the first mixer 7, determined by the delay of the radio signal;

Using the relations 3.37, 3.38 (3, p. 101, 102) and limiting ourselves to components with frequencies no more than (Ω _R ± 2Ω _M ), we write the signal voltage at the output of the first mixer 7 in the form:

where I ₀ , I ₁ , I ₂ are the Bessel functions, respectively, of the zero, first and second order. The signal from the output of the first mixer 7 is fed to the first input of the BF 9, the second input of which receives a two-digit digital code from the second output of the BUS3. Depending on the digital code, the decoder 18, made on a chip similar to K 155 ID4 (5, p. 132), generates control voltages, when exposed to one of the keys 19, 20, 21, 22, similar to the key shown in Fig. 4.9 in (8, p. 175). The signal from the first input of BF 9 through the public key is fed to the input of the corresponding PF 23, 24, 25, 26. The signal with a frequency (Ω _R + Ω _M ) from the output of the corresponding PF through adder 27, made according to the circuit shown in Fig. 6.37 in (8, p. 316), it goes to the output of BF 9. Taking into account relation (18) and filtering properties of PF 23, 24, 25, 26, we write the signal voltage at the output of BF 9 in the form:

From relation (19) it follows that the maximum amplitude of the signal can be obtained when the function I ₁ (m ₁ ) takes the maximum value, this corresponds to the phase modulation index m ₁ ≈ 2, and ΔФ≈π. The signal from the output of the BF 9 is fed to the first input of the second mixer 11, the second input of which receives modulating pulses from the first output of the BUS3. As a result of multiplying the reference signal defined by expression (4) without a constant component, with a signal determined by expression (19), and filtering high-frequency components, the voltage of the converted signal at the output of the second mixer 11 is written as:

The signal from the output of the second mixer 11 is fed to the input of the BO12 and then to the input of the amplifier 33, tuned to a rangefinder frequency corresponding to a given operating range. The signal from the output of the amplifier 33 through the limiter 32 is fed to the input BH31, similar to that shown in Fig. 7.37 in (9, p. 148) with the output voltage shown in fig.3b. The voltage from the output of BH31 is supplied to the input of the first AK28 and to the input of the low-pass filter 29. At a range of IR output, the high level voltage from the output of the first AK28 (Fig.2d) through the first output of BO12 and the first input of BUS 3 comes to the first input of coincidence element 17, to the second the input of which through the second input of the BUS 3 from the second output of the GPN 10 receives a pulse backward (figb) sawtooth voltage. From the output of the coincidence element 17, the reverse pulse (Fig.2e) is supplied to the input of the counter 16. In this case, the value of the digital code at the output of the counter 16 (Fig.2g) will change by one. The digital code from the output of the counter 16 is fed to the second output of the BUS3 and to the second input of the first DCH15, the second input of which receives clock pulses from the output of the GTI13. The first DCH15 is programmable and executed on a 533IE7 binary counter (10, p. 65) and on a 4-bit full binary adder 533IM6 (10, p. 75) according to the circuit shown in Fig. 4. The first DCH15 works in the following sequence. Clock pulses from the output of the GTI13 are supplied to the counting input C2 (figa), while the counter works for subtraction. After setting in the counter 533IE7 a zero code for a positive voltage drop at the input, and then at the time of a negative voltage drop at the input C2, a negative voltage drop appears at the output of loan B (Fig.5b), which goes to the input for recording information V. the counter is written the number (1 + N), where N is the number from the output of the counter 16, and a positive impulse is formed at the output of the loan B (fig.5b). On figb shows the options when a digital code 1000 or 0100 is received at the counter information inputs. Thus, the pulse repetition rate from the output of loan B is determined by the formula:

where Ω _GTI is the pulse repetition rate from the output of the GTI13;

N = (0 ÷ 3) - the number from the output of the two-digit counter 16.

The pulses from the output of the first DCH15 are fed to the input of the second DCH14 with the division ratio of the input frequency by 2. The periodic sequence of pulses from the output of the DCH14 is fed to the first output of the BUS 3, while the repetition frequency of the modulating pulses is written as:

In accordance with the digital code of the counter 16, the decoder 18 generates control voltages, under the action of which one of the keys 19, 20, 21, 22 opens, and one of the PF 23, 24, 25, 26 is connected, the tuning frequency of which is equal to (Ω _R + Ω _M ). At the end of the LFM period, the first AK 28 in BO12 is activated again and the entire cycle described above is repeated. The time constant of the low-pass filter 29 is chosen so that for (5 ÷ 10) periods of the LFM, the voltage at its output (Fig.2z) exceeds the threshold of the second AK 30, while the second AK 30 is triggered and a high level voltage (Fig.2i) through the second output BO12 as an executive command (IR) is issued to the actuator of the weapon. If an interference signal with an amplitude modulation frequency Ω _С = (Ω _R + Ω _M ) is received at the input of the receiving 2 antenna, then at the output of the BH 31 (Fig.2d) a voltage appears at which the first AK28 is triggered, the counter 16 in the BUS 3 will change digital the code per unit, the pulse repetition frequency will change in accordance with formula (22), and the corresponding band-pass filter in BF 9 will be connected, while due to a change in the frequency of the converted signal at the output of the second mixer 11, the interference signal will go beyond the linear section of BH31 and the voltage on the output goes down to zero (Fig. 2d), the first AK 28 will return to its original state (Fig. 2d), the chirp backward pulse will not pass to the input of the counter 16, its digital code will not change, which means that the modulating frequency will not change pulses and frequency settings BF9. In order for the first actuation of the first AK 28 to occur again, the adversary must also synchronously change the frequency of the amplitude modulation, but for this it is necessary to know the order of change of the repetition frequency of the modulating pulses, which is not always possible. The presence of an interference signal in one or even two consecutive periods of the chirp is guaranteed not to trigger the second AK30, which means that there will be no false IR output.

Let us give an example of the choice of the GTI13 frequency and a gain in the sensitivity of a phase-modulated radio sensor. We assume that the weapon is approaching the target, then the delay of the received radio signal is written as:

where R is the distance to the target;

V is the velocity of approach of the weapon with the target;

c is the speed of light.

Using relations (15, 23), we write the Doppler frequency in the form:

Choose a rangefinder frequency at which the voltage at the BH31 output exceeds the thresholds of the first AK28 and the second AK30, taking into account the error due to the Doppler frequency not exceeding 10% of the range error, from the ratio:

Choose the width of the linear section BH31 from the ratio:

The minimum value of the phase modulation frequency is equal to Ω _GTI / 8 and must satisfy the relation:

Using the expression in Table 1, line 2 (6, p. 36) for the spectral voltage density of the PAM, which varies linearly, as well as formula 121 given in (6, p. 73), and integrating in the range from Ω _R to (ΩR + ΔΩ _BH ), the energy of the PAM signal at the output of the first mixer 7 is written in the form:

where K is the rate of rise of the sawtooth voltage.

Using the formula for the cube of the sum (7, p. 27), excluding the terms of the second and third order, the energy of the PAM signal at the output of the first mixer 7 is written as:

The ratio of the energy of the received signal to the energy of the PAM signal at the output of the first mixer 7 without phase modulation of the emitted radio signal is written as:

Using the above methodology and performing integration ranging from (Ω _M + Ω _R ) to (Ω _M + Ω _R + ΔΩ _BH ), we write the energy of the PAM signal at the output of BF 9 in the form:

Using relations (20, 31), the ratio of the energy of the received signal to the energy of the PAM signal at the output of the second mixer 11 with phase modulation of the emitted radio signal can be written as:

The gain in the sensitivity of the radio sensor with phase modulation with respect to the radio sensor without phase modulation can be written as:

Taking into account relations (26, 27) and, setting I ₁ (m ₁ ) = 0.55 I ₀ (m ₂ ) = 0.7 (3, p. 100), we determine the gain in the sensitivity of the radio sensor in the form:

Thus, in the considered RF sensor with chirp and an additional phase modulation of the emitted radio signal, noise immunity from the radio signal of interference with amplitude modulation is provided, as well as at small ranges, all other things being equal, the sensitivity is increased by 21 dB.

Information sources

1. “Inventions of the world” No. 24-2004, p.38 “Radar with frequency modulation” (US application No. 313149 of December 6, 2002).

2. P.A. Bakulev “Radar of moving targets”, Sovetskoe Radio Publishing House, Moscow. - 1964.

3. I. S. Gonorovsky "Radio-technical circuits and signals", publishing house "Soviet Radio", Moscow. - 1964.

4. N.T. Bova et al. "Microwave Control Devices", Technika Publishing House, Kiev - 1973.

5. V.L.Shilo “Popular digital microcircuits”, publishing house “Metallurgy”, Moscow. - 1988.

6. A. L. Zinoviev, L. I. Filippov, “Methods of Analytical Expression of Radio Signals,” Higher School Publishing House, Moscow. - 1966.

7. “Handbook of Mathematics, Mechanics and Physics”, the twelfth edition, the publishing house “Science and Technology”, Minsk. - 1973.

8. “Telemetry Handbook” edited by E.L. Grünberg, Mashinostroenie Publishing House, Moscow. - 1971.

9. “Radio receivers” edited by prof. A.P. Zhukovsky, Higher School Publishing House, Moscow. - 1989.

10. Reference "Digital Integrated Circuits". Authors: P.P. Maltsev, N.S. Dolidze, M.I.Kritenko et al., Radio and Communications Publishing House, Moscow. - 1994.

11. "Radio engineering". "Encyclopedia". Edited by Yu.L. Mazor, E. A. Machusky, V. I. Pravdy, Moscow, Dodeka XXI Publishing House, 2002.

12. “Handbook of Radio-Electronic Systems”, Volume 2, edited by Ph.D. B.Kh. Krivitsky, Moscow, "Energy", 1979.

Claims (1)

A radio sensor comprising a transmitting antenna, a receiving antenna, a directional coupler, a first mixer, the second input of which is connected to the second output of the directional coupler, a sawtooth voltage generator, a voltage controlled generator, the input of which is connected to the first output of the sawtooth voltage generator, and the output is connected to the directional input coupler, a block of control signals connected in series with a second mixer, the second input of which is connected to the first output of the block of control signals, the block characterized in that the second output of the sawtooth generator is connected to the second input of the control signal block, the first output of the processing block is connected to the first input of the control signal block, a phase modulator is added to the radio sensor, the first input of which is connected to the first output of the directional coupler, the second input connected to the first output of the control signal block, and the output connected to the input of the transmitting antenna, a bandpass filter, the input of which is connected to the output of the receiving antenna, and the output is connected nen with the first input of the first mixer, the filter unit, the first input of which is connected to the output of the first mixer, the second input is connected to the second output of the control signal unit, and the output is connected to the first input of the second mixer, and the control signal unit contains a clock generator, a first frequency divider the first input of which is connected to the output of the clock generator, the second frequency divider, the input of which is connected to the output of the first frequency divider, and the output is connected to the first output of the control signal block catch, a coincidence element, the first input of which is connected to the first input, and the second input - to the second input of the control signal block, a counter whose input is connected to the output of the coincidence element, and the output is connected to the second input of the first frequency divider and the second output of the control signal block, the processing unit comprises an amplifier connected in series, the input of which is connected to the input of the processing unit, a limiter, a frequency detector, a first amplitude comparator, the output of which is connected to the first output of the processing unit, connected successive low-pass filters, the input of which is connected to the output of the frequency detector, the second amplitude comparator, the output of which is connected to the second output of the processing unit, which is the output of the executive command of the radio sensor, the filter unit contains a decoder, the input of which is connected to the second input of the filter unit, the first, second , the third and fourth keys, the first inputs of which are connected to the first input of the filter unit, and the second inputs are connected respectively to the first, second, third and fourth outputs of the decoder, the second, third, fourth and fourth bandpass filters, the inputs of which are connected respectively to the outputs of the first, second, third and fourth keys, the adder, the first, second, third and fourth inputs of which are connected respectively to the outputs of the first, second, third and fourth band filters, and the output is connected to the output of the filter unit.

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