KR20110118150A - Mixed format media transmission systems and methods - Google Patents

Abstract

Systems and methods of operating a camera are described. The image signal received from the image sensor may be processed as a plurality of video signals representing the image signal. The encoder may combine baseband and digital video signals for transmission over the cable. The video signal may include baseband and digital video signals that are substantially isochronous. The decoder demodulates the upstream signal to obtain a control scene that controls the position and orientation of the camera and the content of the baseband and digital video signals. It receives various signals, provides synchronization information related to the signals, corrects the phase shift offset of the signals, and uses or detects the encoding method used in the signaling. Systems and methods for detecting the presence of a signal are described.

Description

MIXED FORMAT MEDIA TRANSMISSION SYSTEMS AND METHODS

The present invention relates to a multimedia transmission system, and more particularly to a system and method for transmitting standard analog video and high definition digital video over a signal cable.

With the advent of digital broadcast television and streaming video technology, a variety of video cameras, monitors, and video recorders have become available with improved resolution and enhanced features. Currently, closed circuit television (CCTV) systems provide high quality video output and compressed digital video signals for use in applications such as on-premises surveillance, equipment access control and remote monitoring. However, older computer systems still remain and standard analog video signals are widely used and will continue to be used during transmission to all digital and high definition systems. In particular, coaxial cables ("Coax") have been used to transmit signals from CCTV cameras to surveillance stations. Some used CCTV cameras transmit compressed video signals over local or wide area networks, and these cameras may use the Internet Protocol (“IP”) as a communication means for transmitting the compressed video signals.

1 illustrates a conventional system using Coax to deliver standard analog video. Typically, the base analog camera 10 uses a coax 11 to generate a composite video baseband signal (CVBS) that can be transmitted up to 300 meters. Normally a CVBS signal is provided to a video recording system that often includes a digital video recorder ("DVR") that records the CVBS in a digital format. The conventional monitor 14 may be connected to the DVR 12 to simultaneously display standard analog video, which generally has a resolution of 720x480 pixels.

Digital camera 16 may replace analog camera 10 in some applications. Digital camera 16 may support a serial digital interface (“SDI”) that can be used to transmit uncompressed standard digital video to Coal 17 to DVR 12 at approximately 270 Mbps.

2 is a conventional approach to transmitting high definition video (1920x1080 pixels) in currently used systems. First, the digital camera 20 may support a high definition serial digital interface (HD-SDI) that can be used to transmit uncompressed high definition digital video to the DVR 22 via Coax 21 at a 1.5 Gbps rate. Under this high transmission rate, the supported cable distance is up to 100 meters. Second, IP-based high-definition ("HD") camera 24 generates compressed digital HD video signals over 100Mbps Ethernet using standard category 5 ("CAT5") twisted pair cable (25) for distances up to 100 You may. The signal is received by the DVR 22 and recorded for non real time playback. Coax 26 can now be used to transfer video from camera 24 to DVR 22 using CAT5-to-Coax bridge modems 27 and 29 or other conversion devices. The use of a network that allows the camera to transmit digital video allows these systems to add some upstream communication, typically control and audio signals 28.

Certain embodiments of the present invention provide a camera and system and a method of operating the camera. The processor may receive an image signal from an image sensor and generate a plurality of video signals representing the image signal. The encoder is used to combine the baseband video signal and the digital signal as an output signal through the cable. Video signals may include digital video and baseband video that are substantially isochronous. The camera may operate as a closed circuit high definition television camera.

In accordance with certain aspects of the present invention, the baseband signal may comprise a standard analog video signal, and the digital video signal may be modulated prior to combining with the baseband video signal. The digital video signal may comprise a compressed high definition digital video signal. In particular, when a modulated digital signal is provided to the video recorder, the frame rate of the digital video signal may be less than the frame rate of the image signal.

In a particular embodiment, the decoder is configured to demodulate an upstream signal received from a transmission cable used to carry downstream video or from a wireless communication network. The modulated upstream signal is a signal that controls the position and orientation of the camera, a signal that controls the generation of baseband video signals and digital video signals by a processor, and a signal that selects a portion of an image signal for encoding as a baseband video signal. It may include a control signal including a. The control signal may also include a signal that selects a portion of an image signal for encoding as a digital video signal, and an audio signal used to drive the audio output of the camera, such as a loudspeaker.

Certain embodiments of the present invention provide a method of transmitting a video image. The method comprises frequency division multiplexing a video signal received from a high quality image video to obtain a modulated digital signal, generating an output signal by combining the modulated signal with a baseband analog signal representing the video signal, and monitor and digital Simultaneously transmitting output signals to the video storage device. In some of these embodiments, the monitor displays a baseband analog representation of the video signal, and / or the digital video store records a sequence of high quality frames extracted from the modulated digital signal using a digital video recorder. The digital video signal may be compressed.

In a particular embodiment, transmitting the output signal includes providing the output signal to a coaxial cable and / or a wireless transmitter. The input signal received from the coaxial cable or the wireless network may be demodulated to obtain a control signal. The baseband analog signal may be generated by encoding a portion of the video signal in the composite video signal, and the portion of the video signal to be encoded in the composite video signal may be controlled using a control signal. The control signal may control the position of the camera. Demodulating the input signal may further calculate an audio signal from the input signal.

Certain embodiments of the present invention provide a system and method for operating a camera. The processor receives the image signal from the image sensor to generate a plurality of video signals, and the control logic may be configured to respond to the control signal received by the camera, wherein the modulator modulates the digital video signal to obtain a modulated signal. Can be configured. The plurality of video signals may include a baseband video signal and a digital video signal. Each of the plurality of video signals represents at least a portion of the field of view of the camera, and the control signal may control the digital video signal and the baseband content. Typically, the modulated signal and the baseband video signal are transmitted simultaneously by the camera.

Baseband and digital video signals may be substantially isochronous. The encoder can combine the baseband video signal and the modulated signal as an output signal for transmission over the cable. The control signal can be received wirelessly, for example, from a wireless network. The modulated signal may be transmitted at least partially wirelessly. The digital video signal may be a high quality digital video signal or may be a compressed digital video signal. The control signal shifts some of the field of view represented by one of the video signals.

1 illustrates a conventional system using Coax to deliver standard analog video.
2 is a conventional approach to transmitting high quality digital video.
3 illustrates a system for the transmission of analog and digital video in accordance with certain aspects of the present invention.
4 illustrates a network for the transmission of analog and digital video in accordance with certain aspects of the present invention.
5 illustrates bandwidth allocation for transmission of analog and digital video over coaxial cable in accordance with certain aspects of the present invention.
6 shows an example of CCTV equipment built in accordance with certain aspects of the present invention.
7 shows an example of a modem used in DVR equipment constructed in accordance with certain aspects of the present invention.
8 illustrates an example of a modem used in network switch equipment constructed in accordance with certain aspects of the present invention.
9 is an example of a frame structure used in an ATSC digital television.
10 is an example of a conventional frame synchronization packet.
11 is an example of a data segment of a conventional data frame.
12 provides a simplified diagram of a frame arrangement.
13 is a block diagram of a modulator in accordance with certain aspects of the present invention.
14 is a block diagram of a frame structure used in a particular embodiment of the present invention.
15 illustrates the operation of the convolutional byte interleaver in certain embodiments of the present invention.
16 is a block diagram of an optional code rate punctured lattice coded modulation used in a particular embodiment of the present invention.
17 shows an example of QAM matching.
18 shows a frame synchronization / mode packet.
19 is a simplified frame structure used in certain embodiments of the present invention.
20 is a block diagram of a demodulator in accordance with certain aspects of the present invention.
21 is a block diagram of a camera side modem in accordance with certain aspects of the present invention.
22 is a block diagram of a monitor side modem in accordance with certain aspects of the present invention.
23 illustrates a QAM modulator from baseband to passband on the camera side in accordance with certain aspects of the present invention.
24A and 24B illustrate a QAM demodulator from baseband to passband on the monitor side in accordance with certain aspects of the present invention.
25 illustrates a monitor side digital equalizer and a carrier phase / frequency loop in accordance with certain aspects of the present invention.
Figure 26 shows that attenuation is represented as a function of frequency in a coaxial cable.
27A shows the power spectral density (PSD) of the equalizer input.
27B shows the amplitude response of the converged equalizer taps.
28A, 28B, 29A, and 29B show the loss versus slope of a passband digital video signal at different frequencies.
30 illustrates a monitor side modem having a digital equalizer inside a QAM demodulator in accordance with certain aspects of the present invention.
31 illustrates an analog active filter suitable for equalizing baseband CVBS in accordance with certain aspects of the present invention.
32 shows an example of a filter response in a particular embodiment of the present invention.
33A and 33B are QPSK constellations showing rotation in the complex plane.
34 is a block diagram illustrating a phase correction process according to a particular aspect of the present invention.
35 illustrates an integral-proportional (“IP”) filter in accordance with certain embodiments of the present invention.
36 shows transmitted symbols.
37A, 37B, 37C, and 37D show possible recovered symbols based on the transmitted symbols of FIG. 36.
38 shows the phase shift in the received symbol.
39 shows an example of a constellation diagram transmitted based on the conventional real and imaginary parts of the frame synchronization symbol.
40 is a block diagram of a phase offset corrector used in accordance with certain aspects of the present invention.
41 shows a process for determining reliability associated with frame synchronization.
42 illustrates certain aspects of the equalizer and carrier phase / frequency loops used in certain embodiments of the present invention.
Figure 43 illustrates a phase error detection module and slicer used in certain embodiments of the present invention.
Figure 44 illustrates a multiple exponential LUT module used in certain embodiments of the present invention.
45A and 45B show the real part of the equalized output of QPSK signal (FIG. 45A) and 16-QAM signal (FIG. 45B).
46A, 46B, and 46C show QPSK (FIG. 46A), 16-QAM (FIG. 46B), and 64-QAM (FIG. 46B) generated using one embodiment where the constellation is equalized at R = 58. Is a histogram of the power of the equalized output.
47 is an example constellation diagram at the equalizer output and the carrier phase / frequency recovery loop module input.
48 shows an example of QAM matching to the threshold shown.
49 shows one quadrant of all three constellation diagrams superimposed on the same plot.
50 illustrates the operation of one approach to determining constellation degrees.
51A and 51B illustrate a system for simultaneous transmission of standard and high definition video, with a break or tap of a signal, in accordance with certain aspects of the present invention.
52A and 52B illustrate a process of generating frame synchronization pulses from a noise signal in accordance with certain aspects of the present invention.
53 is a block diagram of a camera side modem having a coax connection indicator in accordance with certain aspects of the present invention.
54 illustrates certain aspects of an automatic gain control loop.

Embodiments of the present invention will be described in detail with reference to the drawings provided as illustrative examples to enable those skilled in the art to practice the present invention. In particular, the following figures and examples are not intended to limit the scope of the invention to a single embodiment, but other embodiments are possible by exchanging some or all of the described or illustrated components. Wherever convenient, the same reference numerals will be used to refer to the same or similar parts throughout the hibernation. Where specific components of these embodiments are implemented, in part or in whole, using known components, only those portions necessary for the understanding of the present invention will be described, and other portions of such known components. The detailed description will be omitted so as not to obscure the present invention. In the detailed description, embodiments that represent a single component are not intended to be limiting, and the invention is intended to include other embodiments that include a plurality of identical components or vice versa, unless expressly stated otherwise herein. Furthermore, Applicants do not intend any word in the description or claims to be such meaning that is not generic or expressly stated in a particular sense. In addition, the present invention does not include present and future equivalents to the components referred to herein by description.

Certain embodiments of the present invention provide a system and method for allowing a camera to simultaneously transmit standard analog video and high definition digital video over coax. High definition cameras are applied to generate digital video signals and analog baseband signals. The digital signal is modulated and transmitted in a frequency band separate from the higher frequencies of the baseband video signal. Analog signals may be encoded according to any standard, including the PAL, SECAM, and NTSC standards and variations thereof.

For purposes of detailed description, an example of a system using a secure link ("SLOC") over coax will be described. SLOC is generally described as having upstream and downstream signals to the camera, with the camera being placed upstream. In the detailed description, examples of SLOC systems provide downstream high definition (“HD”) of the first passband, upstream and control signals of the second passband, and downstream CVBS. Other bandpass signals and bandwidth allocations may be used. For example, the system may use two digital video signals of standard or high definition resolution.

3 illustrates an embodiment of the present invention illustrating a particular theory of operation of the present invention. This example illustrates the use of HD camera 30 in a system where it is desirable to view live video generated by camera 30 while recording high quality video copies on DVR 32. Examples of such systems are security or surveillance systems. The function of the HD camera 30 may be controlled remotely as will be described below. The HD camera 30 can be applied to simultaneously generate a high quality signal 332 and an analog CVBS signal 330. In certain embodiments, high definition signal 332 and analog CVBS signal 330 are isochronous, but may be substantially isochronous if the delays in processing different signals are not equal. In one example, the CVBS signal 330 may be delayed due to digital-to-analog conversion overhead. In another example, high quality signal 332 may be extruded and subject to a possible delay based on the compression ratio. In certain embodiments, the CVBS 330 and the high definition signal 332 may be maintained or synchronized in a certain time relationship with the normal audio signal generated by the camera 30.

Camera 30 may be applied by adding binary components or integrating hardware and software into camera 30. For example, a secure link modem ("SLOC-T") 31 via coax is provided inside the camera. The SLOC-T 31 may be built as a modem integrated as an additive to the camera 30 or implemented using a component already integrated in the camera 30. The SLOC-T 31 causes the multimedia supply to be transmitted downstream via communication as shown, and the SLOC-T 31 delivers a plurality of different resolution signals representing the video generated by the camera 30. Is a device that allows signals to be transmitted over coaxial cable 330. For clarity of description, the SLOC used in the transmission device, such as camera 30, will be referred to herein as "SLOC-T", such as a DVR, network, etc. The SLOC provided at the receiving device will be referred to as “SLOC-R.” The description of SLOC-T and SLOC-R is described in more detail below.

The SLOC-T 31 may cooperate with the components of the camera 30 and / or add enhanced functionality to allow the camera 30 to operate in various modes. In one embodiment, camera 30 may generate an uncompressed HD digital video output, and SLOC-T 31 may provide the ability to compress an HD digital video signal. As a result, the SLOC-T 31 may provide performance beyond modulation and demodulation to improve the functionality of the host camera 30 as needed. Thus, some SLOC-T devices can operate in various modes, some of which are provided in an exemplary manner. In one mode, the SLOC-T 31 receives a standard analog signal version and an HD compressed video signal from the camera and transmits both signals through coax 33. In another example, SLOC-T 31 receives a standard analog signal version and an uncompressed HD video signal from camera 30 and transmits a compressed HD digital signal version along with a standard analog signal via coax 33. do. The SLOC-T 31 transmits a standard analog signal and an HD digital signal extracted from the HD signal received from the camera 30.

In a particular embodiment, SLOC-T 31 uses frequency division multiplexing to generate an output signal for transmission on Coax 33. In the example shown in FIG. 5, the downstream digital signal is provided in the signal band of the frequency 52 which is centered on the carrier 53 of the frequency f _Cd . The band of frequency 52 starts above the highest frequency f 0 of the baseband analog signal 50. This band of separated frequencies 52 may be referred to as a channel. Channel 52 may be selected based on the capacity, available bandwidth, signal bandwidth, and other reasons of SLOC-T 31. In some embodiments, channel 52 may be selected for compatibility with receiving equipment. In one example, the signal may be provided directly to a standard television, and channel 52 may be selected to ensure proper separation from the baseband signal. The frequency band of channel 52 may be selected based on the standard for digital video transmission when standard prescribed encoding of the signal is used. It is contemplated that a single digital signal may be transmitted using two or more different channels to convey a portion of the digital signal.

Any suitable modulation scheme may be used to generate a version of the transmittable digital signal. For example, different types of wired and wireless connections may include phase shift keying (PSK), frequency shift keying (FSK), quadrature amplitude modulation (QAM), orthogonal frequency division multiplexing. It may be used with a modulation scheme such as orthogonal frequency division multiplexing (OFDM). The modulation scheme is typically selected based on the medium used for transmission, the factors including the characteristics of the frame rate of the desired video signal, and other factors affecting the available bandwidth of the channel 52.

SLOC-R modem 35 may be provided to a video capture device, such as DVR 32. The SLOC-R modem 35 may receive digital video and CVBS signals. Typically, the CVBS signal is extracted and delivered directly to the display system 33 for a live view of the video image captured by the camera 30. Display system 33 may be a standard monitor, although it may receive a digitized version of the received analog signal. In one embodiment, the SLOC-R modem 35 may generate a digitized version of the analog signal for use with a digital monitor and a suitably equipped computer. Baseband signals can be affected using analog components or using low pass filters that can be implemented through digital signal processing techniques. The digital HD signal may be extracted separately and provided to the recording portion of the DVR 32. In certain embodiments, the digital HD video signal may be compressed on the DVR prior to recording. In many embodiments, the digital HD video signal is received as a compressed digital signal.

In a particular embodiment, the SLOC-T 31 and SLOC-R 35 are configured to support bidirectional communication of the signal. In an example of a secure installation, and as will be described in detail below with reference to FIG. 6, camera 30 includes a control interface 618 that controls microphone 614, loudspeaker 612, sensor 616, electromechanical actuators. , And other features (see FIG. 6). In this example, SLOC-T 31 and SLOC-R 35 are typically configured to communicate control, audio and other data 36 to camera 30.

Referring again to FIG. 5, in one embodiment, upstream data may be communicated to the camera in one or more channels 54 installed at the upper end of the available bandwidth. The channel search, control and audio signals 54 and other data for communication of the digital multimedia signal 52 depend on the available bandwidth, signal-to-noise ratios detected on the channels 52 and 54, signal standards and / or application specific conditions. Can be selected based on this. In some embodiments, channel configuration, bandwidth, and signal-to-noise ratio are determined upon connection of SLOC-T 31 and SLOC-R 35 using a training sequence. Typically, the practice sequence is used to identify the signal capacity of a predetermined or negotiated channel, select the channel 52 for transmission of digital video, and determine the bandwidth available on the selected channel 52. The characteristics of the selected channel 52 may be used to set the compression level for the digital video signal.

In a particular embodiment, the upstream signal 54 includes a signal capable of controlling the center of the downstream 52 and baseband 50 signals. For example, camera optics 600 may provide a fish-eye field of view monitored by camera 60, and the camera processor may select a portion of the image for transmission as baseband signal 50. It may be controlled. Typically, the downstream digital signal 52 can be recorded on the DVR or provide a complete image for further processing. Baseband signal 50 receives baseband signal 50 for live monitoring of the area under surveillance. Baseband signal 50 may include an adjusted image that corrects the visual effects generated by the fisheye lens. The viewer of the baseband signal 50 may shift the field of view within the field of view of the fisheye lens by selecting a new portion of the captured image for viewing. For example, the viewer may request a "pan-right" to shift the view to the right. The data transmitted in upstream signal 54 causes the camera to extract and process the desired portion of the field of view. In certain embodiments, the field of view request included in baseband signal 50 may cause physical movement of camera 60. As a result, the control data of upstream signal 54 may affect the content of both baseband 50 and downstream digital 52 signals.

In certain embodiments, the downstream audio may be transmitted as part of the HD digital video signal and / or as part of the CVBS signal. Some downstream signals may be carried in separate dedicated channels (not shown). In certain embodiments, upstream communication to camera 30 may be handled outside of a communication method, including, for example, a wired or wireless network. Certain embodiments may alternatively or additionally transmit the downstream digital signal 52 wirelessly. As a result, the baseband signal 50 can be transmitted over Coax, with some combination of upstream 54 and downstream 52 being transmitted over the air. Typically, upstream data 54 includes control signals for downstream 52 and baseband 50 signals regardless of the method of transmission.

In a particular embodiment, the cable 33 may be provided directly to the display system 33 for the display of analog standard video. Typically, the standard monitor or display 33 includes filter circuitry that allows selection between the baseband signal and standard modulated TV channels. Eventually, the monitor 330 may discard the high frequency digitally encoded carrier signal. The DVR 32 may also discard the digital video signal without further processing if the digital video signal is transmitted on a standard channel and uses standard digital encoding. SLOC-R 35 decodes the signal generated by SLOC-T 31 to provide decoded HD digital video and other signals to DVR 32. SLOC-R 35 ) May encode control, audio and other data for transmission to camera 30.

With reference to FIG. 4, an embodiment of the present invention is provided that illustrates a particular operating principle of the present invention. 4 shows an example based on a system where it is desirable to view live video generated by camera 40 while simultaneously providing high quality copies of the video over the network via network switch 44. In one example, the HD video feed is captured or streamed using an internal or external IP video server. Typically the camera 40 is adapted to simultaneously produce a high quality signal and an analog baseband video signal. Camera 40 may be applied by adding external components or integrating hardware software such as SLOC-T 400 into camera 40. The SLOC-T 400 may operate in the same manner as the SLOC-R 31 featured in FIG. 3. However, the SLOC-T 400 may be configured to encode the digital video signal in a manner that facilitates forwarding of the digital video signal over the network. For example, the SLOC-T 400 may be programmed or otherwise configured to provide a digital video signal in accordance with the streaming format supported by the IP video server.

The multiplexed video signal transmitted by camera 40 may be received by network switch 44 optionally equipped with SLOC-R 440. Baseband standard analog signals may be extracted and provided to the display 43. In a particular embodiment, the SLOC-R 440 may extract the digital high definition video signal and forward it to a video server or other network device using a suitable network having sufficient bandwidth to carry the digital video signal. The digital HD video signal may comprise a compressed HD video signal. In a particular embodiment, the digital high definition signal extracted by the SLOC-R 440 is compressed or further compressed for forwarding to a video server or other network device. SLOC-R 440 includes hardware and software that records and / or remodulates digital high definition signals for transmission over a network. For example, SLOC-R 440 is encoded in H for communication over Ethernet. It may also generate a -264 signal.

Referring again to FIG. 6, certain embodiments of the present invention provide improved performance applicable to security systems. In the example shown, camera 60 includes a processor and modem SLOC-T 606 configured and adapted to provide a digitally encoded multimedia signal in accordance with certain aspects of the present invention. Sequential images can be captured using a combination of image sensor 602 and optics 600, including combinations of CCD sensors and lens systems known to those skilled in the art. The processor 604 typically receives a scan signal 603 from an image sensor 602 that provides a captured sequential image according to a desired or predetermined frame rate.

In some embodiments, image sensor 602 may generate a digital video signal by converting a scanned analog signal representing an image captured by one or more sensors. For example, image sensor 602 may include an RGB (red, green, blue) sensor, which image sensor 602 internally processes the RGB sensor output as an output 603 of image sensor 602. An encoded color video signal may be generated. In another embodiment, processor 604 may pre-process signal 603 from image sensor 602 to obtain a raw digital video signal. Raw digital video may be further processed by processor 604 to obtain an initial HD digital video signal, whether acquired internally or received from image sensor 602. The analog standard signal may be obtained by processing the raw digital video signal, the output 603 of the sensor 602, or the initial HD digital video signal. Processor 604 may then format the initial HD digital video signal to obtain one or more HD digital video signals that conform to broadcast and other standards. For example, the processor 604 may generate a signal that conforms to broadcast video standards, such as the ATSC and DVB standards. The processor 604 may further compress the digital video signal.

The camera processor 604 may include a combination of commercially available components and custom hardware and software. In one example, a processor may include one or more microprocessors, digital signal processors, microcontrollers, sequencers, and other programmable devices, in combination with supporting logic and memory to perform sequential steps, instructions, and / or programs. It may be. Storage 610 may be used to store computer readable instructions that perform some or all of the functions described in this application in practice. Camera processor 604 may include some built-in or “hard coded” processes that can be used for building certain embodiments of the present invention. The storage 610 may also be used for program scratch memory and / or to maintain configuration information. In certain embodiments, storage 610 may be used to store a record of video captured by camera 60. Thus, storage 610 may be implemented using volatile and nonvolatile memory, optical and magnetic disks, removable electrically removable memory, USB memory drives and other semiconductor, electromagnetic and optical storage devices.

The signal 605 is received from the upstream control, audio and line 62, the video signal provided by the processor 604 to the SLOC-T 606, and forwarded by the SLOC-T 606 to the processor 604. Other upstream information. The upstream audio information may be decoded, processed, and / or formatted by the processor 604 before the audio information is relayed to the loudspeaker, transducer, or other audio output system 612. The processor may amplify the audio signal or use a separate amplifier for the audio output component 612. Upstream control may include optical control 601 and control signals for external devices that are typically provided via control interface 618. The external device may include a motor or actuator used to interpret, rotate or direct the camera 60. Optical control signal 601 and external control signal 618 may be generated in response to a command predefined by the remote control system. For example, the remote user adjusts the joystick to generate a coded series of commands that are interpreted by the camera 604 to mean "rotate the camera 90 degrees clockwise in the horizontal plane," and the processor 604 Responding by sending a series of pulses to a stepping motor installed perpendicular to 60, the series of pulses may cause the camera to cause the desired rotation about the vertical axis of the camera. Similar commands may adjust the focus, zoom, and iris of the optics 600.

In another example, instructions and data may be provided to upstream control information that may be used to control the functionality of processor 604 and / or sensor 602. The command and data may be used to select an area within the field of view of camera 60 for encoding in one or more of the upstream video signals. In certain embodiments, the commands and data provide one or more virtual cameras that can be remotely adjusted to specify the portion of the field of view to be encoded, whereby the portion operates within the actual field of view determined by the optics of the camera 60. To be selected by virtual pan, zoom and tilt. In certain embodiments, processor 604 additionally causes physical movement of the camera, thereby extending the range of pan, tilt, and zoom functions.

In at least some embodiments, the CVBS and digital signals may each carry a portion of an image captured by image sensor 602. Portions of the image may be from different regions within the field of view superimposed or provided by lens 600. Further, in certain embodiments, additional cameras 60 and / or additional image sensors 602 may be used to expand the available field of view. For example, it may be desirable to configure a plurality of cameras to obtain a panoramic (360 °) field of view of the area. One or more processors 604 may provide analog and digital signals representing a field of view or a portion of a field of view. In one example, the complete panoramic view may be provided to a digital signal that may be recorded on the DVR, and the CVBS signal may provide a selectable field of view within the panorama. The selectable field of view may be controlled using zoom, pan, and other controls. In another example, the CVBS and digital signals may provide a common or different part of the panoramic field of view, which part may be independently controlled by the remote viewer.

7 shows an example of the use of the SLOC-R 700 of the secure digital video recording system 70, similar to the SLOC-R 35 described in FIG. 3. System 70 includes a DVR processor 702, an analog video decoder 704, a digital video decoder 708, and an HD digital display processor connected to the SLOC-R 700, peripherals 710, 712, and 714. 706. As noted above, SLOC-R 700 typically receives and decodes signals from Coax 72, including analog standard video signals and HD digital video signals. SLOC-R 700 also transmits upstream audio and control signals over coax 72. SLOC-R typically splits the analog CVBS signal from the HD digital video signal of the input signal 72 to provide a digital video signal to the processor 702, as a live supply from the camera 60 shown in FIG. Provide CVBS to the standard monitor 74. The SLOC-R 700 provides an analog baseband video signal 701 to the analog video decoder 704, which processes the analog baseband video signal 701 to provide a digital standard video signal 705. ) The display processor 706 multiplexes and / or selects between the digital standard signal 705 and the signal 707 derived from the reproduction of the stored HD digital video. The display processor provides the selected signal in a format displayable by the HD television or monitor 76.

The DVR processor 702 receives the digital HD video signal 703 and optionally stores at least a portion of the signal as a record of the video captured by the camera 60. The recording may be on a hard disk drive 714, network storage (not shown), or other optical electromagnetic or semiconductor storage connected via a network interface 710 and / or a USB / Firewire or other local bus 712. May be stored. The recorded video is further compressed to save storage space. The DVR processor retrieves the recorded video and provides the receivable signal 707 using the digital video decoder 708.

FIG. 8 shows an example of the use of SLOC-R 800 of networked security device 80, similar to SLOC-R 440 described in FIG. 4. Device 80 includes SLOC-R 800 and network switch processor 802, which is typically connected to IP video server 86 by a network. As noted above, SLOC-R 800 typically receives and decodes signals from Coax 82, including analog standard video signals and HD digital video signals. SLOC-R 800 optionally transmits upstream audio and control signals via Coax 82. SLOC-R typically splits the analog CVBS signal from the HD digital video signal of the input signal 82 to provide a digital video signal to the processor 802, as a live supply from the camera 60 shown in FIG. Provide CVBS to a standard monitor 84. In a particular embodiment, the SLOC-R 80 is a component that digitizes the CVBS signal 801 for use as a high-definition display 85, and also as a digital display such as a live supply from the camera 60 shown in FIG. (804, 806, etc.). However, a suitably mounted display device or computing device may receive the CVBS signal 801 to perform digitization of the signal. The switch processor 802 receives the digital HD video signal 803 and optionally transmits the received signal to a network video server 86 that can maintain a record of the video captured by the camera 60. The digital HD video signal 803 may be further compressed before being sent to the video server 86.

5 and 6, certain embodiments of the present invention allow the content of the downstream signal 52 and the baseband analog signal 50 to be selected as desired. In one example, baseband signal 50 and downstream signal 52 comprise the same image, the former being in analog form, the latter being digitally encoded. Digital images can be optionally and selectively transmitted at compressed and uncompressed, standard and high definition, and at full frame or at reduced frame rates. In another example, baseband signal 50 provides a portion of the overall image captured by image sensor 602, and downstream signal 52 carries the entire image. In another example, the baseband signal 50 provides a full image provided by the image sensor, with the downstream including some of the full image. As a result, a highly configurable system allows a user of a digital camera to select from a wide range of options for displaying, recording, and transmitting video images.

Analog Equalizer for Baseband Signals

Certain embodiments of the present invention include systems and methods that improve the effect of significant frequency drop in a cable causing more frequency to decrease as the cable length increases. This slope caused by the cable degrades the baseband analog signal and the passband digital video signal, and the degradation worsens as the cable length increases. Certain embodiments of the present invention typically provide a digital receiver with an equalizer to remove the slope on the digital passband signal to enable reliable decoding of the transmitted symbols.

Certain embodiments of the present invention, including the systems described above, improve the performance of systems and apparatus in which baseband video signals are combined with digital representations and control signals of baseband video signals, thereby improving coaxial cable ("coax"). Enable transmission over a signal such as 3 and 4 show examples of embodiments for providing a SLOC system, and FIG. 5 shows one possible modulation scheme for a SLOC system. Referring to the example of FIG. 3, HD camera 30 provides an output that includes compressed digital HD video 332, and secondary camera output 330 includes an analog standard (“SD”) CVBS. The compressed video signal 332 is modulated into the passband 52 using a SLOC camera side modem 31 that includes a QAM modulator that provides a modulated signal that can be combined with the baseband analog CVBS signal 330. The combined signal is typically transmitted downstream via coaxial cable 33 for a distance that can extend over 300 m. On the monitor side, the SLOC monitor side modem 35 separates the signal representing the baseband CVBS signal from the signal representation of the passband downstream video signal 332. The signal representation of the CVBS is fed to the SD display 34 for live viewing without delay. The high passband downstream signal is demodulated by a QAM demodulator that is output to the host processor and DVR 32 that supports live (perhaps slightly delayed) HD viewing on monitor 34 and non-real-time HD playback for later viewing. .

For example, upstream communication is provided, for example, in accordance with the IP protocol. Upstream communication may additionally be used to send audio and camera control signals 334 from the monitor side to the camera 30. Typically the bit rate for the upstream signal and consequently the required bandwidth is significantly lower than the bit rate required for the downstream passband signal and consequently the required bandwidth. The monitor side SLOC modem 35 includes a QAM modulator that modulates the IP signal into the upstream passband 54. As shown in FIG. 5, upstream passband 54 and downstream passband 52 are located at different spectral positions. On the camera side, the SLOC modem 31 includes a QAM demodulator that receives the upstream signal. This approach provides several advantages over conventional systems and methods as follows.

(1) Increased operating range-increased distance.

(2) A system that uses existing infrastructure and can be used by reusing coaxial cables.

(3) Availability of low-delay, real-time (live) video

(4) Live CVBS and HD video can be viewed in separate locations.

21 is a simplified diagram showing additional details of the SLOC camera side modem 49 of FIG. The IP connection to the HD camera 2100 is interfaced to the QAM modulator 212 and the QAM demodulator 214 via a media independent interface (“MII”) module 210. In one example, MII 210 conforms to the IEEE 802.3 standard. QAM modulator 212 operates using known theory of converting baseband IP data stream 2100 into passband QAM symbol 2120. These symbols are summed with the baseband CVBS signal 2160 and supplied to the diplexer 218. Diplexer 218 carries baseband and low passband downstream signals 2162 coupled to coax and receives a high passband upstream signal 2140 from coax and supplies it to QAM demodulator 214 ( 2-way) may be an analog divider. QAM demodulator 214 typically operates using known theories of demodulating the high passband upstream signal 2140 received from the monitor side and outputting baseband data to MII interface 210.

FIG. 22 is a simplified diagram showing additional details of the SLOC monitor side modem 45 of FIG. The diplexer 220 receives the baseband CVBS and low passband IP signals 2200 coupled downstream from the coaxial cable, so that the signal 2200 is componentized by low pass (LP) and high pass (HP) filtering. Divide into elements 2201-2203. The CVBS signal 2201 may be sent directly to a standard monitor or display device. The low passband signal 2202 can be supplied to a QAM demodulator 222 that supplies the MII interface module 226. The diplexer may also receive a high passband signal 2203 from the QAM modulator 224 and may forward this upstream signal to a coaxial cable. Typically the QAM modulator 222 gets its input from the MII interface 226 which can be connected to a host / DVR that supports the IP protocol.

Coaxial cables typically exhibit significant high frequency roll off characteristics that cause more high frequency attenuation as the cable length increases. This "tilt" may be important within the band of the passband signal, and the slope may cause significant internal symbol interference ("ISI"). Digital equalization may be necessary for QAM demodulator 222 to correctly recover the transmitted data.

At baseband To the passband  module

FIG. 23 shows QAM modulator 212 (FIG. 21) from camera side baseband to passband in more detail. For example, an FEC encoder / matcher that adds air protection data to a data stream received from MII 210 using associated concatenated Reed-Solomon coding, byte interleaving, and / or lattice coding. Data is received from the MII 210. The matcher / encoder 2300 demultiplexes the data into streams 2300 and 2302, with a given bit group size for each stream representing the QAM symbol amplitude levels in the real and imaginary directions, respectively. The isolated transmitted QAM pulse is

Given by

Where d _{R , m} and d _{I , m} are determined by two independent message streams, each representing a real part and an imaginary part of a complex QAM symbol, and m = 1 ... M is a cardinality two-dimensional QAM constellation plot. Where M is the modulation carrier frequency and q (t) is the root raised cosine pulse function.

에 의해 주어지며, 여기서 *는 컨벌루션 (convolution)을 나타내며, c(t)는 채널 임펄스 응답이고, υ(t)는 부가적인 화이트 가우시안 잡음이다. 그 결과, A successive series of transmitted QAM pulses s (t) are propagated through the noise multipath channel at a rate of F _s = 1 / T _S. As a result, the received signal at the input to the QAM receiver

Is denoted by convolution, c (t) is the channel impulse response, and υ (t) is the additional white Gaussian noise. As a result,

Where d [n] is a complex transmitted symbol, and f ₀ and θ ₀ are the frequency of the demodulator local oscillator from the receiver passband to the baseband with respect to the transmitter such that f _L0 = f _c -f ₀ , respectively. Phase offset.

In the passband  Demodulator to baseband

FIG. 24A shows QAM demodulator 222 (FIG. 22) in more detail from monitor side notification band to baseband. Signal r (t) may be received from a coaxial cable, for example, sampled at a rate higher than the symbol rate (see 240) resulting in a sampled signal r (nT _samp ). After sampling below:

After downconversion, resampling at symbol rate 1 / Ts, and matched filtering,

는 샘플링된 복소 필터링된 잡음이며, 완전한 심볼 레이트 샘플 타이밍과 결합된, 펄스 형상화 및 정합된 필터링 q 로 인해,

는 채널 임펄스 응답 c 만으로 인한 것임을 가정한다. Acquire it. here

Is sampled complex filtered noise and, due to pulse shaping and matched filtering q, combined with full symbol rate sample timing,

It is assumed that is due to the channel impulse response c only.

Equalizer and Carrier Phase / Frequency Loop

의 위상 회전 버전과 비교하여, 필터 탭 가중치의 업데이트된 세트를 계산하는데 사용되는 에러 신호를 생성한다. LMS 알고리즘은 아래아 같이 동작할 수도 있다.The digital equalizer and carrier phase / frequency loop of FIG. 24A are described in more detail with reference to FIG. 25. Signal x [k] is applied to adaptive digital equalizer 250, which may include a linear digital filter that is used to compensate for the slope caused by channel impulse response c. Tap weighted adjustments may be obtained using one or more known methods, including LMS algorithms. Equalizer determines two-dimensional ("2-D") slicer for its output y [k]

Compared to the phase rotation version of, produces an error signal that is used to calculate an updated set of filter tap weights. The LMS algorithm may work as follows.

x [k] represents N long equalizer input vectors,

y [k] denotes the equalizer output vector g ^H [k] x [k].

Where g ^H [k] is N long equalizer tap weighting vectors, and superscript H is conjugate transposition (hermitite)

Where μ is a small step size parameter and superscript * denotes a conjugate complex number.

To eliminate the effects of passband cable slope, after conversion, the LMS equalizer tap is close to the inverse of the channel impulse response c.

및 z[k]의 실수부 및 허수부를 독립적으로 나눈다. 위상 에러 검출 모듈 (258)은 z[k] 및

을 수시ㄴ하여 위상 에러 신호

를 형성한다. 로우 패스 ("LP")필터 (256)은 루프가 위상 및 주파수 오프셋 모두를 정정하게 하는 적분 비례 필터일 수 있다. 로우 패스 필터 (256)의 출력은 θ0 및 f0 모두를 정정하는 복소 위상/주파수 정정 요소

를 출력하는 복소 이산 전압 제어된 발진기 ("VCO") (254)에 공급된다. 또한, VCO (254)은 슬라이스 출력

을 "비보정"하는 출력

을 제공하여, 균등화기 탭 업데이트를 위한 에러 신호를 도출하는데 사용될 수 있다. 이는 통상적으로 균등화기가 x[k] 상에서 동작하기 때문에 요구된다. 또한, 도 24a를 참조하면, 균등화기 출력 z[k]은 검출된 실수 및 허수 레벨을 비트 그룹으로 변환하는 심볼 역정합기에 공급된다. 그 후, 디코더는 비터비 (Viterbi) 디코딩, 바이트 역인터리빙, 및/또는 리드-솔로몬 디코딩을 실행하여, 수신된 비트 에러를 정정하며 결과적인 데이터를 MII 인터페이스에 전달한다. 2-D slicer 252 outputs an estimate of d [k] originally transmitted.

And the real part and the imaginary part of z [k] are divided independently. Phase error detection module 258 is z [k] and

The phase error signal

To form. Low pass (“LP”) filter 256 may be an integral proportional filter that allows the loop to correct both phase and frequency offset. The output of low pass filter 256 is a complex phase / frequency correction element that corrects both θ0 and f0

Is supplied to a complex discrete voltage controlled oscillator (" VCO ") 254 that outputs the " In addition, the VCO 254 outputs slices

To "uncalibrate"

Can be used to derive the error signal for the equalizer tap update. This is typically required because the equalizer operates on x [k]. Also, referring to FIG. 24A, the equalizer output z [k] is supplied to a symbol de-matcher for converting the detected real and imaginary levels into bit groups. The decoder then performs Viterbi decoding, byte deinterleaving, and / or Reed-Solomon decoding to correct the received bit error and deliver the resulting data to the MII interface.

Influence of Cable Length

를 유발하며, 그 결과, 전송 신뢰도를 상당히 악화시킬 수 있다. 디지털 신호에서, 균등화기는 이러한 장애를 제거하기 위해 수신기에서 사용될 수 있다. 도 27a 및 도 27b는 균등화기 입력의 전력 스펙트럼 밀도 (PSD) 및 수렴된 균등화기 탭의 응답을 각각 도시한다. 특히, 도 27a는 15.98MHz (통과대역 및 상대적인 기적대역 주파수 모두 도시)의 반송파 주파수로, RG-6 케이블의 2000ft를 통한 전송 이후 균등화기 입력의 PSD 를 도시하며, 도 27b는 수렴된 디지털 균등화기 탭의 진폭 응답을 도시한다. The received video signal may experience attenuation as a function of frequency contributes to certain characteristics of the cable. For the purposes of this discussion, examples of coaxial cables are described. The stringency of the attenuation, referred to as the slope, typically depends on the cable type and length. 26A and 26B show the attenuation as a function of frequency for cable types RG6 and RG59 of various lengths. The slope is equivalent to multipath distortion, with the additional path and the main path having an extremely small delay spread. As the slope increases, the number of many multipath components, and their respective gains, also increases. Multipath distortion can occur in the received signal

And, as a result, can significantly degrade transmission reliability. In digital signals, the equalizer can be used at the receiver to eliminate this obstacle. 27A and 27B show the power spectral density (PSD) of the equalizer input and the response of the converged equalizer taps, respectively. In particular, FIG. 27A shows the PSD of the equalizer input after transmission over 2000 ft of the RG-6 cable, with a carrier frequency of 15.98 MHz (both the passband and relative miraculous band frequencies), and FIG. 27B shows the converged digital equalizer. The amplitude response of the tap is shown.

를 제거할 수 있으며, 전송된 데이터의 신뢰할 수 있는 디코딩을 가능케 한다. 케이블의 길이가 증가함에 따라, 모니터측에서 디지털 통과대역 신호는 디지털 균등화기, 또는 연관된 리스-솔로몬 코딩 및 격자 코딩과 같은 디지털 데이터에 대한 공지된 포워드 에러 방지 방법을 사용하여 신뢰성 있게 수신될 수 있다. 그러나, 케이블 기울기는 기저대역 아날로그 CVBS 신호의 고 주파수에 불리한 영향을 미쳐, 모니터측에서 보이는 컬러의 선명함 및 휘도를 감소시킨다. 따라서, 특정 실시예는 기저대역에서의 케이블 기울기를 보상하기 위해 모니터측에서의 CVBS 신호에 인가될 수 있는, 아날로그 균등화기와 같은 적용가능한 필터를 제공한다. 특정 실시예는 기저대역에서의 기울기 총량을 측정하기 위해 통과대역 디지털 균등화기를 사용하여, 수신된 CVBS 신호에 적용될 기저대역 아날로그 필터들 중 적당한 하나의 세트를 선택한다. Certain embodiments of the present invention may include a digital equalizer capable of undoing the slope introduced by the cable to provide a

Can be eliminated, enabling reliable decoding of transmitted data. As the length of the cable increases, the digital passband signal at the monitor side can be reliably received using a digital equalizer or known forward error prevention methods for digital data such as associated Lis-Solomon coding and lattice coding. . However, the cable slope adversely affects the high frequency of the baseband analog CVBS signal, reducing the sharpness and brightness of the color seen on the monitor side. Thus, certain embodiments provide an applicable filter, such as an analog equalizer, that can be applied to the CVBS signal at the monitor side to compensate for cable slope at baseband. A particular embodiment uses a passband digital equalizer to measure the total amount of slope in the baseband to select a suitable set of baseband analog filters to be applied to the received CVBS signal.

Passband  Efficient Measurement of Tilt

In measuring the slope in the signal band, a frequency band may be selected at a portion that will be approximately linear when the slope of the input signal in PSD is quantified in dB. As a result, the frequencies of -2.67 MHz to 2.67 MHz of the baseband digital equalizer inputs corresponding to 13.31 MHz and 18.65 MHz of the passband input signal provide a suitable range. As shown in FIG. 26A, the slopes of 13.31 MHz and 18.65 MHz are approximately 3.7 dB for 2000 feet of RG-6. To measure the slope of dB from the converged digital equalizer filter tap, the following calculation can be performed.

G [k] is the DFT of the time domain converged equalizer filter tap, and k ₁ and k ₂ correspond to specific frequency bins of the DFT. Since the digital equalizer of FIG. 25 may be performed with time domain convolution, an FFT (or it is also possible to multiply or add N complex numbers to both points) is typically required for the purpose of measuring the slope for a given k1 and k2. In other words,

이며, n=0, 1...N-1은 N 개의 시간 도메인 군등화기 탭이다 (시간 인덱스에 대한 의존성은 생략됨). 1/N 스칼라는 이 계산에서 불필요하다. 유사한 계산이 G(k₂)에 대해 수행될 수도 있다. 그러나, 계산은 주파수 빈을 주의깊게 선택함으로써 상당히 감소될 수 있다. 2.67MHz의 주파수에 대응하여 K₁=N/4 로 함으로써, 수학식 (2)의 복소 지수는 상당히 간단화된다.here,

N = 0, 1 ... N-1 are N time domain equalizer taps (dependence on time index is omitted). 1 / N scalar is unnecessary in this calculation. Similar calculations may be performed for G (k ₂ ). However, the calculation can be significantly reduced by carefully choosing the frequency bins. By making K ₁ = N / 4 corresponding to the frequency of 2.67 MHz, the complex exponent of Equation (2) is considerably simplified.

The real and imaginary parts of the filter frequency response can be calculated using summation.

Finally, the power at this frequency bin is:

By allowing k ₁ = N / 4, the power calculation is greatly simplified. Similarly, by corresponding to the frequency of -2.67 MHz, by setting K ₁ = 3N / 4, the complex index will again be considerably simplified.

Real and imaginary parts:

는 상기와 같이 계산된다. 도 2b에서, 수렴된 필터 탭의 (dB 의) 진폭 응답의 상승 기울기는 탭 잡음 및 64-QAM 신호 대한 보통의 SNR까지 선형이다. 또한, 이러한 방법에서 계산시,

이며, 이는 이 3.7dB의 대역에 걸처 실제 기울기에 상당히 근접한다.Equals power

Is calculated as above. In FIG. 2B, the rising slope of the amplitude response (of dB) of the converged filter tap is linear up to the tap noise and the normal SNR for the 64-QAM signal. In addition, when calculating in this way,

This is quite close to the actual slope over this 3.7dB band.

Baseband CVBS  For tilt correction Passband  Use of slope estimates

After estimating the passband slope for the digital video signal, a suitable baseband analog filter may be selected from one of M different filters. The estimated passband slope of the digital video signal band will represent the severity of the slope of the baseband CVBS signal, which can be approximately corrected with an analog filter. In FIG. 28A, the slopes of the digital video signal bands of 13.31 MHz and 18.65 MHz are shown for RG-6, RG-11, RG-59, and RG-174 and similar lengths with these cables. FIG. 28A shows the slope versus loss at 3.58 MHz in a passband digital video signal for RG-6, RG-11, RG-59, and RG-174 cable types. 28B shows the loss at 6 MHz. The losses at 3.58 MHz and 6 MHz are approximately the same for the four cable types for a given slope. 29A shows the slope versus loss at 3.58 MHz in a passband digital video signal for RG-6, RG-11, RG-59, and RG-174 cable types. 29B shows the loss at 6 MHz. It will be observed that the losses at 3.58 MHz and 6 MHz are approximately the same for the four cable types for a given slope.

Since the estimated passband slope is the only available information about the frequency response of the cable, the ideal scenario is that the frequency response of the cable in the baseband (CVBS signal band) is passband in a known manner regardless of cable type or length. Is related to the slope of the digital signal. 28B, 29A, and 29B confirm this situation of frequency response at DC, 3.58 MHz, and 6 MHz. For example, at a 1.5 dB slope of the passband digital video signal, the loss at DC, the loss at color carrier (3.58 MHz), and the loss at 6 MHz are approximately 0.68 dB and 4.1 dB for all four cables, respectively. , And 5.3 dB. As a result, regardless of whether 1.5 dB of passband slope is derived from 275 ft. Of RG-174, 750 ft. Of RG-59, 825 ft. Of RG-9, or 1825 ft. Of RG-11, the same analog filter The baseband slope of the CVBS signal may be undone.

An example of the algorithm used to select the appropriate analog filter from the M filter sets is shown below.

α ₀ = 1, the other value of α _n is less than 1, and the bit-shifted addition is chosen to be sufficient to calculate R _n . Thus, the monitor-side QAM demodulator of FIG. 24A is typically modified such that the digital equalizer of the passband QAM demodulator provides a signal that selects one of the M analog CVBS filter responses. FIG. 24B shows a modified portion of the monitor side QAM demodulator with analog filter selection output from a digital equalizer operating according to the algorithm described above. 30 shows a full monitor side modem with a digital equalizer in QAM demodulator 304 that provides filter selection signal 305 to CVBS analog equalizer 302.

An example of an analog active filter suitable for equalizing the baseband CVBS signal is shown in FIG. 31. In this example, M = 3, so there are four possible filtering choices. The preferred filter response is selected by closing one of the M + 1 switches of the switch module 310 that in turn grounds each RC pair connected to the switch module 310. Possible filter responses are shown in FIG. 32.

Those skilled in the art will appreciate that the present invention applies to digital communication systems and forward error correction methods that use other passband modulation. One skilled in the art can also use two or more points of the FFT of the passband digital equalizer tap weighting vector g [n] to select an analog filter for the CVBS signal, with values of G ₁ [k] and G ₂ [k]. It will be appreciated that as part of this equalization process, other types of digital equalizer designs for passband signals can be used, including frequency domain equalizers that have already been calculated. Also, such as RLS, known equalizer tap weighted phase imaging methods other than LMS may be used.

In certain embodiments, the CVBS analog filter with the selectable response may take a form other than the form described above. The equalizer for the CVBS signal may also take the form of a digital filter where the CVBS is sampled and digitized prior to equalization. In this case, the tap weight of the digital filter is selected from a predetermined set of M weighted vectors according to the same algorithm described for selecting one of the M analog filter responses.

In digital communication systems Framing

Typically, digital data has some kind of frame structure, so that the data is organized into groups of bits or bytes of uniform size. Systems using block based forward error correction (FEC) have frames organized around the error correction code word size. Also, if the system uses interleaving to combat impulse noise, the frame structure will be arranged with interleaver parameters in mind. If the system uses data randomization to obtain a flat spectrum, the pseudo-random sequence used may be synchronized to the frame structure restarting at the beginning of each frame.

For RF digital communication systems, a receiver typically must first obtain carrier and symbol block synchronization and equalization. The receiver can then recover the matured data. However, in order to understand the applied data stream, the receiver must synchronize to the frame structure. That is, the receiver must know where the error correction code word starts and ends. In addition, the receiver module, such as a deinterleaver, is synchronized to match the interleaver operation of the transmitter so that the resulting deinterleaved bits or bytes must be correctly ordered, and the derandomizer is a pseudo-random used at the transmitter to smooth the spectrum. Matches the starting point of the sequence.

Conventional systems often provide receiver frame synchronization by appending symbols of a known pattern of fixed length to the beginning and end of the frame. This same pattern repeats every frame and consists of a two-level (ie binary) pseudo-random sequence with preferred auto-correlation characteristics. This means that while the auto-correlation of the sequence yields a large value at zero offset, the correlation value (side-lobe) is quite small if the offset is not zero. Also, the correlation for this frame sync sequence with random symbols will yield a small value. Thus, if the receiver performs correlation of the applied symbols with the stored frame sync pattern version, it will yield a large value only at the exact beginning of each frame. The receiver can then easily determine the starting point of each frame.

Example of frame structure

Referring to FIG. 9, the ATSC Digital Television (DTV) Terrestrial Transmission Standard, adopted in 1996, provides a system in which data is transmitted in frames. Frame 90 includes 313 segments, each segment including 832 symbols, for a total of 260,416 symbols per frame. The first four symbols of each segment are segment sync symbols 92 that contain the sequence [+5, -5, -5, +5]. The initial segment of each frame is frame sync segment 94 with 312 data segments 96 and 98. Referring to FIG. 10, frame sync segment 94 comprises segment sync 100, 511 symbol pseudo-random noise (PN511) sequence 101, 63 symbol pseudo-random noise (PN63) sequence 102, Two PN63 sequences 203, and a third PN63 sequence 104. This is followed by 24 mode symbols 105 indicating that the mode is 8 VSBs. Pre-code symbol 107 and reserved symbol 106 complete frame sync segment 94. Segment sync 100 and PN511 101 symbols are known to the receiver and may be used to obtain frame synchronization through a correlation method. All the aforementioned symbols come from the set {+5, -5}. From the last 12 symbol sets {-7 -5 -3 -1 +1 +3 +5 +7} of this segment, it is a copy of the last 12 symbols of the preceding data field. These are referred to as precode symbols (not discussed herein).

Referring also to FIG. 11, for each of the subsequent 312 segments of the field, referred to as data segments, the 828 symbols 32 following the four segment sync symbols 30 take two bits at a time and take them out. After trellis encoding to 3 bits, each unit of 3 bits is matched to 8 level symbols from the set {-7 -5 -3 -1 +1 +3 +5 +7} 207 bytes (1656 bits) of the Reed-Solomon (RS) code-word are generated.

Another example of framing in a digital communication system is shown in an ISDB-T system. Unlike single-carrier ATSC systems, ISDB-T is orthogonal coded frequency division multiplex {COFDM; coded orthogonal frequency division multiplexing}. For example, Mode 1 for ISDB-T uses 1,404 carriers. The frame consists of 204 COFDM symbols, where each COFDM symbol can be thought of as a combination of 1,404 independent QAM symbols, one for each carrier. As a result, the frame is composed of 204 x 1040 = 286,416 QAM symbols. Of these, 254,592 are data, and 31,824 contain pilot information (which can be used for frame synchronization) and mode information distributed through the frame in a known pattern.

A simplified view of this frame configuration is shown in FIG. 12. It can be seen that the pilot and mode information are distributed over the frame in a known pattern. This system has a mode that uses three different QAM constellations -QPSK, 16 QAM, and 64 QAM. The system also supports five different lattice coding rates {1/2, 2/3, 3/4, 5/6, 7/8} based on a single punctured mother code. This known technique is quite economical to build a single Viterbi decoder in a receiver that can be easily adjusted to decode all five of certain codes.

Prior to grating coding at the transmitter, data is formed of 204 byte (1,632 bits) long RS blocks. The number of COFDM symbols per frame is always constant and the number of RS blocks per frame changes depending on the mode selected, but most importantly, the number is always an integer. This facilitates RS block synchronization at the receiver when frame synchronization is established and the lattice code rate is known. For this to be true, the number of data bits per frame before lattice coding should be evenly distributed to 1.632 for all modes.

Table 1 shows the number of data bits per frame for all modes (combination of QAM constellation and lattice code rate). In all cases, the number of data bits per frame is evenly divided into 1632 (data bits mean bits before lattice coding).

Data bits per frame for ISDB-T mode Data Bits / Frames (Before Grid Coding) Bit / frame after grid coding 1/2 2/3 3/4 5/6 7/8 QPSK 254592 336456 381888 424320 445536 509184 16 QAM 509184 678912 763776 848640 891372 1318368 64 QAM 763776 1318368 1145664 1272960 1336608 1527552

Certain embodiments of the present invention provide a frame structure for a modulation system used in a digital communication system. In particular, signal systems and methods are provided that can be used in security systems, including those described above. The convolution byte interleaver interleaves a frame of data, the interleaver is synchronized to a frame structure, and the randomizer is configured to generate a randomized data frame from the interleaved data frame. In one example, the perforated lattice code modulator is operated at a selectable code rate that generates a lattice coded data frame from a randomized data frame. The QAM matcher matches a group of bits of the lattice coded data frame to a modulation symbol to provide a matched frame, and the synchronizer adds a synchronization packet to the matched frame. The perforated grating code modulator can be bypassed as desired to obtain an optimized net bit rate under various white noise conditions, thereby allowing for performance optimization of the system.

In certain embodiments, a new frame structure is provided in a single carrier communication system. Auto-correlation of a known pattern of fixed length symbols at the beginning and end of a frame at an offset of zero yields a large value, and if the offset is not zero, the correlation value (side-lobe) Is quite small. However, the correlation for this frame sync sequence with random symbols yields a small value. Thus, to obtain a large value at the exact beginning of each frame that allows the receiver to determine the start point of each frame, the receiver may perform correlation of the symbols applied to the stored version of the frame sync pattern. The communication system may operate in any of a plurality of modes, and may use various combinations of symbol constellations, lattice codes, and interleaved patterns. The receiver must recognize and understand the mode in order to successfully recover the transmitted data. For this purpose, additional mode symbols may be added to the frame sync pattern. These mode symbols can be reliably received using the correlation method because the mode symbols are transmitted repeatedly every frame. Reliable reception can be made more robust by encoding the mode symbols using block codes.

The frame structure according to certain aspects of the present invention uses a perforated lattice coding and QAM constellation combination similar to those used in ISDB-T. The number of symbols per frame may be a variable integer that depends on the mode, and the number of RS packets per frame is a constant that is independent of the mode. This structure simplifies the design of the receiver to handle blocks such as derandomizers and deinterleavers since the number of RS packets per frame is always fixed. In conventional systems such as ISDB-T, the number of symbols per frame is constant, and the number of RS packets per frame is a variable integer depending on the mode. The frame will be described with reference to the example shown in FIG. 13 of a transmitter structure constructed in accordance with certain aspects of the present invention.

RS encoder 1300 obtains an externally generated frame sync signal that indicates the beginning of each group of byte data 1301 and 315 Reed-Solomon packets 1322. As shown in FIG. 14, each packet 140 includes 207 bytes, 20 of which are parity bytes 142. These 315 Reed-Solomon packets form a forward error correction ("FEC") data frame 1322 comprising 65,205 bytes.

The convolution byte interleaver 1302 is as follows. 15 illustrates an operating mode of the interleaver 1302 to combat impulse noise affecting the transmitted signal. Parameter B of paths 156, 158 is set to 207, and parameter M of paths 152, 154, 156, and 158 is set to one. The frame sync signal 1303 forces the input and output commutators 150 and 151 to the top position 1500, resulting in synchronizing interleaving with the frame structure. Input and output commutators 150 and 151 move down to one location 1502 when bytes are applied to the interleaver and different bytes exit the interleaver. When commutators 150 and 151 reach the bottom 1508, the commutators 150 and 151 move back to the top 1500. Each of the B parallel paths 1506 and 1508 includes shift registers 156 and 158 having the length shown in FIG. 15 (path 1506 has a length of (B-2) M and path 1508 is (B-1) has a length of M).

The randomizer 1306 is on 65,205 x 8 = 521,640 bits of an FEC data frame 1324 with a 209-1 length PN sequence shortened by resetting the PN (pseudo-random noise) sequence generator at every frame sync time. By performing an exclusive OR operation, a randomized FEC data frame 1328 is generated by operating on 65,205 x 8 = 521,640 bits of the FEC data frame 1324.

An example of a selectable code rate punctured trellis coded modulation ("PTCM") module 1308 is shown in detail in Figure 16. PTCM 1308 uses methods known to those skilled in the art. Starting with a state 1/2 rate coder and starting puncturing to obtain any one of five different code rates In addition, in certain embodiments, PTCM 1308 may be fully bypassed (code rate = 1). This allows for a selectable tradeoff between white noise performance and net bit rate for the system Similar lattice coding techniques are used in ISDB-T and DVB-T systems PTCM is used for every bit provided at input 1328. Generate two bits 1332 at the output per step, but some of the output bits 1332 are discarded according to the selected code rate and corresponding puncturing pattern. The bits are taken into groups of 2, 4, or 6 from 1332 and matched to QPSK, 16 QAM, or 64 QAM symbols, respectively, An example of such a match is provided in FIG.

Module 1312 adds a frame-sync / mode symbol packet (all symbols are QPSK) at the beginning of each FEC data frame 1334. Referring to FIG. 18, the first portion 180 of this packet includes 127 symbols and includes an identifying binary PN sequence for both real and imaginary parts for the symbol. Other PN sequence lengths are possible, with the real and imaginary parts having opposite signs. The second portion 182 of this packet contains data indicative of the transmission mode—selected QAM constellation diagram and selected grid code rate. This mode data can be encoded using a block error correction code for added reliability at the receiver. Methods that can be used include BCH coding and other block codes. In one example, six possible grating code rates are possible including bypass. In addition, three constellations are possible, yielding 18 modes. Thus, five bits are needed to represent each of the possible mode selections. Five bits may be encoded into a 16 bit code word using an extended BCH code. Since each QPSK symbol includes two bits, eight mode symbols may be required.

FIG. 19 illustrates a frame structure 1336 (see FIG. 13) provided for passband modulation (“PB Mod”). Payload 190 includes 315 RS packets (521,640 bits). The number of QAM symbols to which 315 RS packets are matched may change according to mode selection. The PB Mod module 1314 then modulates the baseband QAM symbols into the passband using any suitable method known to those skilled in the art.

The frame structure according to certain aspects of the present invention advantageously overcomes certain drawbacks and failures of conventional frames. In particular, the frame structure is for all modes;

Constant integer RS packets per frame, regardless of mode, and

The number of QAM symbols per frame is a variable integer for all modes.

-Integer number of perforation pattern periods per frame for all modes

To provide. Since an FEC data frame must have exactly I x 207 data bytes, providing an integer QAM symbol per frame is a minor achievement, where I is an integer selected to have a fixed integer RS packet per frame. Thus, the number of data bits per frame before lattice coding must be an integer, as well as equally distributed to 207 x 8 = 1656 for all modes. In addition, the number of grid coder output bits per QAM symbol is 2, 4, and 6 bits for QPSK, 16 QAM, and 64 QAM, respectively (see Table 2 showing code rate = 1 for grid code bypass). . Lattice coding also adds bits. The number of data bits per symbol before lattice coding is shown in Table 2, where each entry is calculated as follows.

(Right column entry-code rate)

Data bits per symbol (input bits to grid coder per matched QAM symbol) Constellation Grid code rate 1/2 2/3 3/4 5/6 7/8 One QPSK 1.00 4/3 1.50 5/3 1.75 2.00 16 QAM 2.00 8/3 3.00 13/3 3.50 4.00 64 QAM 3.00 4.00 4.50 5.00 5.25 6.00

The number of data bits per symbol is a fractional request, and the RS packet size per frame and the number of RS packets are correctly selected. With 315 packets per frame and RS packet size of 207, a true number of symbols per frame are obtained. As shown in Table 3, each entry is calculated as follows.

(Data bits per frame / data bits per symbol = 521640 / entry from Table 2)

Symbol per frame Constellation Grid code rate 1/2 2/3 3/4 5/6 7/8 One QPSK 521640 391230 347760 312984 298080 260820 16 QAM 260820 195615 173880 156492 149040 130410 64 QAM 173880 130410 115920 104328 99360 86940

This frame provides the added benefit of having an integral number of perforation pattern (pp / frame) periods for every mode. To decode the perforated lattice coded data, the receiver's decoder must know how the perforation pattern is aligned with the data. The bit-wise puncturing pattern applied to the output of the mother code is shown in the second column of the table of FIG. The number of ones in each puncture pattern is the puncture pattern length. In the proposed system, the puncturing pattern is always lined up with the beginning of the FEC data frame. This allows the use of frame synchronization at the receiver, which properly aligns the de-puncturer at the Viterbi decoder with the bit stream. Preferred alignment is shown in Table 4, which shows the true numbers of pp / frame for all modes. The per-symbol perforation pattern ("pp / symbol") entry is calculated as follows.

(# of grid coder output per symbol / pp)

pp / frame entry;

Symbols per Frame from Table 3 / (pp / symbol)

Perforation Pattern Per Frame code
Rate pp
Length QPSK
(2 beats / symbol) 16QAM
(4 beats / symbol) pp / symbol pp / frame pp / symbol pp / frame pp / symbol pp / frame 1/2 2 One 521640 2 521640 3 521640 2/3 3 2/3 260820 4/3 260820 2 260820 3/4 4 1/2 173880 One 173880 3/2 173880 5/6 5 1/3 134328 2/3 134328 One 134328 7/8 8 1/4 74520 1/2 74520 3/4 74520 One NA NA NA NA NA NA NA

Other combinations of number of packets per frame and RS packet size can be used to achieve the same desired result. The numbers provided herein are disclosed for illustrative purposes only.

및

(2019)를 형성한다. 프레임 동기 모듈 (2000)은 이진 프레임-동기 PN 시퀀스의 저장된 복사본으로, 실수부 및 허수부 모두에 대해 별개로, 인가하는 슬라이스된 QAM 심볼 (2019)상에서 연속적인 상호-상관 (cross-correlation)동작을 수행한다. 저장된 복사본의 각각의 수는 -1 또는 +1의 값을 가진다. 이 동작은,As shown in FIG. 20, certain embodiments of the present invention provide a receiver configured to handle a frame constructed in accordance with certain aspects of the present invention. Module 2000 receives the data transmitted in the passband signal and converts it to a baseband QAM symbol. The operations performed by module 2000 may typically include symbol clock synchronization, equalization (to eliminate mid-symbol interference), and carrier recovery using a sub-module. Thus, module 2000 may include an equalizer that outputs the recovered baseband QAM symbol 2001. The baseband QAM signals 2001 are provided to a two-level slicer 2018 that slices in both real and imaginary directions, thereby providing a sequence to the frame-sync module 2020.

And

2020 is formed. The frame synchronization module 2000 is a stored copy of the binary frame-synchronous PN sequence, which is a continuous cross-correlation operation on the applying sliced QAM symbol 2019, separately for both real and imaginary parts. Do this. Each number of stored copies has a value of -1 or +1. This behavior is

Lt; / RTI >

Where s is a copy stored in 127 long frame-sync PN sequences. The maximum amplitude of b _R or b _I indicates the beginning of the FEC data frame.

When the frame sync start position is placed, the position of the code words including the mode bit (the degree of constellation and the lattice code rate) is known. The code word can be reliably decoded, for example, by a BCH decoder or by selecting a code word that correlates the received code word with all possible code words and yields the best result. Since this information is transmitted repeatedly, additional reliability can be obtained by requiring that the same result occur multiple times before the result is accepted.

This derived frame-sync signal 2021 is used to indicate which symbols will be removed in the " removal frame-sync / mode symbol " module 2004 before the symbols are fed to the soft de-matcher 2006. do. In one example, 127 frame-sync symbols and eight mode symbols are removed from the stream to ensure that only the symbols corresponding to the RS packets are passed to the soft de-matcher 2006. Soft inverse-matcher (2006) calculates the soft bit matrix using algorithms known in the art, including, for example, the algorithm described by Akay and Tosato. For correct operation, the soft de-matcher 2006 needs to know which puncturing pattern (which grid code) was used at the transmitter and the alignment of the pattern with the received bits. This information 2021 is provided by the frame-synchronization module 2020 which provides a repeating frame synchronization signal in which the puncturing patterns are aligned and decodes the mode information, regardless of the current mode. These soft bit matrices are fed to a Viterbi decoder 2008 that operates in a manner known in the art to reach an estimate of the bits that have been input to the transmitter's PTCM encoder.

The de-randomizer 2013, the byte de-interleaver 2014, and the RS decoder 2016 are all synchronized by the frame-synchronization signal 2021, and de-randomize, de-interleave, and byte data respectively. Decode to obtain the data originally applied to the RS encoder of the transmitter.

Carrier Phase Offset Correlation

Certain embodiments of the present invention employ carrier phase offset correction systems and methods. In a particular embodiment, the receiver receives an equalized signal representing a quadrature amplitude modulated signal to obtain a phase offset corrector that derives a phase-corrected signal from the equalized signal, slices the equalized signal to obtain a real sequence and an imaginary sequence. A two-level slicer, a real time sequence and an imaginary sequence and a frame synchronizer for performing correlation with the corresponding portion of the stored frame-synchronous pseudo-random sequence, and a phase correction signal provided to the phase offset corrector by the frame synchronizer. Based on the maximum real and imaginary values of the phase correction signal correlation. The frame synchronizer performs continuous cross-correlation for the applied sliced quadrature amplitude modulated symbols. Continuous cross-correlation is performed separately for real and imaginary sequences with stored copies of binary frame-synchronous pseudo-random noise sequences.

From baseband To the passband  Modulation

Certain wireless digital communication systems, including broadcast, wireless LAN, and wide area mobile systems, use some form of QAM. QAM also enables two dual-side-band suppressed-carrier modulated waves to occupy the same channel bandwidth, with each wave using a square-carrier multiplexing modulated by an independent message. Used in all digital cable television standards. As noted above, FIG. 23 shows a simple QAM modulator that may serve as the PB mode 1314 in the example of FIG. 13. The isolated transmitted QAM pulse is

Lt; / RTI >

Where d _{R , m} and d _{I , m} are determined by two independent streams and represent the real and imaginary parts of a complex QAM symbol (see eg FIG. 17), respectively, m = 1 ... M Is the two-dimensional QAM constellation of the radix, where M is the modulated carrier frequency and q (t) is the root raised cosine pulse function.

에 의해 주어지며, 여기서 *는 컨벌루션 (convolution)을 나타내며, c(t)는 채널 임펄스 응답이고, υ(t)는 부가적인 화이트 가우시안 잡음이다. 그 결과, A successive series of transmitted QAM pulses s (t) are propagated through the noise multipath channel at a rate of F _s = 1 / T _S. As a result, the received signal at the input to the QAM receiver

Is denoted by convolution, c (t) is the channel impulse response, and υ (t) is the additional white Gaussian noise. As a result,

Where d [n] is a complex transmitted symbol, and f ₀ and θ ₀ are the frequency of the demodulator local oscillator from the receiver passband to the baseband with respect to the transmitter such that f _L0 = f _c -f ₀ , respectively. Phase offset.

In the passband  Demodulator to baseband

FIG. 24A shows QAM demodulator 222 (FIG. 22) in more detail from monitor side notification band to baseband. Signal r (t) may be received from a coaxial cable, for example, sampled at a rate higher than the symbol rate (see 240) resulting in a sampled signal r (nT _samp ). After sampling below:

After demodulation, resampling at symbol rate 1 / Ts, and matched filtering,

는 샘플링된 복소 필터링된 잡음이다. 완전한 심볼 레이트 샘플 타이밍과 결합된, 펄스 형상화 및 정합된 필터링 q로 인해, 임의의

는 채널 임펄스 응답 c 만으로 인한 것임을 가정한다. 복조 이후, 완전한 균등화를 가정하는 경우, 균등화기 출력에서의 근처 기저대역 복소 시퀀스 z[k]는,here

Is sampled complex filtered noise. Pulse shaping and matched filtering q, combined with full symbol rate sample timing,

It is assumed that is due to the channel impulse response c only. After demodulation, assuming full equalization, the near baseband complex sequence z [k] at the equalizer output is

Is obtained by.

As a result, the recovered near baseband sequence represents a transmitted constellation diagram with rotation at phase offset θ ₀ , frequency f ₀ . In order to reliably recover the transmitted d _R and d _I , for example, the use of an equalizer, a two-dimensional slicer, combined with a phase and frequency offset recovery loop, must remove the frequency offset f ₀ , which rotates the constellation. The remaining static phase offset θ _0, which may have the constellation diagram in the static rotational position, must be removed.

In order to understand phase / frequency recovery, the QAM constellation diagram at baseband must be understood. In the simple example of FIG. 33A, a QAM modulation known as QPSK consists of four symbols. In the example shown, the real and imaginary parts of d [k] can each take two different values (such as ± 3). The effect of the phase offset θ ₀ on the recovered d [k] is shown in FIG. 33B, indicating rotation in the complex plane. Effect of f ₀ It is understood by recognizing the call ships with time in a counter-clockwise direction or counterclockwise direction by rotation depends on the sign of f _0.

Equalizer and Carrier Phase / Frequency Loop

의 위상 회전된 버전과 비교하여, 업데이트된 필터 탭 가중치 세트를 계산하는데 사용되는 에러 신호를 생성한다. 이 필터는 채널 임펄스 응답 c에 의해 유발된

를 제거한다.In FIG. 34, signal x [k] 340 is received by digital equalizer and carrier phase / frequency loop 248 (see, eg, FIG. 24A) dp. The equalizer 341 component typically includes a linear digital filter, and using known or proprietary methods such as least mean square (LMS), the equalizer 341 outputs the output y [k] of the equalizer 341. Slicer decision

Compare the phase rotated version of to generate an error signal that is used to calculate the updated set of filter tap weights. This filter is caused by the channel impulse response c

Remove it.

를 출력한다. z[k] 및

모두 위상 에러 검출 모듈 (346)으로 인가하여

에 의해 주어진 위상 에러 신호를 형성한다. 적분-비례 ("IP") 필터 (345)는 도 35의 필터 또는 당업자에 공지된 임의의 균등물을 포함할 수 있다. IP 필터 (345)는 루프가 이상 및 주파수 오프셋을 보정하도록 승인한다. IP 필터 (345)의 출력은 θ₀ 및 f₀ 모두를 보정하는 복수 위상/주파수 보정 성분

을 출력하는 복소 전압 제어된 발진기 (VCO) (344)에 공급된다. 또한, VCO (344)는 슬라이스 출력

을 "보정하지 않는"

를 출력하여,

는 균등화기 탭 업데이트에 대한 에러 신호를 도출하는데 사용될 수 있다.The 2-D slicer 342 slices the real and imaginary parts of z [k] independently, which is an estimate of d [k] originally transmitted.

. z [k] and

Are all applied to the phase error detection module 346

Form the phase error signal given by Integral-proportional (“IP”) filter 345 may include the filter of FIG. 35 or any equivalent known to those skilled in the art. IP filter 345 authorizes the loop to correct for anomalies and frequency offsets. The output of the IP filter 345 is a multiple phase / frequency correction component that corrects both θ ₀ and f _0.

Is supplied to a complex voltage controlled oscillator (VCO) 344 that outputs the output. In addition, the VCO 344 outputs slices

"Do not calibrate"

By printing

Can be used to derive the error signal for the equalizer tap update.

This approach appears because the equalizer operates on x [k] which contains both θ ₀ and f ₀ .

로서 수신된다. 따라서, 위상 복구 루프는 신호를 회전시켜 θ₀를 보상함으로써,

은 a로 라인업된다. 그러나, 2-D 슬라이서 (162)의 결정은, b가

에 더 근접하기 때문에 보정 심볼은 b라는 것이 될 것이다. 이는 위상 복구 루프가, 성좌도를 회전시켜

가 b로 라인업되는 방법으로 수렴하게 할 수 있다. 이 경우 최종 위상은 있어야 할 곳으로부터 오프셋 -π/2 이다. In certain embodiments, efficiency may be obtained by implementing discrete form of VCO 344 due to the delay of one integrator that is fed to a complex exponential look-up table (LUT). However, the final correction for θ ₀ may have an ambiguous π / 2 meaning that the recovered phase is accurate (offset = 0), or may have π / 2, offset π, or offset 3π / 4. These results are shown in FIGS. 35 and 37, where the actual transmitted symbols are shown in FIG. 36, and possible recovered symbols with respective offsets are shown in FIGS. 37A and 37D. Typically, since 2-D slicer 342 performs the closest neighbor operation, the receiver may not know which of the four possible symbols were actually transmitted. 38 shows at the equalizer input that transmitted symbol a is θ ₀ as shown.

Is received as. Thus, the phase recovery loop rotates the signal to compensate for θ ₀ ,

Is lineup with a. However, the determination of the 2-D slicer 162 is that b is

Since it is closer to, the correction symbol will be b. This is because the phase recovery loop rotates the constellation

Can be converged by b lineup to b. In this case the final phase is the offset -π / 2 from where it should be.

및 를 형성하는 2-D 레벨 슬라이서 (342)에 의해 실수 방향 및 허수 방향 모두로 슬라이스된다. 프레임 동기 모듈 (2020)은 이진 프레임-동기 PN 시퀀스의 저장된 복사본으로, 실수부 및 허수부 모두에 대해 개별적으로, 인가되는 슬라이스된 QAM 심볼상에서 연속적인 상호-상관 동작을 수행한다. 저장된 복사본의 각각의 멤버는 -1 또는 +1의 값을 가진다. 이 동작은 구체화될 수도 있어 결과는,Certain embodiments of the present invention provide a method for eliminating and / or minimizing this problem in a grid coded system, including the group of perforated grid codes used in some of the presently disclosed embodiments. As described above, the output of the equalizer is a sequence supplied to the frame-synchronization module 2020 (see FIG. 20).

And Sliced in both real and imaginary directions by a 2-D level slicer 342 which forms. The frame synchronization module 2020 is a stored copy of the binary frame-sync PN sequence that performs successive cross-correlation operations on the applied sliced QAM symbols separately for both real and imaginary parts. Each member of the saved copy has a value of -1 or +1. This behavior may be specified so that the result is

to be.

Where s is a stored copy of 127 long frame-sync PN sequences. The maximum amplitude of b _R or b _I indicates the beginning of the FEC data frame.

Sign of max b _R Sign of max b _I Required phase correction + + 0 - + + π / 2 - - + π + - -π / 2

가 주어지는 경우, 이 동작은, For the frame sync symbol, the real part and the imaginary part have the same sign, and the constellation diagram of the real part and the imaginary part is shown in FIG. As a result, the sign of the maximum amplitude b _R or b _I are all positive for zero rotation. For rotation of π, both b _R and b _I are negative, for rotation of π / 2, the maximum amplitude b _R is positive and the maximum amplitude b _I is negative. This is summarized in Table 5 above. As a result, each sign of the maximum amplitudes b _R and b _I combine to represent the quadrant of the complex plane where the final phase offset converges. This allows for additional phase correction to be applied to the signal as shown in FIG. The sign of at most b _R and b _I is transmitted from the correlation based frame-sync module to the phase offset corrector. The operation of the phase offset corrector module, in which the LUT operation 404 is shown as an example, is shown in FIG. 40.

If is given, this behavior is

1. About φ = + θ

2. About φ = + π / 2

3. About φ = -π / 2

Can be executed.

40 is a block diagram representation of a phase offset corrector that derives a phase-corrected signal by indexing the LUT with the sign of the largest real and imaginary values of the correlation in accordance with certain aspects of the present invention.

multiple- mode QAM  Constellation degree detection

를 최소화하기 위해 표준 CMA 로 재시작한다. 균등화기 출력은 정확하게 스케일링될 수 있으며, 이 단계 이후 감소된 성좌도 반송파 복구 (RCCR; reduced constellation carrier recovery) 및 반송파 복구에 대한 결정이 수행되어, 결합된 균등화기 반송파 주파수/위상 루프에 의해 반송파 주파수 및 위상을 복구한다. 미공지 QAM 성좌도를 결정하는 또 다른 방법에서, 균등화기는 초기에, 수정된 CMA 를 사용하여

를 최소화한다. 균등화기 출력이 프로세스 주 이 포인트에서 정확히 스케일링되지 않을 수도 있지만, 균등화기 반송파 주파수/위상 루프는 성좌도를 알지 못하면서 RCCR을 사용하여 반송파 주파수 및 위상을 복구한다. 복구된 위상은 잡음일 수 있다. 수신기는 QAM 성좌도가 전송되고 있는 신호 프레임에 내장된 정보를 판독할 수도 있다. 그 후 균등화기 동작은 미공지 성좌도에 기반하여 표준 CMA로 재시작하며, 반송파 복구에 대한 RCCR 및 결정이 후속한다. Certain embodiments provide a system and method for determining an unknown QAM constellation from a set of possible received QAM constellations. One method is the power of a signal after intermediate-symbol-interference (ISI) is minimized with a modified constant modulus algorithm (CMA) equalizer, but before the carrier frequency and phase are fully recovered. Use a histogram. The unknown constellations are then determined from the histogram. The equalization processor is based on the unknown constellation

Restart to standard CMA to minimize this. The equalizer output can be scaled accurately, and after this step, decisions about reduced constellation carrier recovery (RCCR) and carrier recovery are performed, so that the carrier frequency and phase are combined by the combined equalizer carrier frequency / phase loop. Restore the phase. In another method of determining the unknown QAM constellation, the equalizer initially uses a modified CMA

Minimize. Although the equalizer output may not be scaled exactly at this point in the process note, the equalizer carrier frequency / phase loop uses the RCCR to recover carrier frequency and phase without knowing the constellation. The recovered phase may be noise. The receiver may read information embedded in the signal frame in which the QAM constellation is being transmitted. The equalizer operation then restarts with a standard CMA based on the unknown constellation diagram, followed by RCCR and decision on carrier recovery.

Certain embodiments of the invention use perforated lattice coding and QAM constellation combinations similar to those used in the ISDB-T described above. As used herein, constellation degrees are understood to mean matching in the complex plane of possible symbols for the modulation method. The number of symbols per frame is a variable constant that depends on the mode, and the number of RS packets per frame is a constant that is independent of the mode. This configuration has been described in more detail above and simplifies the design of the receiver.

Referring back to FIG. 20, the frame synchronization module 2020 is a stored copy of the applying frame-sync PN sequence, which is consecutively on the applying sliced QAM symbol 1219, separately for both real and imaginary parts. -Perform the correlation operation. Each member of the saved copy has a value of -1 or +1. This operation given by Equation 10 is repeated here.

Where s is a copy stored in 127 long frame-sync PN sequences. The maximum amplitude of b _R or b _I indicates the beginning of the FEC data frame.

As will be explained in more detail below, there is an ambiguity of π / 2 in the recovered carrier phase. This results in any additional recovered phase offset of 0, ± π / 2, or π. For the frame sync symbol, the real part and the imaginary part have the same sign, and the constellation diagram transmitted for the frame sync symbol is shown in FIG. As a result, for the zero phase offset, the signs of the maximum amplitudes b _R and b _I are both positive. As summarized in Table 404 of FIG. 40, the -π / 2 offset yields a negative maximum amplitude b _R and a positive maximum amplitude b _I , and for an offset of π, both b _R and b _I will be negative and π / For an offset of 2 the maximum amplitude b _R will be positive and the maximum amplitude b _I will be negative. As a result, each sign of the maximum amplitudes b _R and b _I combine to represent the quadrant of the complex plane where the final phase offset converges. This allows for additional phase correction to be applied to the signal of the phase offset correlation module 2002. The sign of at most b _R and b _I may be sent from the correlation based frame-synchronization module 2020 to the phase offset corrector 2002.

가 주어지는 경우, 동작 (402)는, Referring to FIG. 40, the operation of certain aspects of the phase offset corrector 2002 of the example of FIG. 20 may be better understood. LUT 400 generates an output based on the signs of maximum amplitudes b _R and b _I (see component 404 in FIG. 40).

If is given, the operation 402,

1) About φ = + π

2) for φ = + π / 2

3) About φ = -π / 2

Can be executed.

When the frame sync start position is placed and the m [pi] / 2 phase offset is corrected, the position of the code word including the mode bits (constellation degree and lattice code rate) is known. The code word can be reliably decoded, for example, by a BCH decoder or by selecting a code word that correlates the received code word with all possible code words and yields the best result. Since this information is transmitted repeatedly, additional reliability can be obtained by requiring that the same result occur multiple times before the result is accepted.

41 shows an example of a process that may be performed by the frame sync module 2020. In response to the frame-synchronization signal 2021, at step 4100, the received constellation code words are cross-correlated with all valid code words. Cross-correlation yields a value that can be used to select the best match. In one example, the valid code word that produces the maximum correlation value is selected at step 4102. This selected code word can be used to identify the current constellation diagram. In step 4104, the identity of the current constellation diagram is compared with the identity of a previously identified constellation diagram that has been recorded or stored. In step 4104, if the current constellation and the previously identified constellation are the same constellations, then the reliability counter may be incremented. In step 4104, if it is determined that the previously identified constellation is different from the current constellation, in step 4107 the current constellation is recorded as the previously identified constellation, in step 4107 the reliability counter is decremented, and in step 4109 another sync frame is removed. Waiting. Subsequent to step 4106, which is an increase in the reliability counter, if the reliability counter is tested in step 4108 and determined to exceed a predetermined or configured threshold, a determination of the signal constellation may be made in step 4110. Repetition of this process may be performed until the reliability counter exceeds a predetermined or configured threshold.

Equalizer and Carrier Phase / Frequency Loop

를 제거한다. 균등화기 (420)의 출력 y[k]는 임의의 나머지 반송파 위상 및 주파수 오프셋을 감소시키기 위해 (421)에서 위상 회전된다. 위상 회전된 출력 z[k]은 적분-비례 (IP) 필터 (426)에 공급되는 위상 에러 값 e_θ[k]을 계산하는 위상 에러 검출기 모듈 (427) 및 슬라이서에 의해 처리된다. IP 필터 (426) 출력은 반송파 위상 및 주파수 오프셋을 보정하기 위해 루프에서 사용된 복소 지수 값을 계산하는 복수 지수 LUT (424) 및 적분기로 공급된다. 슬라이서 및 위상 에러 검출 모듈 (427)은 그 위상이

로 곱셈에 의해 "보정되지 않으며" 에러 계산기 모듈 (422)에서 사용되는 가장 근접한 이웃 2-차원 슬라이스된 심볼 결정을 출력한다. 에러 계산기 모듈 (422)은 이 입력뿐만 아니라 x[k]를 사용하여 에러 신호 e[k]를 계산한다. 도시된 바와 같이, 에러 계산기 모듈 (422) 및 슬라이스 및 위상 에러 검출기 모듈 (427)의 내부 동작은 단계 제어기 (423)에 의해 결정되는 현재 동작 단계 (1, 2, 또는 3)에 의존한다. Referring to FIG. 42, a particular aspect of the equalizer and carrier phase / frequency loop 248 of FIG. 24A is described. Signal x [k] is applied to a carrier phase / frequency loop 248 and a digital equalizer that may include an equalizer 420 that includes a linear digital filter. The error calculation module 422 calculates an error signal e [k] that can be used to calculate the updated set of filter tap weights using any suitable method known to those skilled in the art. In one example, an LMS algorithm may be used. The filter is caused by the channel impulse response c

Remove it. Output y [k] of equalizer 420 is phase rotated at 421 to reduce any remaining carrier phase and frequency offset. The phase rotated output z [k] is processed by the phase error detector module 427 and slicer which calculates the phase error value e _θ [k] supplied to the integral-proportional (IP) filter 426. The IP filter 426 output is fed to a multi-exponential LUT 424 and integrator that computes the complex exponential values used in the loop to correct the carrier phase and frequency offset. Slicer and phase error detection module 427

Outputs the nearest neighbor two-dimensional sliced symbol decision used in error calculator module 422, which is not "corrected" by multiplication. The error calculator module 422 uses this input as well as x [k] to calculate the error signal e [k]. As shown, the internal operation of the error calculator module 422 and slice and phase error detector module 427 depends on the current operation step (1, 2, or 3) determined by the step controller 423.

In a particular embodiment, the LMS algorithm is used to calculate the equalizer filter tap weights and operates as follows:

x [k] represents N long equalizer input vectors, y [k] represents equalizer output vector y [k] = g ^H [k] x [k], where g ^H [k] is N Long equalizer tap weighting vectors, the superscript H being the conjugate transposition (hermitite). The updated e [k] is then calculated in the error calculator module 422 using, for example, the method described below.

Where μ is a small step size parameter and superscript * denotes a conjugate complex number.

The step controller 423 takes the carrier phase / frequency loop 428 and the equalizer through three operating steps, whereby switching from step 1 to step 2, step 3 results in a simple count of input data samples x [k]. It is executed based on the threshold. In addition, more complex step switching is possible based on an estimate of the error at the equalizer output. Three steps are summarized in Table 6.

Equalizer and Carrier Phase / Frequency Loop Steps step e _θ [k] calculation method e [k] calculation method Frequency / Phase Recovery Steps One CMA Always 0 Constellation diagram spin 2 CMA Based on small constellation diagram (RCCR) Phase / Frequency Recovers Gradually 3 DD Based on the entire constellation diagram Phase noise reduction

에 의해 주어진 심볼의 전력 z[k]이 임계값 ξ를 초과하는 경우, z[k]는 성좌도의 코너 심볼 중 하나이며, RCCR 은 도시된 제 2 스위치 (432)를

를 산출하는 상위 위치로 설정함으로써 인에이블된다. 또는,

인 경우, 제 2 스위치 (432)는 반송파 루프를 디스에이블하는 더 낮게 도시된 위치에 있다. 그 결과, 심볼들 중 서브세트만이 단계 2 동안 반송파 복수에 기여할 수 있다. 임계값 ξ성좌도 코너 주변 영역에 더 많은 심볼을 포함하기 위해 감소될 수 있지만, 결과적인 위상 보정 텀 (term) e_θ[k]은 더 잡음이 많을 것이다. 단계 3 동안, 스위치 (430)는

를 산출하는 최저로 도시된 위치에 있으며, 여기서

는 가장 근접한 이웃 2-차원 슬라이스된 심볼 결정

의 켤레 복소수이다. 균등화기 탭들이 수렴하도록 충분한 시간이 흘렀으며, 슬라이드된 심볼 결정이 신뢰할만 하도록 반송파 위상이 실질적으로 보정되었음을 가정한다. 특히, 관계

및

은 단일의 복소 평면 내에서 효율적으로 동작한다. 이는 전술한 바와 같이 복구된 반송파 위상에서 mπ/2의 모호함을 유발한다. Slicer and phase error detector module 427 is shown in more detail in FIG. 43. The switch 430 is set in accordance with one of three operational stages 434. During step 1, switch 430 is in the highest position, e _θ [k] = 0. This effectively turns off the carrier loop so that there is no carrier phase during this phase. During step 2, the switch 430 is in an intermediate position and the loop operates using a reduced constellation carrier recovery (RCCR) algorithm.

If the power z [k] of the symbol given by < RTI ID = 0.0 > is < / RTI >

It is enabled by setting to a higher position that calculates. or,

If, then the second switch 432 is in the lower shown position to disable the carrier loop. As a result, only a subset of the symbols may contribute to carrier plurality during step two. The threshold ξ constellation can also be reduced to include more symbols in the area around the corner, but the resulting phase correction term e _θ [k] will be more noisy. During step 3, the switch 430 is

Is the lowest shown location that yields

Determines the nearest neighbor two-dimensional sliced symbol

Is a complex number. Suppose sufficient time has passed for the equalizer taps to converge, and the carrier phase has been substantially corrected so that the slid symbology is reliable. In particular, the relationship

And

Operates efficiently within a single complex plane. This leads to ambiguity of mπ / 2 in the recovered carrier phase as described above.

)을 출력하는 LUT (444)로 공급되는 위상 에러 신호 θ[k]를 형성하기 위해 입력의 한 단계 지연된 (442) 버전에 모듈로 (modulo) 부가된 2π (440) (도 44 참조) 이다. LUT (444)는 슬라이서 출력

을 "보정하지 않는" 출력 (446) (

)을 제공하여, 출력 (446) (

)은 균등화기 탭 업데이트에 대한 에러 신호를 도출하는데 사용될 수 있다. 이는 균등화기가 θ₀ 및 f₀ 모두를 포함하는 x[k]사에서 동작하기 때문에 필요하다. An example of an IP filter (see FIG. 42) is shown in more detail in FIG. 35. IP filter 426 allows the loop to correct both phase and frequency offset. The output of the IP filter 426 is fed to the integrator complex index LUT module 424, which is described in more detail in FIG. The input of the integrator / LUT 424 is a phase correction component 445 that corrects for both θ ₀ and f ₀ (

Is modulo added to the one step delayed 442 version of the input to form a phase error signal θ [k] which is fed to LUT 444 (see FIG. 44). LUT 444 output slicer

"Do not calibrate" output (446) (

By providing the output 446 (

) Can be used to derive the error signal for the equalizer tap update. This is necessary because the equalizer operates at x [k], which includes both θ ₀ and f ₀ .

Error Calculation Module and Step Action Summary

The error calculator 422 can use a different method of calculating e [k] depending on the step. For steps 1 and 2, e [k] is typically CMA;

Calculated using a process based on. Where R is

Is a predetermined constant given by.

를 최소화하 것과 균등하다. Where E is the expected actuator and d [k] is the symbol (see FIG. 17). This e [k], which yields a tap update of Equation 11 above, is independent of the symbol determination and the phase of x [k], and depends only on the statistics of the equalizer output, the equalizer input, and the constellation plot. During steps 1, 2, and 3, the use of the CMA error to derive Equation 11 is true even if the constellation spins due to carrier frequency and phase offset.

Is equivalent to minimizing

를 최소화한다.

가 최소화된 이후, 단계 2 가 시작하여 루프는 RCCR에 대해 턴온되며, 도 43에 관하여 전술한 바와 같이, 반송파 위상/주파수 복구는 성좌도의 코너 심볼만을 사용하여 시작한다. 단계 2의 종료시, 반송파 위상 및 주파수가 성공적으로 복구되어, 도 43의 2-차원 슬라이서 (436)는 신뢰할만한 심볼 결정

을 출력하기 시작한다.As a result, during phase 1, the phase / frequency recovery loop is disabled, and the equalizer uses the CMA error function

Minimize.

After is minimized, step 2 begins and the loop is turned on for the RCCR, and as described above with respect to FIG. 43, carrier phase / frequency recovery begins using only the corner symbols of the constellation diagram. At the end of step 2, the carrier phase and frequency have been successfully recovered so that the two-dimensional slicer 436 of FIG.

Will start printing.

로서 계산될 수도 있다. 이러한 설명의 목적을 위해, R 이 이들 성좌도 각각에 대해 상이하기 때문에, 수신기는 도 17의 3개의 성좌도 중 어느 것이 전송되고 있는지를 결정한 것으로 가정한다. 또한, RCCR은 성좌도의 지식, 특히 성좌도의 코너 심볼의 전력 지식을 요구한다. Decision detected (DD) errors may be used in step 3. DD error

May be calculated as For the purpose of this description, it is assumed that since R is different for each of these constellations, the receiver has determined which of the three constellations in FIG. 17 is being transmitted. RCCR also requires knowledge of the constellation diagram, in particular the power knowledge of the corner symbols of the constellation diagram.

Unknown Constellation  Have CMA

In the example described herein, one of three different constellation diagrams may be transmitted, and the equalization and phase / frequency recovery described above require knowledge of the transmitted constellation diagrams. Constellation selection is encoded in the mode symbol, equalization and phase / frequency recovery precede frame synchronization (see FIG. 20), and this information can be directly decoded as described above (see, for example, FIGS. 28, 20 and 41). Can be. As a result, in certain embodiments, the degree of constellation is determined within the equalizer and the carrier recovery algorithm itself.

R (as provided in Equation 12) is also constellation dependent. With continued reference to certain embodiments and FIG. 17, the real and imaginary parts of symbols for 64-QAM are selected from the set ± {1,3,5,7} and the real and imaginary parts of symbols for 16-QAM. The part is selected from the set ± {2,6}, and the real part and the imaginary part of the symbol for QPSK are selected from the set ± 4. According to Equation 12, the value of R is

to be.

에 의해 스케일링된 도일한 값의 세트로 수렴되게 하며, 균등화기는 출력은 유사하게 스케일링된다.

는 그럼에도 불구하고 최소화된다. 성좌도가 미공지된 일 예에서, R은 58로 설정될 수 있으며, 전송된 성좌도에 관계없이,

는 단계 1 동안 최소화될 것이다. 전술한 예에 대해, 범위 32-58 내의 임의의 값 R이 사용될 수 있다. 그러나 최대값 (즉, 58)의 선택은 균등화기 출력에서 초대 밀도의 성좌도 (여기서는 64-QAM)의 압축을 방지하며, 균등화기 성능에 대한 부담을 감소시킨다. For any of the three constellation views of FIG. 17, the use of the scaled value αR for the CMA error calculation is determined by the equalizer filter tab.

Converges into a set of scaled values that are scaled by the equalizer, and the output is similarly scaled.

Is nevertheless minimized. In one example, where the constellation is unknown, R may be set to 58, regardless of the transmitted constellation,

Will be minimized during step 1. For the above example, any value R within the range 32-58 can be used. However, choosing the maximum value (ie 58) prevents compression of the super-density constellation (here 64-QAM) at the equalizer output and reduces the burden on equalizer performance.

The use of scaled CMA parameter R causes a scaling up of the equalized output by the converged filter taps, so that the statistics of the equalizer output are

의 제거 및 성좌도와 관계없음을 가정한다. 그 결과, QPSK 에 대해,

가 단계 1 동안 최소화된 이후, 균등화기 출력은 아래와 같이 스케일링될 것이다.Will be complete

It is assumed that there is no relation to the removal and constellation of. As a result, about QPSK,

After is minimized during step 1, the equalizer output will be scaled as follows.

를 제거하는 솔루션에 수렴함에 따라, R=58의 값으로 인해 출력은

에 의해 스케일링된다. 도 45b는 θ₀=f₀=0 인 경우에 대해, 16-QAM 를 갖는 시스템에 대한 균등화된 출력의 실수부를 나타낸다.

이 상대적으로 1에 근접하기 때문에, 균등화기 출력의 실수부는 약간 스케일링된 것으로만 보인다. 그 결과, 실제 스케일링은 균등화기 수렴 동안 명백하다. 45A shows the real part of the equalized output for a system with QPSK for the case of θ ₀ = f ₀ = 0. The equalizer

As we converge on a solution that eliminates, the value of R = 58

Is scaled by. 45B shows the real part of the equalized output for a system with 16-QAM for the case of θ ₀ = f ₀ = 0.

Since this is relatively close to 1, the real part of the equalizer output only appears to be slightly scaled. As a result, the actual scaling is evident during equalizer convergence.

How to protect the constellation

의 전력 히스토그램을 고려한다. 히스토그램은 균등화기가 R=58로 수렴된 이후의 전력을 나타낸다. 균등화기의 출력이 위상에 독립적이며, 각각의 성좌도에 대한 히스토그램은 실질적으로 상이하기 때문에, 전송된 성좌도는 균등화기 출력 전력 히스토그램으로부터 수신기에서 결정될 수 있다. In certain embodiments, prior to entering step 2, a histogram may be used to determine the degree of constellation. Even if the carrier phase and frequency have not yet been recovered, the constellation may be determined. Equalizer output shown in FIGS. 16A, 46B, and 46C for QPSK, 16-QAM, and 64-QAM constellation views, respectively

Consider the power histogram. The histogram shows the power after the equalizer has converged to R = 58. Since the output of the equalizer is phase independent and the histogram for each constellation is substantially different, the transmitted constellation can be determined at the receiver from the equalizer output power histogram.

이다. 16-QAM 성좌도에 해대, 균등화기 출력에 대한 확률 질량 함수는,Without additional or tap noise, the power of each equalizer output sample for the QPSK constellation plot

to be. Compared to the 16-QAM constellation plot, the probability mass function for the equalizer output is

Similarly, the probability mass function for the equalizer output is plotted against the 64-QAM constellation plot,

to be.

및 탭 업데이트 잡음으로 인해, 예를 들어, 실질적인 30dB의 SNR에 대해서도 이들 값 주변에 히스토그램의 일부 확산이 존재한다. 균등화기 출력상의 잡음을 심볼에 부가적이며 독립적으로 모델링시, 출력에는

가 없음을 가정하면,Additional noise on the input signal

And due to tap update noise, there is some spread of the histogram around these values, for example, even for an actual 30 dB SNR. When modeling the noise on the equalizer output in addition to the symbol and independently,

Assuming there is no

항과 관련된 변동은 심볼 전력이 증가함에 따라 증가한다. 히스토그램 형상에서, 이 현상은 증가하는 심볼 전력과 함께 증가하는 주어진 성좌도 전력 주변의 확산, 즉 변동으로 나타난다. 16-QAM 경우, 심볼 ±2.1±j2.1의 성좌도 전력에 대한 확산은 심볼 ±6.3±j6.3의 성좌도 전력에 대한 확산 미만이다.Subject to the conditions of the given symbol,

The variation associated with the term increases as the symbol power increases. In the histogram shape, this phenomenon is manifested by the spread, or fluctuations, around the power, given a constellation that increases with increasing symbol power. For 16-QAM, the spread for constellation power of the symbol ± 2.1 ± j2.1 is less than the spread for constellation power of the symbol ± 6.3 ± j6.3.

Certain other relationships can be observed from the histogram of the equalizer output power.

The region T1 of the QPSK histogram falls approximately between R2 and R3, which are respectively the second and third regions of the 16-QAM histogram. Thus, the area declaring which symbol power was transmitted does not overlap for the QPSK and 16-QAM constellation diagrams.

를 나타낸다. 그 결과, 영역 T₁에 대한

의 비교를 위해,

는 영역 T₁의 외부에 있을 가능성이 높다.A comparison of QPSK histograms to 64-QAM histograms is shown for 64-QAM.

Indicates. As a result, for the area T ₁

For comparison,

Is likely outside of the area T ₁ .

는 9/16의 확률로 세트 {2,18,26,34,58,98}로부터 값을 취한다. 그 결과, 기본적인 성좌도가 64-QAM 인 경우, 잡음을 무시하면,In the absence of noise in the 64-QAM example,

Takes a value from the set {2,18,26,34,58,98} with a probability of 9/16. As a result, if the basic constellation is 64-QAM, ignoring the noise,

는 OR 를 나타낸다. 따라서, 전송된 성좌도가 64-QAM 이며

가 영역 R₁, R₂, 및 R₃ 와 비교되는 경우,

는 이들 영역의 외부에 있을 가능성이 높다.here

Represents OR. Thus, the transmitted constellation is 64-QAM

Is compared with the regions R ₁ , R ₂ , and R ₃ ,

Is likely outside of these areas.

Certain embodiments use algorithms based on these observations.

이 영역 R₁, R₂, 및 R₃ 내에 있는 경우 증가하며, 그렇지 않은 경우 감소한다. The algorithm is initialized after the equalizer converges and, in the first part, increases the QPSK counter λ ₄ [k] if the equalizer output power is in the N equalizer output sample frequency regions T ₁ . If the equalizer output power is not in region T ₁ , the counter is decremented. Similarly, the 16-QAM counter λ ₁₆ [k] is

It increases if it is within the regions R ₁ , R ₂ , and R ₃ , and decreases if it is not.

는 QPSK 및 16-QAM 영역의 외부에 속할 것이다. 전송된 성좌도가 QPSK 또는 16-QAM 인 경우, 전송된 성좌도의 카운터는 상당히 클 것이다.After N equalizer output samples, the histogram is correctly characterized. If the basic constellation is 64-QAM, the QPSK and 16-QAM counters will be much smaller, and the power estimate

Will fall outside of the QPSK and 16-QAM regions. If the transmitted constellation is QPSK or 16-QAM, the counter of transmitted constellation will be quite large.

As a result,

The threshold M can be determined experimentally, but should be relatively small compared to N. The algorithm is extremely robust and reliably selects a low signal-to-noise ratio (SNR) when QPSK, 16-QAM, or 64-QAM is transmitted. If the constellation is determined reliably, R can be set to correct the equation (13) and step 1 reaches completion. The equalizer output will be scaled appropriately, and step 2 may begin with the knowledge of the threshold ξ required for RCCR.

임계값 ξ을 초과하는 경우, 단계 2 RCCR에 대한 키는 심볼들만의 고려사항이다. z[k]는 성좌도의 코너 심볼들 중 하나인 것으로 가정될 수도 있다. 균등하게,

는 코너 심볼을 나타낼 수 있다. 64-QAM 성좌도에 대해 도 48a에서 설명한 바와 같이 성좌도가 공지된 경우 , ξ에 대한 값을 선택하기가 상대적으로 용이하다. 도 48은 반송파 위성/주파수 복구 루프 모듈 입력 및 균등화기 출력에서의 3개의 성좌도를 모두 도시한다. 코너 포인트에 대해

일 수 있다. 예를 들어, 점선원 (484)에 의해 표시된 임계값

은 코너 포인트만이 선택됨을 보장한다. 유사하게, 원 (482)에 의한

및 원 (480)에 의한

은 16-QAM 및 QPSK 각각에 대해 상당한 마진으로 사용될 수 있다.Another method for determining constellation degrees before the equalizer enters stage 3 is described. In this way, step 1 is executed and allowed to complete with R = 58. As a result, as described, all constellations will be scaled at the equalizer output, resulting in y [k] as shown in the three constellations of FIG. 47 even though these constellations are likely to spin. As discussed with respect to FIG. 43, the power of the symbol is given by z [k]

If the threshold value ξ is exceeded, the key for step 2 RCCR is a symbol only consideration. z [k] may be assumed to be one of the corner symbols of the constellation diagram. Evenly,

May represent a corner symbol. When the constellation diagram is known as described in FIG. 48A for the 64-QAM constellation diagram, it is relatively easy to select a value for ξ. 48 shows all three constellation diagrams at the carrier satellite / frequency recovery loop module input and equalizer output. About corner point

Can be. For example, the threshold indicated by the dashed circle 484

Ensures that only corner points are selected. Similarly, by circle 482

And by circle 480

May be used with significant margin for each of 16-QAM and QPSK.

인 경우, 점선 외부에 속하는 QPSK 및 16-QAM 대한 코너 포인트만이 RCCR에 의해 사용될 것이다. 그러나 64-QAM 이 수신된 경우, 5개의 성좌도 포인트 (4개의 비-코너)가 원 외부에 속하며, RCCR에 의해 사용될 것이다. 복구된 위상이 덜 잡음이 있기 때문에 코너 성좌도 포인트들이 사용된 경우에만 RCCR은 통상적으로 최상으로 동작한다. 그러나 위상 잡음이 발생할지라도, 일부 부가적인 포인트가 사용되는 경우에도 RCCR은 성공적으로 위상을 복구한다. 따라서, 단계 2는

로 초기에 동작할 수 있어, 모든 3개의 성좌도에 대한 적합한 초기 반송파를 허용하며, 성좌도는 수신기에 미공지로 유지된다.FIG. 49 shows the overlapping notation of one quadrant of all three constellation diagrams.

If, only corner points for QPSK and 16-QAM belonging outside the dashed line will be used by the RCCR. However, if 64-QAM is received, five constellation points (four non-corners) fall outside the circle and will be used by the RCCR. Because the recovered phase is less noisy, RCCR typically works best only when corner constellation points are used. However, even if phase noise occurs, the RCCR successfully recovers the phase even if some additional points are used. Therefore, step 2

It can be operated initially, allowing a suitable initial carrier for all three constellations, which remain unknown to the receiver.

As described with respect to FIG. 20, the equalizer 2000 supplies the two-level slicer 2018 and the two-level slicer 2018 in turn to the frame synchronization module 2020. The frame synchronization module 2020 may perform a continuous cross-correlation operation on the sign of an applied sliced QAM symbol with a stored copy of the binary frame-sync PN sequence, as described by Equation 10. Successive cross-correlation operations can be performed separately for both real and imaginary parts. Each member of the stored copy has a value of -1 or +1. The maximum amplitude of b _R or b _I indicates the beginning of the FEC data frame. The only difference here is that for 64-QAM constellation diagrams, the 2-level slicer (2018) operates on signals with some additional phase noise. However, this additional phase noise has a detrimental effect on the highly robust two-level slicing and subsequent cross-correlation based frame synchronization even in the presence of phase noise. Decoding of constellation codewords is quite robust to phase noise as described above.

50 illustrates the operation of this another approach to determining the constellation diagram, which can be summarized as follows.

(1) The equalizer and phase / frequency loop enter step 2 after completing step 1 with R = 58.

(2) Instead of waiting for step 3, correlation-based frame-sync 2020 receives input data during step 2, retrieves frame sync, and also decodes the constellation codeword.

(3) The determined constellation degree information 2021 is sent back to the phase / frequency loop and equalizer 2000 which returns to step 2 using the R value appropriately responding to the determined constellation degree.

(4) Steps 1, 2, and 3 are completed as before.

The main difference between the systems shown in FIGS. 50 and 20 is an additional connection 5000 to frame equalizer 2020 to equalizer / carrier recovery 2000 that carries constellation information.

Coax  On a secure link SPOT  Modulation

Certain embodiments of the present invention include the embodiments described above, where the baseband video signal is combined with a digital representation and control signal of the baseband video signal to enable transmission over a single cable, such as a coax. Improve the performance of systems and devices that can. Referring to Figure 4, one embodiment of the present invention provides a secure link ("SLOC") system over coax. 5 shows one possible modulation method for a SLOC system. For example, HD camera 30 provides an IP output 41 that includes a compressed digital HD video 332 and an auxiliary camera signal that includes analog SD CVBS 330. The compressed HD video IP signal 332 is modulated into the passband 52 using a SLOC camera side modem 49 that includes a QAM modulator (see modulator 212 of modem 32 in FIG. 21). Modulator 212 provides a modulated signal that can be combined with baseband analog CVBS signal 330. The combined signal is typically transmitted downstream via coaxial cable 41 for a distance that can extend over 300 m. On the monitor side, the SLOC monitor side modem 45 separates the baseband CVBS signal 330 from the passband downstream IP signal 332. Separate CVBS signal 330 is fed to SD display 43 for live, delay-free viewing. The passband downstream IP signal 332 is demodulated with a QAM demodulator (see demodulator 222 of FIG. 22) which outputs the signal to a host network switch 44 or a processor / DVR (not shown in FIG. 4).

In one example, upstream communication is provided in accordance with IP protocol requirements. Upstream communication 334 may additionally be used to send audio and camera control signals 42 from the monitor side to camera 40. Typically the bit rate for the upstream signal and the corresponding required bandwidth will be much lower than required for the downstream passband signal. Monitor-side SLOC modem 45 includes a QAM modulator (see modulator 224 of FIG. 22) that modulates the IP signal into upstream passband 44. As shown in FIG. 5, upstream passband 54 and downstream passband 52 are located at different spectral positions. On the camera side, the SLOC modem 49 includes a QAM demodulator (see demodulator 214 of the modem of FIG. 21) that receives the upstream signal. This approach offers several advantages over conventional systems and methods, including increased operating range, ease of adoption using existing coax infrastructure, low-delay, and acquisition of real-time video. 21 and 22 illustrate the SLOC camera side modem 49 of FIG. 4 and the SLOC monitor side modem 45 of FIG. 4 and have been described in detail above.

51 shows a filtered tab 519 provided between coaxial cable segment 512 and coaxial cable segment 514 such that tab 513 and cable segment 512 and 514 connect the camera side equipment with the monitor side component. 4 illustrates a SLOC system based on the system described in FIG. Typically filtered tap 513 is used to extract at least a portion of baseband CVBS signal 5100 to camera-side SD display 5130. Display 5130 may be provided near camera 510 for testing, setup, and / or local monitoring. Typically, the filtered tap 513 includes a low pass filter that blocks unwanted signals such as modulated digital, IP and / or control signals that may interfere with the display function 5130. In addition, tab 513 may include a filter or switch that blocks transmission of signals between modem 511 and modem 515. For example, a test modem 5131 may be connected via a tab 513 to enable initial setup or high-frequency repair of the camera-side modem 511, which may cause the display-side modem 515 to interfere with signals and / or It may be separated to prevent deterioration. As shown in FIG. 5, the SLOC camera-side modem 511 typically outputs a low passband QAM signal based on the camera-generated portion of the signal 5102 in addition to the baseband CVBS signal 5100, and the SLOC monitor The side modem 515 outputs a high passband QAM signal based on the control signal of the signal 5170. One or more filters may be provided by tap 513 to prevent unwanted interference seen on SD display 5130 and / or 516 and to block IP and control signals. Some displays and monitors lack the filtering necessary to block the molten frequency signal (for baseband CVBS signal 5100) in the passband signal.

51B shows that the cable 514 between the camera side and the monitor side is temporarily disconnected at the camera side, and the SD display device or monitor 5130 is directly connected to the SLOC camera side modem 511 via the cable segment 519. 3 illustrates a SLOC system based on the system described with reference to FIG. 3. The test modem 5191 may be optionally connected for test / setup purposes. The SD display device 5130 displays the baseband CVBS signal and provides the ability to monitor video from the camera 510 near the physical location of the camera 510, and reconfiguration of the connection facilitates setup and troubleshooting. It may be desirable. In FIG. 51B, low passband QAM signal 5102 can cause undesirable visual interference on SD display 5130 lacking high frequency filtering.

In the example shown in FIGS. 51A and 51, partial or complete disconnection of the signal between modem 511 and modem 515 may occur. Partial disconnection of the signal leaves the QAM signal transmission path completely. However, the reconfiguration of some connections causes separation of the QAM signal between the camera side modem 511 and the monitor side SLOC modem 515. Certain embodiments of the present invention provide a mechanism where the camera side modem 511 stops passband QAM transmission and outputs only a CVBS signal when the connection between modem 511 and modem 515 is interrupted. Temporary replacement of the test modem 5151 for the display-side modem may include disconnection between the modem 511 and the modem 515, establishment of a connection between the modem 511 and the modem 5121, and the modem 511 and the modem. And typically includes a disconnection between 5111 and a re-establishment of the connection between modem 511 and modem 515. Disconnection of the QAM signal may be detected using various functional components of the modem 511. Thus, the operation of the SLOC system is described in more detail below.

SLOC  For the system QAM  Modulation structure

As noted above, FIG. 19 illustrates a frame structure 1336 provided to a passband modulation (PB mode) module 1314 (see FIG. 13). The lattice coding of FIG. 16 adds bits, and the number of data bits per matched QAM symbol prior to lattice coding is shown in Table 2. The number of QAM symbols to which 315 RS packets (521640 bits) of FIG. 14 match is changed according to mode selection. As shown in Table 3, with 315 packets per frame and RS packet size of 207, symbols of true number per frame are obtained. The PB mode module 1314 modulates the baseband QAM symbols into a passband using any bonded method known to those skilled in the art (see, e.g., the foregoing with reference to FIG. 24).

및

(2019)를 형성한다.As described above, with respect to FIG. 20, the QAM demodulator of FIGS. 21 and 22 is further described. Module 2000 receives the data transmitted in the passband signal and converts it to a baseband QAM symbol. The operations performed by module 2000 may typically include symbol clock synchronization, equalization (to eliminate mid-symbol interference), and carrier recovery using a sub-module. Thus, module 2000 may include an equalizer that outputs the recovered baseband QAM symbol 2001. The baseband QAM signals 2001 are provided to a two-level slicer 2018 that slices in both real and imaginary directions, thereby providing a sequence to the frame-sync module 2020.

And

2020 is formed.

The frame synchronization module 2020 is a stored copy of the binary frame-sync PN sequence that performs successive cross-correlation operations on the applied sliced QAM symbols separately for both real and imaginary parts. Each member of the saved copy has a value of -1 or +1. This operation is given by Equation 1, where it is regenerated.

{Equation 10}

Where s is a copy stored in 127 long frame-sync PN sequences. The maximum amplitude of b _R or b _I indicates the beginning of the FEC data frame. When this FEC data frame start point is detected in the stream, a frame sync pulse or other sync signal is communicated to one or more of the receiver modules.

52A and 52B illustrate components of a process that can reliably generate frame sync pulses when a noise signal is received. 52A is part of a processor to determine frame length. The frame length can be changed depending on the selected transmission mode (Table 3). The process commencing at step 5200 is repeatedly executed upon receipt of a symbol, and the symbol counter records the number of symbols between executions that result in a value exceeding a predetermined threshold. In step 5201, cross correlation is performed for each arriving symbol, and the symbol counter is incremented until the predetermined threshold is determined to be exceeded in step 5202. The symbol counter is incremented (5203) for each symbol until the threshold is exceeded. If the threshold is exceeded in step 5202, the symbol counter is cleared (5204), cross-correlation 5205, increment of symbol counter 5207, and reception of a new symbol 5208 are thresholded in step 5206. This is repeated until it is determined to be exceeded. The intermediate symbol counter is written at step 5208 and the symbol counter is reset at step 5209. The steps of cross-correlation 5210, incrementing 5212 of symbol counters, and receiving 5213 of new symbols are repeated in step 5211 until it is determined that the threshold is exceeded. If in step 5214 the symbol counter is equal to the intermediate symbol counter recorded in step 5208, then in step 5215 the frame length is returned as the value of the symbol counter. In the example described, the frame length may be determined after two successive constant counters. However, the same counters required in succession may be selected as desired.

52B shows one process in which the frame sync module 2020 generates frame timed frame sync pulses even when the received signal is quite noisy. In addition, the process provides for the acquisition of a new frame sync position in the event of a temporary interruption of the signal, or after a transmitter transmission mode change causes a corresponding change in frame_size. The free running counter counts received symbols using modulo frame_size operations, where the frame_size was determined by the steps described in connection with FIG. 52A. In the result of Equation 10, if cross-correlation exceeds a selected threshold, the symbol counter value will always have the same value. If the value is constant, the confidence counter is incremented by a maximum of the selected maximum value, e.

As a result, upon receipt of the symbol at step 5250, cross correlation is performed at step 5251, and if the result at step 5252 exceeds the threshold, the current maximum is set to the threshold and the maximum point is the symbol at step 5253. It is set to the current value of the counter. In the illustrated example, the reliability counter is set to 4 or more (5254), and if the current symbol counter indicates a frame synchronization point (5255), a frame synchronization signal is output in step 5256. Next, at step 5257 the symbol counter is incremented using modulo 4 addition here. The next symbol waits at step 5277 unless the symbol counter is determined to be zero at step 5270. If the symbol counter is zero, then at step 5271 the current maximum is reset. In step 5272, if the current maximum point is equal to the frame synchronization point, the reliability counter is incremented in step 5273, the next symbol waits in step 5277, otherwise the reliability counter is decremented in step 5274. In the presently described example, if it is determined in step 5275 that the reliability has dropped below 2, then in step 5276 the frame synchronization point is set to the current maximum point. In any case, the next symbol is waited at step 5277.

In summary, according to the described process, it is determined that frame synchronization has been obtained reliably if the reliability counter exceeds a predetermined value—eg, four. Thereafter, the frame synchronization module may provide a frame synchronization pulse at a time that is cleared and purified. If the reliability counter exceeds 4, the frame sync pulse will be output at the correct time-typically at the time corresponding to the start of the frame, although noise often causes Equation 10 to produce a low value.

If the transmission mode is changed, the reliability counter will eventually return to zero. This can be used to trigger a recalculation of the frame length of determining the new frame length (eg, using the process of FIG. 52A). As described below with respect to carrier recovery, there may be an ambiguity of π / 2 in the recovered carrier phase that may cause binary additional recovered phase offset 0, ± π / 2, or π. For the frame sync symbol, the real part and the imaginary part have the same sign, and the transmitted constellation diagram is shown in FIG.

As a result, for the zero phase offset, the signs of the maximum amplitudes b _R and b _I are both positive. As summarized in Table 5, the -π / 2 offset yields a negative maximum amplitude b _R and a positive maximum amplitude b _I , and for an offset of π, both b _R and b _I will be negative and an offset of π / 2 For a maximum amplitude b _R will be positive and a maximum amplitude b _I will be negative. As a result, each sign of the maximum amplitudes b _R and b _I combine to represent the quadrant of the complex plane where the final phase offset converges. This allows for additional phase correction to be applied to the signal of the phase offset correlation module 2002. The sign of at most b _R and b _I may be sent from the correlation based frame-synchronization module 2020 to the phase offset corrector 2002.

가 주어지는 경우, 동작 (402)는, Referring to FIG. 40, the operation of certain aspects of the phase offset corrector 2002 of the example of FIG. 20 may be better understood. LUT 400 generates an output based on the signs of maximum amplitudes b _R and b _I (see Table 5).

If is given, the operation 402,

1) About φ = + π

2) for φ = + π / 2

3) About φ = -π / 2

Can be executed.

When the frame sync start position is placed and the m [pi] / 2 phase offset is corrected, the position of the code word including the mode bits (constellation degree and lattice code rate) is known. The code word can be reliably decoded, for example, by a BCH decoder or by selecting a code word that correlates the received code word with all possible code words and yields the best result. Since this information is transmitted repeatedly, additional reliability can be obtained by requiring that the same result occur multiple times before the result is accepted. 41 shows an example of a process that may be performed by the frame sync module 2020.

Continuing with the system of FIG. 20, the frame-sync signal 2021 may be used to indicate which symbol will be removed in module 2004 before the symbol is supplied to the soft de-matcher 2006. In one example, 127 frame-sync symbols and eight mode symbols are removed from the stream to ensure that only the symbols corresponding to the RS packets are passed to the soft de-matcher 2006. Soft inverse-matcher (2006) calculates the soft bit matrix using algorithms known in the art, including, for example, the algorithm described by Akay and Tosato. For correct operation, the soft de-matcher 2006 must know which puncture pattern (which grid code) was used at the transmitter and the alignment of the pattern with the received bits. The information 2021 is provided by the frame-sync module 2020 which provides a repeating frame sync signal in which the puncturing patterns are aligned and decodes the mode information, regardless of the current mode. These soft bit matrices are fed to a Viterbi decoder 2008 that operates in a manner known in the art to reach an estimate of the bits that have been input to the transmitter's PTCM encoder. The de-randomizer 2013, the byte de-interleaver 2014, and the RS decoder 2016 are all synchronized by the frame-synchronization signal 2021, and de-randomize, de-interleave, and byte data respectively. Decode to obtain the data originally applied to the RS encoder of the transmitter.

Phase switching

Certain embodiments use step switching based on the mean squared error at the output of the equalizer. An accurate estimate of the mean square error (MSE) of the equalizer output may be obtained from the series of errors e [k] calculated by the error calculation module 422 of FIG. For example, the estimate

It may be obtained by. Where β <1 is the forgetting factor. Other methods of averaging e [k] are known and can be used. Equation 18 can be compared with a predetermined threshold and used by the step control module 423 of FIG. 42 to switch the operation from step 1 to step 2 when MSE [k] falls below the threshold. Produces possible results. The resulting value may be compared with a second predetermined threshold to switch the operation from step 2 to step 3 when MSE [k] falls below the second threshold.

Disconnection and Reconnection Detection

Certain embodiments provide a system and method for detecting disconnection and reconnection events at the camera side of a communication link. Referring again to FIGS. 51A and 51B, partial or complete disconnection of the signal between modem 511 and modem 515 may occur in normal operation. The particular disconnection affects the QAM signal between the camera side modem 511 and the monitor side SLOC modem 515. In particular, the signal carrying the image captured by the HD camera 510 is encoded and / or modulated by the modem 511 for transmission via the cable 514 to the display side modem 515. Multiple methods of detecting disconnection and reconnection events associated with coax 514 may be performed by camera side SLOC modem 511. In response to a disconnect or reconnect event, modem 511 may stop, initiate, or resume downstream passband QAM transmission. In some embodiments, a "coax connected" signal sent from QAM demodulator 530 to a QAM modulator may be used to control the transmission for connection-related events.

Referring to FIG. 53, for example, the camera side QAM modulator 530 transmits the downstream passband signal 533 only when the coax connected signal 531 is claimed by the camera side QAM modulator 532. It may be configured to. Camera-side QAM demodulator 532 can determine the presence of an input signal 534 transmitted by a monitor-side QAM modulator (not shown) using the Viterbi method. Typically, when reception of the input signal 534 is reliably confirmed, when constellation degree identification is confirmed, and / or when confirmation of frame synchronization is obtained, the coax connected signal 531 is a camera-side QAM demodulator 532. Is claimed.

The method for detecting the presence of the input signal 534 includes a method based on an automatic gain control (AGC) loop. As found in a communication receiver that includes a QAM demodulator, AGC is used to control signal levels at various stages and points of the receiver. An example illustrating an AGC loop 540 added to the receiver front end of FIG. 24 is shown in FIG. 27. In the AGC loop 540, the amplitude of the complex signal is determined at 541 and subtracted at 541 from the predetermined reference level 543. The result is filtered by low pass filter (LPF) 544 to suppress noise and short term changes. LPF 544 provides an output that is supplied to an accumulator that includes adder 545 and delay component 546. The accumulator output is used as the gain control signal 547 which is fed back from the system input 549 to the gain block 548. In one example, gain control signal 547 is used as a multiplier or gain component to determine the gain provided by gain block 548 such that the gain provided by gain block 548 increases as gain control 547 increases. Increase within a predetermined limit. If the input 549 is disconnected (eg, the coax is disconnected), the output of the amplitude block 541 tends to be significantly lower. Typically, the coax connected signal 531 may only be asserted if the amplitude block output exceeds a predetermined threshold. Also, the gain control signal 547 is typically quite high when the input 549 is disconnected. As a result, the coax connected signal 531 may only be asserted if the gain control signal is below a predetermined threshold. The AGC loop 540 can be used to monitor the connection state of the input 549 only if the loop is found elsewhere in the QAM demodulator 532.

The method of detecting the presence of the input signal 534 is based on the carrier phase / frequency rope step and equalizer shown in FIG. 43 (see Equation 18). In particular, when the QAM modulator step controller 434 of the QAM demodulator 532 switches to step 2 based on the result of equation (18 at the beginning of step 1), the coax connected signal 531 may be asserted. . The transition from step 1 to step 2 only occurs when the coax is connected and the QAM demodulator 532 is actively receiving an upstream signal from the monitor side QAM modulator. Subsequent disconnection of the coax will cause a loss of signal, an increase in the MSE calculated by Equation 18, and will lead to a return to step 1. The coax connected signal 531 may be reset or reverse-claimed if the QAM retrograde 532 is in step 1. In some embodiments, camera-side QAM demodulator 532 may be required to have step 3 obtained prior to asserting coax connected signal 531.

Another method of detecting the presence of the input signal 534 is based on the demodulator frame synchronization reliability counter discussed with respect to FIG. 52B. In particular, the ocax connected signal 531 may be asserted by the camera side QAM demodulator 532 only if the reliability counter registers a value greater than a predetermined threshold. In one example, the threshold may be four. Thus, the coax connected signal 531 will only be asserted when the coax is connected and the monitor side modem is sending a SLOC frame to the camera. If the frame synchronization process continues to go free, even if no symbol is being received, the disconnection will return the reliability counter and eventually drop below 4, and the coax connected signal 531 will be de-asserted.

Another method of detecting the presence of the input signal 534 is based on a higher layer protocol. Referring again to FIG. 51A, the HD camera 30 and monitor side host system 38 may communicate using networking protocols. For the purposes of this discussion, ubiquitous internet protocol (IP) may be used as an example of a networking protocol. Some IP modes are essentially two-way, allowing data to be sent both upstream and downstream. If the cable is disconnected, the network controller or processor of the HD camera 30 and / or modem 32 recognizes that there are no return IP packets arriving from the monitor side, and the camera side SLOC modem ( 32) can be notified. In one example, such notification may include, for example, sending a special predetermined data packet from HD camera 30 to modem 32 via MII interface 536 shown in FIG. 53.

Additional Description of Certain Aspects of the Invention

The foregoing description of the invention is intended to be illustrative, and not intended to be limiting. For example, those skilled in the art will appreciate that the present invention may be practiced with various combinations of the foregoing functionality and capabilities, and may include fewer or more components than those described above. Certain additional aspects and features of the invention are given more below, and can be obtained using the functionality and components described in more detail above, as will be understood by those skilled in the art after being taught by the present disclosure.

Certain embodiments of the present invention provide a system and method associated with a camera. These embodiments include a processor that receives an image signal from an image sensor and generates a plurality of video signals representing the image signal, and an encoder that combines the baseband video signal with the digital video signal as an output signal for transmission over a cable. . In some of these embodiments, the video signal includes a baseband video signal and a digital video signal. Baseband and digital video signals may be substantially isochronous. In some of these embodiments, the camera is a closed circuit high definition television camera. In some of these embodiments, the baseband video signal comprises a standard analog video signal. In some of these embodiments, the digital video signal is modulated prior to combining with the baseband video signal. In some of these embodiments, the digital video signal comprises compressed digital video. In some of these embodiments, the digital video signal is a high definition digital video signal. In some of these embodiments, the frame rate of the digital video signal is less than the frame rate of the image signal. In some of these embodiments, the modulated digital signal is provided to a video recorder.

In some of these embodiments, the decoder is configured to demodulate the upstream signal received from the cable. In some of these embodiments, the demodulated upstream signal includes a control signal. In some of these embodiments, the control signal comprises a signal for controlling the position and orientation of the camera. In some of these embodiments, the control signal includes a signal that controls the generation of the baseband video signal and the digital video signal by the processor. In some of these embodiments, the control signal includes a signal that selects some of the image signal for encoding as a baseband video signal. In some of these embodiments, the control signal comprises a signal that selects some of the image signal for encoding as a digital video signal. In some of these embodiments, the demodulated upstream signal includes an audio signal that drives the audio output of the camera.

Certain embodiments of the present invention provide a system and method for transmitting video images. Some of these embodiments include frequency division multiplexing a video signal received from a high definition image device to obtain a modulated digital signal, and generating an output signal by combining the modulated digital signal with a baseband analog signal representing the video signal. And simultaneously transmitting the output signal to the display system and the digital video capture and / or storage device. In some of these embodiments, the display system displays an image derived from the baseband analog representation of the video signal. In some of these embodiments, the digital video store records a sequence of high quality frames extracted from the modulated digital signal using a digital video recorder.

Some of these embodiments include compressing the video signal. In some of these embodiments, frequency division multiplexing the digital video signal comprises compressing the video signal prior to modulation. In some of these embodiments, transmitting the output signal includes providing an output signal to the coaxial cable. Some of these embodiments include demodulating an input signal received from a coaxial cable to obtain a control signal. Some of these embodiments include generating a baseband analog signal by encoding a portion of the video signal in the composite video signal. Some of these embodiments include using a control signal to select, in the composite video signal, a portion of the video signal to be encoded. Some of these embodiments include controlling the position of the camera using control signals. In some of these embodiments, demodulating the input signal includes extracting an audio signal from the input signal.

Certain embodiments of the present invention provide a system and method for operating a camera. Some of these embodiments include a processor that receives image signals from an image sensor and generates a plurality of video signals, control logic configured to respond to control signals received by the camera, and modulate the digital video signal as a modulated signal. It includes a modulator. In some of these embodiments, the plurality of video signals includes a baseband video signal and a digital video signal. In some of these embodiments, each of the plurality of video signals represents at least a portion of the field of view of the camera. In some of these embodiments, the control signal controls the digital video signal and the baseband content. In some of these embodiments, the modulated signal and the baseband video signal are transmitted simultaneously by the camera.

In some of these embodiments, the baseband and digital video signals are substantially isochronous. Some of these embodiments include an encoder that combines a baseband video signal and a modulated signal as an output signal for transmission over a cable. In some of these embodiments, the control signal is received as a wireless signal. In some of these embodiments, the modulated signal is transmitted wirelessly. In some of these embodiments, the digital video signal is a high definition digital video signal. In some of these embodiments, the digital video signal comprises a compressed digital video signal. In some of these embodiments, the control signal shifts some of the field of view represented by one of the video signals.

Some of these embodiments provide equalizers for use as baseband analog signals and digital signals separated by frequency and carried by cable. Some of these embodiments include a digital equalizer that removes distortion from the digital signal received at the receiver. Some of these embodiments include an analog equalizer that compensates for the attenuation of the analog signal caused by the cable. In some of these embodiments, the analog equalizer applies one of the baseband analog filter sets to compensate for the attenuation. In some of these embodiments, the applied baseband analog filter is selected based on an estimate calculated by the digital equalizer of the difference in attenuation at different frequencies.

In some of these embodiments, digital and analog signals are transmitted between a transmitter and a receiver embedded in the camera, and the receiver provides an equalized signal representing the analog signal to the monitor. In some of these embodiments, the cable comprises a coaxial cable. In some of these embodiments, the distortion increases with the length of the cable. In some of these embodiments, the distortion includes multiple paths. In some of these embodiments, the estimate of the attenuation is calculated from a frequency band having a power spectral density whose slope is approximately linear. In some of these embodiments, the slope is calculated using a fast Fourier transform for the plurality of filter taps. In some of these embodiments, frequency bins within the frequency band are selected to allow calculation of the frequency response of the filter of the digital equalizer using summation.

Where G [k] is the DFT of the time domain converged equalizer filter tap, and k ₁ and k ₂ correspond to specific frequency bins of the DFT. In some of these embodiments, the digital signal includes a high quality representation of the video image captured by the camera, and the analog signal includes a standard representation of the video image.

Certain embodiments of the present invention provide a method of equalizing an analog signal in a cable that carries a digital signal separated from the analog signal by frequency. In some of these embodiments, the method is performed by a modem that receives analog and digital signals and outputs baseband video signals. Some of these embodiments include calculating the slope in the digital signal. In some of these embodiments, the slope characterizes attenuation as a function of frequency that may contribute to the cable. Some of these embodiments include equalizing the digital signal based on the calculated slope. Some of these embodiments include configuring an analog equalizer by using the calculated slope to select one of a set of baseband analog filters. Some of these embodiments equalize the analog signal using the selected baseband analog filter.

In some of these embodiments, the analog signal comprises a baseband video signal and the digital signal comprises a high quality version of the baseband video signal. In some of these embodiments, the cable comprises a coaxial cable, the slope of which varies with the length of the cable. In some of these embodiments, the slope is derived from multi-path distortion. In some of these embodiments, calculating the slope includes estimating the attenuation within a frequency band with a power spectral density of which the slope is approximately linear. In some of these embodiments, estimating the attenuation includes using a fast Fourier transform for the plurality of filter taps. In some of these embodiments, estimating the attenuation includes selecting a frequency bin within a frequency band. In some of these embodiments, the selected frequency bin optimizes the efficiency of the step of calculating the slope.

Certain embodiments of the present invention provide a digital communication system using a new frame structure. Some of these embodiments include a convolution byte interleaver that interleaves a frame of data, the interleaver being synchronized to the frame structure. Some of these embodiments include a randomizer configured to generate a randomized data frame from interleaved data frames. Some of these embodiments include perforated lattice code modulators operated at selectable code rates that generate lattice coded data frames from randomized data frames. Some of these embodiments include a QAM matcher that matches a group of bits of a lattice coded data frame to a modulation symbol to produce a matched frame. Some of these embodiments include a synchronizer that adds a synchronization packet to the matched frame.

In some of these embodiments, the perforated grating code modulator is bypassed to obtain an optimized net bit rate based on the measured white noise performance of the system. In some of these embodiments, the same synchronization packet is added to each subsequent sequence of matched frames. In some of these embodiments, the same synchronization packet is added to each matched frame. In some of these embodiments, the portion of the synchronization packet includes 127 symbols. In some of these embodiments, some of the synchronization packets include different binary sequences for the real and imaginary parts of the modulation symbols. In some of these embodiments, some of the synchronization packets contain the same binary sequence for both real and imaginary parts of the modulation symbol. In some of these embodiments, the synchronization packet includes data indicating the transmission mode for the matched frame. In some of these embodiments, the indication of the transmission mode includes the selected QAM constellation diagram and the selected grid code rate. In some of these embodiments, the system generates a constant true Reed-Solomon packet for each data frame regardless of the transmission mode. In some of these embodiments, the system generates a variable integer modulation symbol for each data frame regardless of the transmission mode. In some of these embodiments, the system generates a true number of puncturing pattern periods for each data frame regardless of the transmission mode.

Certain embodiments of the present invention provide a method of framing for a variable net bit rate digital communication system. Some of these embodiments include providing different sets of quadrature amplitude modulation (QAM) constellations. In some of these embodiments, generating the data packet frames using perforated grid code combinations, each combination corresponding to an associated mode. Some of these embodiments include providing a frame having a variable true number of QAM symbols. In some of these embodiments, the number of QAM symbols corresponds to the selected mode. In some of these embodiments, the number of related bytes and Reed-Solomon packets per frame is constant. In some of these embodiments, generating a frame of data packets using the perforated lattice code combination includes generating a puncturing pattern period of true number per data frame regardless of the associated mode. In some of these embodiments, the number of data bits per QAM symbol is a fraction for one or more modes. In some of these embodiments, the number of grating coder puncture pattern periods per frame is an integer for all modes.

Certain examples of the present invention provide a system for correcting phase offset. Some of these embodiments include a phase offset corrector that receives an equalized signal representing a quadrature amplitude modulated signal and derives a phase-corrected signal from the equalized signal. Some of these embodiments include a two-level slicer that slices the equalized signal to obtain real and imaginary sequences. Some of these embodiments include frame synchronizers that perform correlations between real and imaginary sequences and corresponding portions of stored frame-sync pseudo-random sequences. Some of these embodiments include phase correction signals provided by the frame synchronizer to the phase offset corrector. In some of these embodiments, the phase correction signal is based on the maximum real and imaginary values of the correlation. In some of these embodiments, the frame synchronizer performs continuous cross-correlation on the applied sliced quadrature amplitude modulated symbols. In some of these embodiments, continuous cross-correlation is performed separately for real and imaginary sequences with stored copies of binary frame-synchronous pseudo-random noise sequences. In some of these embodiments, quadrature amplitude modulated signals are modulated using a perforated grating code. In some of these embodiments, quadrature amplitude modulated signals are modulated using quadrature phase shift keying modulation. In some of these embodiments, the QAM signal is modulated using 16-QAM. In some of these embodiments, QAM is modulated using 64-QAM. In some of these embodiments, the frame sync symbols of the quadrature amplitude modulated signal have the same sign, and the sign of the largest real and imaginary values of the correlations represents the phase rotation in the equalized signal. In some of these embodiments, the phase correction signal provided by frame synchronization includes the sign of the largest real and imaginary values of the correlation. The phase offset corrector derives the phase-corrected signal by determining the phase correction value by indexing the LUT with the sign of the largest real and imaginary values of the correlation.

Certain embodiments of the present invention provide a method of correcting a carrier phase offset of a quadrature amplitude modulated signal at a receiver. Some of these embodiments include equalizing the signal. Some of these embodiments include slicing the equalized signal, thereby obtaining real and imaginary sequences from the equalized signal. Some of these embodiments include identifying frame synchronization sequences of real and imaginary sequences. In some of these embodiments, identifying the frame synchronization sequence includes correlating the stored pseudo-random sequence with real and imaginary sequences. In some of these embodiments, identifying the frame synchronization sequence includes determining the start of the frame from the maximum correlation value associated with the real and imaginary sequences. Some of these embodiments include correcting the phase error of the equalized signal based on the maximum correlation value.

In some of these embodiments, the correlating step includes performing continuous cross-correlation of the series of sliced quadrature amplitude modulated symbols and the stored copy of the binary frame-synchronous pseudo-random noise sequence. In some of these embodiments, the correlating step includes separately performing continuous cross-correlation to the stored copies of the frame synchronization sequence in real and imaginary sequences. In some of these embodiments, the frame sync symbols of the frame sync sequence have the same sign. In some of these embodiments, correcting the phase error includes determining the phase rotation of the equalized signal based on the sign of the maximum correlation value. In some of these embodiments, correcting the phase error of the equalized signal includes indexing the LUT with the sign of the real and imaginary maximum correlation values.

Certain embodiments of the present invention provide a method for correcting a carrier phase offset of a quadrature amplitude modulated signal. In some of these embodiments, the method may be implemented in a system that includes one or more processors configured to execute instructions. Some of these embodiments include executing instructions configured to equalize the signal on one or more processors. Some of these embodiments include executing, on one or more processors, instructions configured to slice the equalized signal, thereby obtaining real and imaginary sequences from the equalized signal. Some of these embodiments include executing instructions on one or more processors configured to identify frame synchronization sequences of real and imaginary sequences. In some of these embodiments, identifying the frame synchronization sequence includes performing continuous continuous cross-correlation individually to the stored copy of the frame synchronization sequence in real and imaginary sequences. In some of these embodiments, identifying the frame synchronization sequence includes determining the start of the frame from the maximum correlation value associated with the real and imaginary sequences. In some of these embodiments, executing instructions on one or more processors configured to correct phase error of the equalized signal based on the maximum correlation value. In some of these embodiments, the frame sync symbols of the frame sync sequence have the same sign. In some of these embodiments, correcting the phase error includes determining the phase rotation of the equalized signal based on the sign of the maximum correlation value.

Certain embodiments of the present invention provide a method of identifying constellation degrees of a symbol. In some of these embodiments, the method is performed by one or more processors in a multi-mode quadrature amplitude modulated communication system. Some of these embodiments include executing instructions that cause one or more processors to characterize the power distribution of the signal. In some of these embodiments, power distribution statistically tracks the occurrence of detected power levels in the signal. Some of these embodiments include executing instructions that cause one or more processors to determine one or more peak occurrences of power levels within the power distribution. Some of these embodiments include executing instructions that cause one or more processors to determine constellation degrees based on the distribution of peak occurrences.

In some of these embodiments, one or more processors determine constellation degrees based on the spread of one or more peak occurrences. In some of these embodiments, the signal is an equalized signal and one or more processors determine the constellation degree by examining a plurality of sections of the histogram of power distribution. In some of these embodiments, each section corresponds to a range of power levels associated with one but not all of the plurality of constellation candidates. In some of these embodiments, the plurality of constellation degree candidates includes an orthogonal phase shift key constellation diagram and a QAM constellation diagram. In some of these embodiments, the plurality of constellation degree candidates includes 16-QAM and 64-QAM constellation views. In some of these embodiments, the plurality of constellation degree candidates includes a 256-QAM constellation diagram.

Some of these embodiments include executing instructions that cause one or more processors to establish the reliability of the identified constellation degrees by performing the steps for each succession of constellation degrees determination. In some of these embodiments, this step includes incrementing the counter if the subsequent determination confirms the identity of the constellation diagram. In some of these embodiments, this step includes decrementing the counter if subsequent determinations identify different constellation degrees. In some of these embodiments, this step includes providing a measure of reliability based on the value of the counter.

In some of these embodiments, a counter is provided to each of the plurality of constellation degrees candidates, the constellation degrees being identified when the corresponding counter of the constellation degrees exceeds a threshold. In some of these embodiments, the peak occurrence of the power level corresponds to the corner symbol of the constellation diagram. In some of these embodiments, the constellation diagram is identified before the signal is equalized.

Certain embodiments of the present invention provide a method of identifying constellation degrees of a symbol in a multi-mode quadrature amplitude modulated communication system. In some of these embodiments, the method is performed by a processor of a modem of a communication system. In some of these embodiments, the processor includes executing an instruction to extract mode information from a data frame responsive to the detection of the start of a data frame received at the modem. Some of these embodiments include executing a command to cause the processor to determine the current constellation degree by selecting a code that most closely matches the corresponding code of the mode bit among the plurality of potential constellation degree codes. Some of these embodiments include that the processor executes an instruction to increase the confidence metric associated with the previously identified constellation if the current constellation also matches a previously determined constellation. Some of these embodiments include the processor executing the instruction to record the current constellation diagram as the previously identified constellation diagram and to reduce the reliability metric if the current constellation degree is different from the previously identified constellation diagram. Some of these embodiments include repeating the step of causing the processor to extract mode information, determine the current constellation degree, and adjust the confidence metric for subsequent data frames until the confidence metric exceeds a predetermined threshold. do. In some of these embodiments, constellation degrees are identified when the constellation degree metric exceeds a predetermined threshold.

In some of these embodiments, selecting the constellation degree code includes causing the processor to cross-correlate each of the plurality of potential constellation degree codes with corresponding code bits. In some of these embodiments, constellation degrees are identified in the equalized signal carrying the data frame and subsequent data frames. In some of these embodiments, the constellation diagram is identified while the processor is recovering the carrier from the signal. Some of these embodiments include executing instructions that cause the processor to compute an error signal using a constant counting algorithm that converges the equalizer filter taps to allow equalization of the signal. In some of these embodiments, the error signal is calculated using scaled CMA parameters to improve equalization performance. In some of these embodiments, performing equalization of the signal includes analyzing a histogram of the power of the equalized signal. In some of these embodiments, analyzing the histogram includes using a probability mass function. In some of these embodiments, performing signal equalization includes executing instructions that cause the processor to calculate power associated with the plurality of symbols of the equalized signal. In some of these embodiments, performing signal equalization includes executing instructions that cause the processor to identify corner symbols of the constellation diagram by using a threshold power level. In some of these embodiments, the threshold power level represents the identity of the constellation diagram.

Certain embodiments of the present invention receive from the video two signals each representing a sequence of images captured by the camera, transmit one of the two signals as a composite passband video signal, and transmit the other signal with the baseband signal. A system for transmitting a video signal comprising a camera-side modem configured to modulate and transmit as a non-overlapping passband video signal. In some of these embodiments, the camera-side modem includes a mixer that combines baseband and passband video signals to provide a transmission signal. In some of these embodiments, the camera-side modem includes a diplexer configured to transmit the transmission signal over the communication line and extract the passband signal received from the transmission line. In some of these embodiments, the camera side modem includes a detector that monitors the camera side modem and generates an enable signal when a received passband signal is identified. In some of these embodiments, the enable signal controls the transmission of at least one of a baseband video signal and a passband video signal.

In some of these embodiments, the passband video signal is transmitted only when the enable signal is generated. In some of these embodiments, the received passband signal is quadrature amplitude modulated. In some of these embodiments, the detector monitors an estimate of the mean squared error in the quadrature amplitude modulator, and an enable signal is generated if the estimate exceeds a threshold. In some of these embodiments, the detector monitors the constellation degree detector. In some of these embodiments, the enable signal is generated based on a measure of reliability provided by the constellation degree detector. In some of these embodiments, the measure of reliability is based on a sequence of frame synchronization. In some of these embodiments, the detector monitors an estimate of the mean squared error of the equalizer. In some of these embodiments, the enable signal is generated if the estimate exceeds a threshold. In some of these embodiments, the detector detects the gain component of the automatic gain control module of the camera side modem. In some of these embodiments, the enable signal is generated if the gain component has a value below the threshold. In some of these embodiments, the detector monitors the amplitude of the received passband signal. In some of these embodiments, the enable signal is generated if the amplitude has a value above the threshold. In some of these embodiments, the received passband signal includes data encoded according to the Internet protocol.

Certain embodiments of the present invention provide a method of controlling a signal in a security system. Some of these embodiments include detecting, in an upstream modem, the presence of an upstream QAM signal in the composite signal transmitted on the coaxial cable. Some of these embodiments include allowing a strict stream modem to transmit a composite baseband video signal and a passband video signal on a coaxial cable when it is determined that an upstream QAM signal is present. In some of these embodiments, the composite baseband video signal and the passband video signal are simultaneous representations of the sequence of images captured by the video camera. Some of these embodiments include allowing the upstream modem to transmit a composite baseband video signal on the coaxial cable and to prevent transmission of the passband video signal if it is determined that the upstream QAM signal is absent.

In some of these embodiments, an upstream QAM signal is determined to be present if the gain value of the automatic gain control signal exceeds a threshold. In some of these embodiments, the upstream QAM signal is determined to be present if the estimate of the amplitude of the upstream QAM signal is below a threshold. In some of these embodiments, the upstream QAM signal is determined to be absent if the estimate of the mean squared error of the equalizer exceeds the threshold. In some of these embodiments, the upstream QAM signal is determined to be absent if an Internet Protocol data packet is identified in the upstream QAM signal.

Certain embodiments of the present invention provide a system that can be automatically configured to transmit a video signal. Some of these embodiments include an upstream modem configured to receive two signals from a video camera. In some of these embodiments, each signal represents a sequence of images captured by the camera. In some of these embodiments, the upstream modem transmits one of the two signals as a composite baseband video signal and is configured to modulate and transmit the other signal as a passband video signal that does not overlap with the baseband signal. Some of these embodiments include a downstream modem configured to receive a composite baseband video signal and a passband video signal from an upstream modem and transmit the upstream passband signal to the upstream modem. In some of these embodiments, the upstream modem stops transmitting at least one of the two signals when the upstream modem detects a degradation in the upstream passband signal.

Although the present invention has been described with reference to specific exemplary embodiments, it will be apparent to those skilled in the art that various changes and modifications may be made without departing from the broad spirit and scope of the invention. For example, a system has been described that provides compressed digital HD video simultaneously with a baseband analog video signal. Yet another embodiment of the present invention provides simultaneous standard digital and analog supplies. Another embodiment provides full frame rate digital HD video analog with baseband analog video. The description and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.

Claims (109)

A processor that receives the image sensor from the image sensor to generate a plurality of video signals representing the image signal, the video signal comprising a baseband signal and a digital video signal; And
And an encoder for combining the baseband video signal and the digital video signal as an output signal for transmission over a cable. The method of claim 1,
And the combined baseband and digital video signal is substantially isochronous. The method of claim 1,
The baseband video signal comprises a standard analog video signal. The method of claim 1,
And the digital video signal is modulated prior to combining with the baseband video signal. The method of claim 4, wherein
And the digital video signal is a high definition digital video signal. The method of claim 4, wherein
The frame rate of the digital video signal is less than the frame rate of the image signal. The method of claim 4, wherein
The modulated digital signal is provided to a video capture device. The method of claim 1,
And a decoder configured to demodulate an upstream signal received from the cable, wherein the demodulated upstream signal comprises control signals. The method of claim 8,
The control signal includes a signal for controlling the position and direction of the camera. The method of claim 8,
The control signal comprises a signal for controlling generation of the baseband video signal and the digital video signal by a processor. The method of claim 10,
The control signal comprises a signal for selecting a portion of the image signal that encodes the baseband video signal. The method of claim 10,
The control signal comprises a signal for selecting a portion of the image signal that encodes as the digital video signal. The method of claim 8,
The demodulated upstream signal comprises an audio signal for driving an audio output of the camera. Frequency division multiplexing the video signal received from the video imaging video to obtain a modulated digital signal;
Generating an output signal by combining the modulated digital signal with a baseband analog signal representing the video signal; And
And transmitting the output signal to one or more devices including a digital video capture device and a device displaying a video image derived from the baseband analog signal. The method of claim 14,
The digital video capture device records a sequence of frames extracted from the modulated digital signal. The method of claim 14,
And the digital video capture device comprises a video server. The method of claim 14,
Frequency division multiplexing the digital video signal,
Compressing the video signal; And
Modulating the compressed video signal on a carrier wave. The method of claim 14,
The step of transmitting the output signal includes providing the output signal to a coaxial cable, and demodulating an input signal received from the coaxial cable to obtain a control signal. 19. The method of claim 18,
Generating the baseband analog signal by encoding a portion of the video signal in a composite video signal; And
Using the control signal to select a portion of the video signal to be encoded in the composite video signal. 19. The method of claim 18,
Controlling the position of the camera using the control signal. 19. The method of claim 18,
Demodulating the input signal comprises extracting an audio signal from the input signal. Receiving an image signal from an image sensor to generate a plurality of video signals, each of the plurality of video signals representing at least a portion of a field of view of the camera, wherein the plurality of video signals comprises a baseband video signal and a digital video signal, A processor;
Control logic configured to respond to a control signal received by the camera, the control signal controlling content of the baseband and digital video signals; And
A modulator configured to modulate the digital video signal, wherein the modulated digital video signal and the baseband video signal are simultaneously transmitted by the camera. The method of claim 22,
The control signal shifts a portion of the field of view represented by at least one of the plurality of video signals. The method of claim 22,
The control signal is received as a wireless signal. The method of claim 22,
The modulated digital video signal is transmitted wirelessly. The method of claim 22,
And the baseband and digital video signals are substantially isochronous. Further comprising an encoder combining the baseband video signal and the modulated digital video signal as an output signal for transmission over a cable. A system for use as a baseband analog signal and a digital signal separated by frequency and carried by a cable, the system comprising:
A digital equalizer for removing distortion from the digital signal received at the receiver; And
An analog equalizer for compensating for the attenuation of the analog signal by the cable,
The analog equalizer applies one of a set of baseband analog filters to compensate for the attenuation, and the applied baseband analog filter is selected based on an estimate calculated by the digital equalizer of the difference in the attenuation at different frequencies. The method of claim 28,
The digital signal and the analog signal are transmitted between a receiver and a transmitter embedded in a camera, the receiver providing an equalized signal representing the analog signal to a monitor. 30. The method of claim 29,
And the cable comprises a coaxial cable. The method of claim 30,
The distortion increases with the length of the cable. The method of claim 28,
Wherein the distortion comprises multipath distortion. The method of claim 32,
The difference in the attenuation comprises an estimate calculated from a frequency band having a power spectral density of approximately tilt. 34. The method of claim 33,
Wherein the slope is calculated using a fast Fourier transform for the plurality of filter taps. 34. The method of claim 33,
Frequency bins in the frequency band are summed:

Is selected to accept the frequency response of the digital equalizer using
Where G [k] is the fast Fourier transform (DFT) of the time-domain converged equalizer filter tap and k1 corresponds to a specific frequency bin of the DFT. The method of claim 28,
The digital signal comprises a high quality representation of a video image captured by a camera and the analog signal comprises a standard representation of the video image. A method of equalizing an analog signal in a cable carrying a digital signal separated from an analog signal by frequency, the method being performed by a modem receiving the analog signal and the digital signal and outputting a baseband video signal, The method,
Calculating a slope in the digital signal, the slope characterizing the attenuation as a function of frequency contributing to the cable;
Equalizing the digital signal based on the calculated slope;
Configuring an analog equalizer by using the calculated slope to select one of a set of baseband analog filters; And
Equalizing the analog signal using the selected baseband analog filter. The method of claim 37, wherein
And the analog signal comprises a baseband video signal and the digital signal comprising a high quality version of the baseband video signal. The method of claim 38,
The cable comprises a coaxial cable, the slope varying with the length of the cable. 40. The method of claim 39,
And the slope is derived from multi-path distortions. The method of claim 37, wherein
The step of calculating the slope includes estimating the attenuation in a frequency band having a power spectral density whose slope is approximately linear. 42. The method of claim 41 wherein
Estimating the attenuation uses a Fast Fourier Transform for a plurality of filter taps. The method of claim 42, wherein
Estimating the attenuation comprises selecting frequency bins in the frequency band, the selected frequency bins optimizing the efficiency of the step of calculating the slope. A convolutional byte interleaver interleaving a frame of data and synchronized to the frame structure;
A randomizer for generating a randomized data frame from the interleaved data frame;
A punched trellis code modulator operating at a selected code rate that generates a grid coded data frame from the randomized data frame;
A QAM matcher that matches a group of bits in the lattice coded data frame to modulation symbols to provide a matched frame; And
And a synchronizer for adding a synchronization packet to the matched frame. The method of claim 44,
And the perforated grating code modulator is bypassed to obtain an optimized net bit rate against r measured white noise performance of the system. The method of claim 44,
The synchronization packet is added to each sequence of matched frames. The method of claim 46,
And a portion of the synchronization packet includes different binary sequences for real and imaginary parts of the modulation symbols. The method of claim 46,
And a portion of the synchronization packet includes the same binary sequence for the real part and the imaginary part of the modulation symbols. The method of claim 48,
And the synchronization packet includes data indicative of a transmission mode for the matched frame. The method of claim 49,
The data indicative of the transmission mode comprises a selected QAM constellation diagram and a selected grid code rate. The method of claim 44,
The system generates a constant integral number of Reed-Solomon packets for each data frame regardless of the mode of transmission. The method of claim 44,
The system generates a true number of modulation symbols for each data frame regardless of the transmission mode. The method of claim 44,
And the system generates a true number of puncturing pattern periods for each data frame regardless of the transmission mode. A method of framing for a variable net bit rate digital communication system,
Providing a set of different quadrature amplitude modulation (QAM) constellations;
Generating a frame of data packets using a punched grid code combination corresponding to each associated mode; And
Providing a frame having a variable true number of QAM symbols,
Wherein the number of QAM symbols corresponds to the selected mode, and the number of related bytes and Reed-Solomon packets per frame is a constant. The method of claim 54,
Generating a frame of data packets using the perforated grid code combination comprises generating a puncturing pattern period of true number per frame of data regardless of the associated mode. Frame way. The method of claim 54,
Wherein the number of lattice coder puncture pattern periods per frame is an integer for all modes. 56. The method of claim 55,
The method of frame for a variable net bit digital communication system, wherein the number of data bits per QAM symbol for one or more modes is a fraction. A phase offset corrector that receives an equalized signal representative of a quadrature amplitude modulated signal and derives a phase-corrected signal from the equalized signal;
A two-level slicer that slices the equalized signal to obtain real and imaginary sequences;
A frame synchronizer for correlating the real and imaginary sequences with corresponding portions of a stored frame-synchronous pseudo-random sequence; and
A phase correction signal provided by the frame synchronizer to the phase offset corrector, the phase correction signal based on the maximum real and imaginary values of the correlation. The method of claim 58,
And the frame synchronizer performs continuous cross-correlation on an applied sliced quadrature amplitude modulated symbol. The method of claim 59,
Wherein the continuous cross-correlation is performed separately for the real and imaginary sequences with a stored copy of a binary frame-synchronous pseudo-random noise sequence. The method of claim 58,
And the quadrature amplitude modulated signal is modulated using a perforated grating code. The method of claim 58,
And the quadrature amplitude modulated signal is modulated using quadrature phase shift keying modulation. The method of claim 58,
And the quadrature amplitude modulated signal is modulated using 16-QAM. The method of claim 58,
And the quadrature amplitude modulated signal is modulated using 64-QAM. The method of claim 58,
The frame sync symbols of the quadrature amplitude modulated signal have the same sign, and the maximum real and imaginary values of the correlations represent the phase rotation of the equalizer signal. 66. The method of claim 65,
And the phase correction signal provided by the frame synchronizer comprises a sign of the maximum real and imaginary values of the correlation. 66. The method of claim 65,
And the phase offset corrector slaughters a phase-corrected signal and indexes a lookup table with the sign of the maximum real and imaginary values of the correlation to determine the phase correction value. A method of correcting a carrier phase offset of a quadrature amplitude modulated signal at a receiver, the method comprising:
Equalizing the signal;
Slicing an equalized signal to obtain real and imaginary sequences from the equalized signal; And
Identifying a frame sync sequence in the real and imaginary sequences,
Identifying the frame sync sequence,
Correlating a stored pseudo-random sequence with the real and imaginary sequences;
Determining the start of a frame from the maximum correlation value associated with the real and imaginary sequences; And
Correcting a phase error in the equalized signal based on the maximum correlation value. The method of claim 68, wherein
Wherein said correlating step comprises performing continuous cross-correlation on a series of sliced orthogonal amplitude modulated symbols with a copy of a binary frame-synchronous pseudo-random noise sequence. The method of claim 68, wherein
And said correlating step comprises performing continuous cross-correlation on stored copies of frame synchronization sequences individually in said real and imaginary sequences. The method of claim 70,
And frame synchronization symbols of the frame synchronization sequence have the same sign. The method of claim 71, wherein
Correcting the phase error comprises determining the right-hand rotation of the equalized signal based on the sign of the maximum correlation value. The method of claim 72,
Correcting the phase error in the equalized signal comprises indexing a lookup table with real and imaginary maximum correlation values. A method of correcting a carrier phase offset of a quadrature amplitude modulated signal, the method being implemented in a system comprising one or more processors configured to execute instructions, the method comprising:
Executing instructions on the one or more processors configured to equalize the signal;
On the one or more processors, executing the instructions configured to slice the equalized signal to obtain real and imaginary sequences from the equalized signal;
Executing, on the one or more processors, an instruction configured to identify a frame synchronization sequence in the real and imaginary sequences, wherein identifying the frame synchronization sequence comprises:
Performing continuous cross-correlation on stored copies of the frame synchronization sequence separately in the real and imaginary sequences; And
Determining a start of a frame from a maximum correlation value associated with the real and imaginary sequences, wherein
Executing instructions on the one or more processors configured to identify a frame synchronization sequence; And
Executing, on the one or more processors, an instruction configured to correct phase error in the equalized signal based on the maximum correlation value,
The frame sync symbol of the frame synchronization sequence has the same sign, and correcting the phase error between includes determining the phase rotation of the equalized signal based on the sign of the maximum correlation value. A method of identifying constellation degrees of symbols, the method being performed by one or more processors in a multi-mode quadrature amplitude modulated communication system, the method comprising:
Executing instructions to characterize the power distribution, wherein the one or more processors characterize power distribution in a signal, wherein the power distribution statistically tracks the occurrence of a detected power level in the signal. Doing;
Executing instructions to cause the one or more processors to determine one or more peak occurrences of power levels in the power distribution; And
Executing instructions that cause the one or more processors to determine the constellation degree based on the distribution of peak occurrences. 76. The method of claim 75,
And wherein the one or more processors determine the constellation degree based on a spread of the one or more peak occurrences. 76. The method of claim 75,
The signal is an equalized signal and the one or more processors determine the constellation degree by examining a plurality of sections of the histogram of the power bonsai, each of the sections having a range of power levels associated with one but not all of the plurality of constellation degrees candidates. Corresponding to the constellation of symbols. 78. The method of claim 77 wherein
Wherein the plurality of constellation degrees candidates comprises a quadrature phase shift key constellation diagram and a quadrature amplitude modulation (QAM) constellation diagram. The method of claim 78,
Wherein the plurality of constellation diagram candidates comprises 16-QAM and 64-QAM constellation diagrams. The method of claim 78,
And the plurality of constellation degree candidates comprise 256-QAM constellation views. 76. The method of claim 75,
Executing the instructions that cause the one or more processors to establish the reliability of the identified constellation degrees by performing a step for each successive series of constellation degrees determinations, wherein the steps for each successive series of constellation degrees determinations include:
Incrementing a counter if a subsequent determination confirms the identity of the constellation diagram;
Decrementing the counter if the subsequent determination identifies a different constellation degree; And
Providing a measure of confidence based on the value of the counter. 82. The method of claim 81 wherein
And the constellation diagram is reliably identified if the counter exceeds a threshold. 82. The method of claim 81 wherein
A counter is provided for each of a plurality of constellation diagram candidates, wherein the constellation diagram is reliably identified when the corresponding counter of the constellation diagram exceeds a threshold. 76. The method of claim 75,
And generating a peak of the power level corresponds to a corner symbol of the constellation diagram. 76. The method of claim 75,
And the constellation diagram is identified after the signal is equalized. A method of identifying constellation degrees of a symbol in a multi-mode quadrature amplitude modulated communication system, the method being performed by a processor in a modem of the communication system,
In response to determining the start of a data frame received at the modem, executing an instruction to cause the processor to extract mode information from the data frame; the processor to correspond to the code of the mode information from a plurality of potential constellation diagram codes. Executing a command to determine a current constellation degree by selecting a code that most closely matches.
Executing the instruction to cause the processor to increase a reliability metric associated with the previously identified constellation when the current constellation matches the previously determined constellation;
If the current constellation is different from the previously identified constellation, executing the instruction to cause the processor to reduce the reliability metric and record the current constellation as the previously identified constellation; And
Repeating the step of causing the processor to extract mode information, select a current constellation degree, and adjust the confidence metric for subsequent data frames until the confidence metric exceeds a predetermined threshold; And wherein the constellation diagram is identified when the reliability metric exceeds the predetermined threshold, the constellation diagram of a symbol in a multi-mode quadrature amplitude modulated communication system. 89. The method of claim 88 wherein
Selecting the constellation diagram code comprises causing the processor to cross-correlate the plurality of potential constellation diagram codes with the corresponding code bits for the symbol in a multi-mode quadrature amplitude modulated communication system. How to identify constellation diagrams. 87. The method of claim 86,
24. A method of identifying a constellation of symbols in a multi-mode quadrature amplitude modulated communication system, wherein an always constellation is identified in an equalized signal carrying the data frame and subsequent data frames. 89. The method of claim 88 wherein
And the constellation diagram is identified while the processor recovers a signal from the signal. 89. The method of claim 88 wherein
And executing, by the processor, a command to converge an equalizer filter tap using a constant module algorithm (CMA) to authorize equalization of the signal. How to identify constellation diagrams. 89. The method of claim 88 wherein
And the error signal is calculated using a scaled CAM parameter to improve equalization performance. 89. The method of claim 88 wherein
Performing equalization of the signal includes analyzing a histogram of the power of the equalized signal, and analyzing the histogram comprises using a rate mass function. A method of identifying the constellation diagram of a symbol in a communication system. 89. The method of claim 88 wherein
Performing the equalization of the signal may be performed by the processor.
Calculate power associated with a plurality of symbols in the equalized signal; And
Executing a command to identify a corner symbol of the constellation diagram by using a threshold power level, wherein the threshold power level represents the constellation diagram of the symbol in a multi-mode quadrature amplitude modulated communication system, representing the identity of the constellation diagram. How to identify. Receive two signals from a video camera, each representing a sequence of images captured by the video camera, and transmit one of the two signals as a composite baseband video signal, and do not overlap the other signal with the baseband signal. A system for transmitting a video signal comprising a camera side modem configured to modulate and transmit as a passband video signal, the camera side modem comprising:
A mixer for combining a strong baseband video signal with the passband video signal to provide a transmission signal;
A diplexer configured to transmit the transmission signal via a transmission line and to extract a passband signal received from the transmission line; And
A detector for monitoring the camera side modem to generate an enable signal if the received passband signal is identified;
The enable signal controls transmission of at least one of the baseband video signal and the passband video signal. 95. The method of claim 94,
And the passband video signal is transmitted only when the enable signal is generated. 95. The method of claim 94,
And the received passband signal is quadrature amplitude modulated. 97. The method of claim 96,
The detector monitors an estimate of the mean squared error of a quadrature amplitude demodulator and the enable signal is generated if the estimate exceeds a threshold. 97. The method of claim 96,
The detector monitors a constellation degree detector and the enable signal is generated based on a measure of the reliability provided by the constellation degree detector. 97. The method of claim 96,
And the measure of reliability is based on a sequence of frame synchronization. 95. The method of claim 94,
The detector monitors an estimate of the mean squared error of the equalizer, and wherein the enable signal is generated if the estimate exceeds a threshold. 95. The method of claim 94,
The detector monitors a gain component of an automatic gain control module of the camera side modem, wherein the enable signal is generated when the gain component is below a threshold value. 95. The method of claim 94,
The detector monitors the amplitude of the received passband signal and the enable signal is generated when the amplitude has a value above a threshold. 95. The method of claim 94,
And the received passband signal comprises data encoded according to an internet protocol. An upstream modem for determining the presence of an upstream QAM signal of the composite signal transmitted on the coaxial cable,
If it is determined that the upstream QAM video signal is present, causing the upstream modem to transmit a composite baseband video signal and a passband video signal over a coaxial cable, wherein the composite baseband video signal and the passband video signal are video cameras. The transmitting, which is a simultaneous representation of the sequence of images captured by the; And
If the upstream QAM signal is determined to be absent, causing the upstream modem to transmit the composite baseband video signal over the coaxial cable, and preventing transmission of the passband video signal. Way. 105. The method of claim 104,
And said upstream QAM signal is determined to be present if the gain value of the automatic gain control signal exceeds a threshold. 105. The method of claim 104,
And the upstream QAM signal is determined to be present if the measurement of the amplitude of the upstream QAM signal is below a threshold. 105. The method of claim 104,
And said upstream QAM signal is determined to be absent if an estimate of the mean squared error of the equalizer exceeds a threshold. 105. The method of claim 104,
And said upstream QAM signal is determined to be absent if an internet protocol data packet is identified in said upstream QAM signal. Receive two signals from a video camera, each representing a sequence of images captured by the video camera, transmit one of the two signals as a composite baseband video signal, and do not overlap the other signal with the baseband signal. An upstream modem configured to modulate and transmit as a passband video signal; And
A downstream modem configured to receive an always synthesized baseband video signal and a passband video signal from the upstream modem and to transmit the upstream passband signal to the upstream modem;
And the upstream modem stops transmitting at least one of the two signals when the upstream modem detects degradation of the upstream passband signal.

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