CN102365832B - Mixed format medium transmission system and method - Google Patents

Abstract

Describe the system and method for operating video camera.The multiple vision signals that can be used as token image signal from the picture signal of imageing sensor reception process.Base band and digital video signal can be combined in output signal to transmit on cable by encoder.Base band when vision signal can comprise basic grade and digital video signal.Decoder to stream signal demodulation to obtain control signal, this control signal for control video camera position and towards and the content of base band and digital video signal.The system and method described receives multi-signal, provides the synchronizing information associated with signal, the phase shift offset in correction signal, and adopts or detect the encoding scheme used in signaling procedure.Describe the system and method existed for detection signal.

Description

Mixed format medium transmission system and method

Background of invention

Invention field

The present invention relates generally to multi-media transmission system and relate more specifically to transmit the system and method for high-definition digital video and SD analog video on single cable.

Background technology

Along with the appearance of digital broadcast television and stream video technique, many have the digital camera, monitor and the video recorder that strengthen definition and advanced and become available.Nowadays closed-circuit television (CCTV) system provides the digital video signal of HD video output and compression for such as building management, the access control of facility and the occasion of telemonitoring.But old system rests on original place and SD analog video signal is in and widely uses and still can continue to use in the process being transitioned into digital, high definition system.Specifically, coaxial cable (" coaxially ") has been deployed the signal of carrying from CCTV video camera to monitoring station.Some CCTV video cameras disposed are in local area network (LAN) or the compressed vision signal of wide area transfers on network, and these video cameras can use Internet protocol (IP) as means of communication to transmit compressed vision signal.

Fig. 1 illustrates the legacy system using coaxial cable carrier SD analog video.Basic analog video camera 10 generally produces and coaxial cable 11 can be used to transmit the composite video baseband signal (CVBS) reaching 300 meters.CVBS signal is provided to a kind of video recording system usually, and this video recording system often comprises the digital video recordings device (DVR) 12 recording CVBS in a digital format.Conventional monitors 14 can be connected to DVR12 to show this SD analog video simultaneously, and this SD analog video generally has the resolution of 720*480 pixel.

Digital camera 16 can replace analog video camera 10 in some applications.Digital camera 16 can support serial digital interface (SDI), and the standard-definition digital video of uncompressed the speed of about 270Mbps can be transferred to DVR by this serial digital interface on coaxial cable 17.

Fig. 2 illustrates the conventional method transmitting HD video (1920 × 1080 pixel) in current deployment system.First, digital camera 20 can support high definition serial digital interface (HD-SDI), and the high-definition digital video of uncompressed the speed of 1.5Gbps can be transferred to DVR22 by HD-SDI on coaxial cable 21.The cable distance supported under such high transmission rates can reach 100 meters.Secondly, IP-based high definition (HD) video camera 24 can use standard class 5 (CAT5) twisted-pair cables 25 on 100Mbps Ethernet, produce compressed digital HD video signal, and its transmission range is 100 meters.This signal DVR22 receives and is recorded to do non real-time playback.Existing coaxial cable 26 can be used to use class 5-coaxial cable bridge joint modulator-demodulator 27,29 or other conversion equipment that video is transferred to DVR22 from video camera 24.Make the use of the networking network of camera transmissions digital video allow these systems to add some upstream communications, be generally control and audio signal 28.

Summary of the invention

Some embodiment of the present invention provides the system and method for video camera and operation video camera.Processor can receive picture signal from imageing sensor and produce multiple vision signals of token image signal.Encoder is for being combined as the output signal transmitted on cable by baseband video signal and digital video signal.Baseband video signal when vision signal can comprise basic grade and digital video signal.Video camera can be used as closed circuit HDTV (High-Definition Television) video camera.

According to some aspect of the present invention, baseband video signal can comprise SD analog video signal and digital video signal can be modulated before combining with baseband video signal.Digital video signal can comprise compressed high-definition digital video signal.The frame rate of digital video signal can lower than the frame rate of picture signal, especially when modulated digital signal is supplied to video recorder.

In certain embodiments, configuration decoder is to carry out demodulation to from transmission cable or from the stream signal that cordless communication network receives, and this transmission cable is used for carrying downstream visual.Stream signal through demodulation can comprise control signal, described control signal comprise control the position of video camera and direction signal, control to produce the signal of baseband video signal and digital video signal by processor and select a part for picture signal using the signal carrying out encoding as baseband video signal.Control signal also can comprise selects a part of picture signal to carry out the signal of encoding and the audio signal for driving the camera audio of such as loud speaker to export as digital video signal.

Some embodiment of the present invention provides the method for transmitting video image.These methods can comprise: carry out frequency division multiplexing to obtain modulated digital signal to the vision signal received from high definition imaging device; Output signal by modulated digital signal is combined to produce with the base-band analog signal characterizing vision signal and output signal be sent to monitor and digital video storage device simultaneously.These embodiments some in, the baseband analog of monitor display video signal characterizes and/or Digital video storage device uses digital video recordings device to record a series of high definition frames extracted from modulated digital signal.Digital video signal can be compressed.

In certain embodiments, transmission output signal comprises and output signal is supplied to coaxial cable and/or is supplied to transmitting set.Can the input signal that receives from coaxial cable or wireless network of demodulation to obtain control signal.Base-band analog signal is by producing a part of Video signal encoding in composite video signal, and the plan part vision signal be coded in composite video signal can use control signal to control.Control signal can control the position of video camera.Demodulation input signal can produce an audio signal from input signal extraly.

Some embodiment of the present invention is provided for the system and method operating video camera.Processor can receive picture signal from imageing sensor and produce multiple vision signal, and control logic can be configured to respond to the control signal received by video camera and modulator can be configured to modulation digital vision signal to obtain modulated signal.Multiple vision signal can comprise baseband video signal and digital video signal.Each in multiple vision signal represents camera field at least partially, and control signal can control the content of base band and digital video signal.Modulated signal and baseband video signal are transmitted by video camera usually simultaneously.

Base band and digital video signal can be substantially etc. time.Encoder can be combined as the output signal for transmitting on cable by baseband video signal with through modulation signal.Such as, can wirelessly from wireless network reception control signal.Can wirelessly transmit at least in part through modulation signal.Digital video signal can be high-definition digital video signal, and can be compressed digital video signal.Control signal moves the part by the ken characterized of in vision signal.

Accompanying drawing is sketched

Fig. 1 explains orally the prior art systems using coaxial cable carrier SD analog video.

Fig. 2 explains orally the art methods of transmission high-definition digital video.

Fig. 3 describes the system for transportation simulator and digital video according to some aspect of the present invention.

Fig. 4 describes the networked systems for transportation simulator and digital video according to some aspect of the present invention.

Fig. 5 illustrates the allocated bandwidth for transportation simulator and digital video on coax according to some aspect of the present invention.

Fig. 6 explains orally the example of the CCTV camera equipment according to some aspect of the present invention structure.

Fig. 7 explains orally the example of the modulator-demodulator used in DVR equipment according to some aspect of the present invention structure.

Fig. 8 explains orally the example of the modulator-demodulator used in network exchange equipment according to some aspect of the present invention structure.

Fig. 9 is the example of the frame structure used in ATSC Digital Television.

Figure 10 is the example that conventional frame is synchronously divided into groups.

Figure 11 is the example of the data sectional in conventional data frame.

The simplification diagram that Figure 12 provides frame to configure.

Figure 13 is the block diagram of the modulator according to some aspect of the present invention.

Figure 14 is the block representation of the frame structure adopted in some embodiments of the invention.

Figure 15 illustrates the operation of convolutional byte interleaver in some embodiments of the invention.

Figure 16 is the block diagram of the selectable code rate puncture trellis coded modulation adopted in certain embodiments of the invention.

Figure 17 explains orally the example that QAM maps.

Figure 18 illustrates frame synchronization/mould grouping.

Figure 19 is the simplification frame structure adopted in some embodiments of the invention.

Figure 20 is the block diagram of the demodulator according to some aspect of the present invention.

Figure 21 is the block diagram of the camera side modulator-demodulator according to some aspect of the present invention.

Figure 22 is the block diagram of the monitor side modulator-demodulator according to some aspect of the present invention.

Figure 23 explains orally the camera side base band-passband QAM modulation device according to some aspect of the present invention.

Figure 24 A and 24B explains orally the monitor side passband-base band qam demodulator according to some aspect of the present invention.

Figure 25 explanation is according to the monitor side digital equalizer of some aspect of the present invention and carrier phase/frequency loop.

Figure 26 illustrates because of the decay that the frequency become in coaxial cable is described.

Figure 27 A describes the power spectral density (PSD) of equalizer input.

Figure 27 B illustrates the amplitude response of the equalizer tap of convergence.

Figure 28 A, 28B, 29A and 29B illustrate that loss under different frequency in passband digital video signal is relative to crooked.

Figure 30 illustrates the monitor modulator-demodulator in qam demodulator with digital equalizer according to some aspect of the present invention.

Figure 31 describes the Active Analog Filter being suitable for equalization base band CVBS according to some aspect of the present invention.

Figure 32 illustrates the example of the filter response in certain embodiments of the invention.

Figure 33 A and 33B is the QPSK planisphere of the rotation explained orally in complex plane.

Figure 34 explains orally the block diagram according to the phase correction process of some aspect of the present invention.

Figure 35 describes integration ratio (IP) filter according to some aspect of the present invention.

Figure 36 explains orally the code element of a transmission.

Figure 37 A, 37B, 37C and 37D explain orally the possible recovery code element based on the transmission code element of Figure 36.

Figure 38 illustrates the example of the phase shift in receiving symbol.

Figure 39 illustrates the example based on the typical real part of frame synchronization code element and the transmission planisphere of imaginary part.

Figure 40 is the block diagram of the expression phase offset correction device adopted in some embodiments of the invention.

Figure 41 explanation is for determining the process of the reliability relevant with frame synchronization.

Figure 42 describes some aspect of equalizer and the carrier phase/frequency loop adopted in some embodiments of the invention.

Figure 43 illustrates the sheer and phase error detector module that adopt in some embodiments of the invention.

The complex exponential LUT module that Figure 44 explanation adopts in some embodiments of the invention.

Figure 45 A, 45B illustrate the real part that the equilibrium in QPSK signal (Figure 45 A) and 16-QAM signal (Figure 45 B) exports.

Figure 46 A, 46B and 46C be when constellation be QPSK (Figure 46 A), 16-QAM (Figure 46 B) and 64-QAM (Figure 46 C) time, the distribution map of the power using equalizer convergence to export in the equilibrium that embodiment of R=58 produces.

Figure 47 explains orally the example recovering the constellation of loop module input in equalizer output and carrier phase/frequency.

Figure 48 illustrates the example that the QAM with the threshold value depicted maps.

Figure 49 illustrates the right upper quadrant of all three constellations covered on the same area.

The operation of a kind of way of constellation is determined in Figure 50 explanation.

Figure 51 A and 51B describes the system for transmitting SD and HD video simultaneously according to some aspect of the present invention, and this system has tap or signal interruption.

Figure 52 A and 52B explains orally the process producing frame-synchronizing impulse according to some aspect of the present invention from noise signal.

Figure 53 is the block diagram with the camera side modulator-demodulator of the indicating device that coaxial cable connects according to some aspect of the present invention.

Figure 54 explains orally some aspect of automatic gain control loop.

Embodiment

Be described in detail embodiments of the invention referring now to accompanying drawing, these accompanying drawings provide to enable those skilled in that art realize the present invention as illustrative example.Obviously, accompanying drawing below and example are not intended to scope of the present invention to be limited in the scope of single embodiment, but make other embodiment become possibility by exchanging some or all of described or shown key element.No matter where, as long as convenient, namely run through institute's drawings attached and use identical Reference numeral to represent same or analogous parts.When some key element of these embodiments can use known elements partly or entirely to realize, only be described understanding the present invention's those parts necessary in these known elements, and save with unlikely, the present invention is obscured to the detailed description of the other parts of these known elements.In this manual, illustrate that the embodiment of single part should not be considered as restriction; On the contrary, the present invention is intended to contain other embodiment comprising multiple same parts, and vice versa, unless explicit state is really not so in this article.In addition, applicant is not intended to make any term in specification or claims to ascribe common or not special implication to, unless clearly illustrated, the way it goes.In addition, the present invention contain herein by the parts explaining orally citation at present and future known equivalent.

Some embodiment of the present invention provides the system and method allowing video camera simultaneously to send high-definition digital video and SD analog video on coax.High-definition camera is suitable for producing compressed digital video signal and analog baseband signal.Digital signal is modulated and is sent in the frequency range be separated with the upper frequency of baseband video signal.The standard that can need according to any conjunction is encoded to analog signal, comprises PAL, SECAM and NTSC standard and distortion thereof.

For the purpose of description, the example of the system adopting safety chain (SLOC) on coaxial cable will be described.In addition, SLOC is generally described as and has upstream and downstream signal relative to video camera: position for video camera is in upstream.In this manual, the example of SLOC system is provided in downstream high definition (HD) vision signal in the first passband, the upstream audio frequency in the second passband and control signal and downstream composite video baseband signal (CVBS).To understand, other passband signal and allocated bandwidth can be used.Such as, system can adopt the two-way digital video signal of SD or high definition resolution.

Fig. 3 describes the embodiments of the invention explaining orally some operation principle of the present invention.This example is described to wish and is about to the video record of high definition version to the deployment watching the HD video camera 30 in the system of the live video produced by video camera 30 while DVR32.The example of this kind of system is safety or surveillance.Can the function of like that remotely control HD video camera 30 as described in more detail below.HD video camera 30 is suitable for producing high-definition signal 332 and simulation CVBS signal 330 simultaneously.In certain embodiments, high-definition signal 332 and simulation CVBS signal 330 are when waiting, if but such as unequal in the time delay of process unlike signal, then they may be when roughly waiting.In one example, due to number-mode convertion expense, CVBS signal 330 can be time delay.In another example, compressible high-definition signal 332 make it experience different time delays based on compression ratio etc.In certain embodiments, CVBS330 and high-definition signal 332 can be synchronous with the common audio signal produced by video camera 30 or keep with it constant time relationship.

By adding external component or hardware and software being integrated into video camera 30 and adaptive video camera 30.In this example, safety chain modulator-demodulator (SLOC-T) on coaxial cable is provided in video camera 30.SLOC-T31 can be configured to the modulator-demodulator integrating or use the parts being integrated into video camera 30 to realize as the additives of video camera 30.SLOC-T30 allows multimedia feeds over the communication channels to downstream transmission: as shown in the figure, SLOC-T31 is a kind of equipment, this equipment allows to transmit multiple signal on coaxial cable 33, described multiple signaling bearer to characterize by video camera 30 produce the different resolution signal of video.In order to clearly demonstrate, the SLOC be deployed in transmitting apparatus (such as video camera 30) is called SLOC-T herein, and the SLOC be arranged in receiving equipment (such as DVR, the network switch etc.) is called SLOC-R.To provide in more detail hereinafter the description of SLOC-T and SLOC-R equipment.

SLOC-T31 can cooperate with other parts of video camera 30 and/or can increase the enhancing function that video camera 30 is worked in each mode.In one example, video camera 30 can produce unpressed HD digital video and to export and SLOC-T31 can provide the ability of compression HD digital video signal.Therefore, SLOC-T31 can provide the ability beyond modulation and demodulation to strengthen the function of main frame video camera 30 as required.Therefore, several SLOC-T equipment can work in each mode, and some of them pattern is provided by example.In a kind of pattern, SLOC-T31 receives the SD analog version of compressed HD vision signal and signal from video camera 30 and transmits this two kinds of signals at coaxial cable 33.In another kind of pattern, SLOC-T31 receives the HD vision signal of uncompressed with the SD analog version of signal from video camera 30 and transmit compressed HD digital signal version together with SD analog signal on coaxial cable 33.The SD analog signal that SLOC-T31 can send HD digital signal and derive from the HD signal that video camera 30 receives.

In certain embodiments, SLOC-T31 uses frequency division multiplexing to produce the output signal of transmission on coaxial cable 33.In the example that Fig. 5 explains orally, with frequency f _cdcarrier wave 53 centered by single frequency range 52 in downstream digital signal is provided.Frequency range 52 is from the highest frequency f exceeding base-band analog signal 50 ₀position start.This different frequency range 52 can be called as channel.Channel 52 can be selected based on the ability of SLOC-T31, available bandwidth, signal bandwidth and other reason.In certain embodiments, can according to receive the compatibility of equipping and carry out selective channel 52.In one example, signal directly can be supplied to Standard Definition Television and can selective channel 52 to guarantee the suitable isolation with baseband signal.When using the standard definition coding of signal, the multiple frequency ranges in selective channel 52 also can be carried out based on the standard of digital video transmission.Individual digit signal is transmitted it is conceivable that all parts by using two or more different channels to carry digital signal.

Any suitable modulation scheme can be used to produce the transmitted version of digital signal.Such as, dissimilar wired and wireless connections can with the coupling of more modulation scheme, such as phase shift keying (PSK), frequency shift keying (FSK), quadrature amplitude modulation (QAM), OFDM (OFDM) etc.Modulation scheme normally based on some selecting factors, these factors comprise for the characteristic of the medium transmitted, the vision signal of requirement frame rate and affect the other factors of the available bandwidth in channel 52.

SLOC-R modulator-demodulator 35 can be arranged in the video capture device of such as DVR32.SLOC-R modulator-demodulator 35 can receive and fast acquisition of digital video and CBS signal.Typically, CVBS signal is extracted and is directly passed to display system 33 to watch the video image of being caught by video camera 30 in real time.Display system 33 can be SD monitor, although display system also can receive the digital version of received analog signal.In one example, SLOC-R modulator-demodulator 35 digitized version that can produce analog signal with data monitor or the computer coupling that is suitably equipped with.The extraction of baseband signal typically uses that low pass filter realizes, and this low pass filter can be used analog component or be realized by Digital Signal Processing.Digital HD signal can be extracted separately and provide it to the recording portion of DVR32.In certain embodiments, can compressed digital HD vision signal in DVR before recording.In many examples, digital HD video signal is received as compressed digital signal.

In certain embodiments, SLOC-T31 and SLOC-R35 is configured to the transmitted in both directions of supporting signal.In the example of safety device, and as described in detail referring to Fig. 6, video camera 30 can comprise microphone 614, loud speaker 612, transducer 616, for the control interface 618 of controller electric brake and further feature (see Fig. 6).In this example, SLOC-T31 and SLOC-R35 is configured to control, audio frequency and other data 36 to convey to video camera 30 usually.

Refer again to Fig. 5, in one embodiment, at the one or more channels 54 being arranged in available bandwidth upper end, upstream data can be conveyed to video camera.For digital multimedia signal 52, to control and the Channel assignment of audio signal 54 and other data communication can be selected based on the signal to noise ratio recorded in available bandwidth, channel 52 and 54, signaling standard and/or specialized requirement.In certain embodiments, once use training sequence to establish the connection of SLOC-T31 and SLOC-R35, channel configuration, bandwidth sum signal to noise ratio is namely determined.Typically, training sequence is used to determine signalling capability that is predetermined or negotiated channel, to select the channel 52 for transmission digital video and to determine the available bandwidth in selected channel 52.The characteristic of selected channel 52 can be used for the compression level setting digital video signal.

In certain embodiments, stream signal 54 comprises the signal of the content that can control downstream 52 and base band 50 signal.Such as, camera optics device 600 can provide the flake visual field of the position monitored by video camera 60 and can control camera processes device to select for a part of image of transmitting as baseband signal 50.Typically, downstream digital signal 52 can provide the complete image being recorded on DVR or supplying process further.Baseband signal 50 can be in the region under supervising with real time monitoring by receiving baseband signal 50.Baseband signal 50 can comprise through adjustment image, this visual effect formed by fish eye lens through adjustment image rectification.The observer of baseband signal 50 carries out observing by selecting a new part through catching image and the visual field is moved in the fish-eye ken.Such as, observer can ask " to right translation " to move right to make the ken.In stream signal 54, the data of transmission then make camera processes device extract and process the requirement part of the ken.In certain embodiments, make the request of the ken movement be included in baseband signal 50 that the physics of video camera 60 can be caused to move.Therefore, the control data in stream signal 54 may affect the content of base band 50 and downstream digital 52 signal.

In certain embodiments, downstream audio frequency can be used as the part of HD digital video signal and/or the fractional transmission as CVBS signal.Some downstream signalings can be carried in independent dedicated channel (not shown).In certain embodiments, out-of-band communication method can be used realize with the upstream communication of video camera 30, comprise and such as use wired or wireless network.In certain embodiments, it is conceivable that wirelessly sending downstream digital signal 52 as an alternative or additional to select.Therefore, while some combinations wirelessly transmitting upstream 54 and downstream 52, by co-axial cables transport baseband signal 50.Typically, upstream data 54 comprises the control signal of downstream 52 and base band 50, and no matter transmission method why.

In certain embodiments, cable 33 directly can be supplied to display system 33 with display simulation SD video.SD monitor or display 33 generally include the filter circuit allowing to select between baseband signal and the TV channel of classical modulation.As a result, the carrier signal of the discardable high-frequency digital coding of monitor 33.If use the digital encoding scheme transmission digital video signal of standard definition in the channel of standard definition, then DVR32 also can receive digital video signal and without the need to extra process.SLOC-R35 decodes the signal that produced by SLOC-T31 be supplied to DVR32 by through the HD digital video of decoding and other signal.SLOC-R35 also can paired domination number certificate, voice data and other data carry out encoding for transmitting to video camera 30.

Referring now to Fig. 4, provide embodiments of the invention, this embodiment explains orally some operation principle of the present invention.Fig. 4 describes an example, and this example is based on the system being provided HD video version while requiring to watch the live video that produced by video camera 40 by the network switch 44 on network.In one example, HD video feed is captured and uses inside or external IP video server to carry out stream and send.HD video camera 40 is suitable for producing high-definition signal and Analog Baseband vision signal usually simultaneously.By adding external component or hardware and software being integrated into the adaptive video camera 40 of video camera 40, such as SLOC-T400.SLOC-T400 can be identical with the SLOC-T31 shown in Fig. 3 mode operate.But the mode that SLOC-T400 can be configured to be beneficial to forward digital video signal on network is to digital encoding video signal.Such as, the stream that SLOC-T400 can support according to IP video server send form programme or to be configured to separately provide digital video signal.

The multiplex video signal sent by digital camera 40 can be received by the network switch 44, and the described network switch 44 is furnished with SLOC-R440 alternatively.Base band SD analog signal can be extracted and provide it to display 43.In certain embodiments, SLOC-R440 can extract digital high-definition video signal and use the suitable networks of enough bandwidth with carrying digital HD video signal to forward it to video server or other network equipment.Digital HD video signal can comprise compressed HD vision signal.In certain embodiments, the digital high-definition signal extracted by SLOC-R440 is compressed or compresses further to be forwarded to video server or other network equipment.SLOC-R440 can comprise for record and/or again modulation digital high-definition signal with the hardware and software in transmission over networks; Such as SLOC-R440 can produce the H-264 signal of coding to transmit over ethernet.

Referring now to Fig. 6, some embodiment of the present invention provides the enhancing ability being applicable to safety system.In described example, video camera 60 comprises modulator-demodulator SLOC-T606 and processor, and processor is configured and is adapted to some aspect according to the present invention and provides through digitally coded multi-media signal.The combination of optics 600 and imageing sensor 602 can be used to catch image sequence, and this combination can comprise the combination of the known lens combination of those skilled in that art and ccd sensor.Processor 604 receives sweep signal 603 from imageing sensor 602 usually, the image sequence that described imageing sensor 602 provides as requested or predefine frame rate is caught.

In certain embodiments, imageing sensor 602 can comprise the carrying out changing through scanning analog signal and producing hardware and the logic of digital video signal of the image of sign being caught by one or more transducer.Such as, imageing sensor 602 can comprise RGB (red, green, blue) transducer and imageing sensor 602 can inter-process RGB transducer export using produce through digitally coded colour-video signal as its export 603.In other embodiments, processor 604 can preliminary treatment from the signal 603 of imageing sensor 602 to obtain raw digital video signal.No matter obtain from inside or receive from imageing sensor 602, raw digital video all can be for further processing to obtain initial HD digital video signal by processor 604.Simulation standard definition signal obtains by the output 603 of process raw digital video signal, transducer 602 or initial HD digital video signal.Then processor 604 can format to obtain to initial HD digital video signal the one or more HD digital video signals meeting broadcast and other standard.Such as, processor 604 can produce the signal of the broadcasting video standard meeting such as ATSC and DVB standard.Processor 604 can additionally compressed digital video signals.

Camera processes device 604 can comprise the combination of commercially available parts and custom hardware and software.In one example, processor can comprise microprocessor, digital signal processor, microcontroller, sequencer and other and memory combination and support to perform series of steps, instruction and/or program logic programming device in one or more.Storing 610 to can be used to store computer-readable instruction, when performing this instruction, performing some or all functions described in the application.Camera processes device 604 can comprise some built-in or " hard coded " processes, and these processes can be used to build some embodiment of the present invention.Store 610 also can be used to buffer programming and/or maintain configuration information.In certain embodiments, store 610 to can be used for storing the videograph of being caught by video camera 60.Therefore, storing 610 can use volatile and nonvolatile storage, CD and disk, removable electricity erasable memorizer, USB memory driver and other semiconductor, electromagnetism and light storage device to realize.

Signal 605 comprises and is supplied to the vision signal of SLOC-T606 by processor 604 and receives from circuit 62 and the upstream being transmitted to processor 604 by SLOC-T606 controls, audio frequency and other upstream information.Can decode by processor 604 pairs of upstream audio-frequency informations, process and/or format before by voice frequency relay to loud speaker, transducer or other audio output system 612.Processor can amplify audio signal or can adopt independent amplifier in audio output part 612.Upstream control can comprise optics control 601 and control the signal of external equipment, and this signal is provided by control interface 618 usually.External equipment can comprise for making video camera 60 translation, rotation or with the motor of orientation separately or actuator.Optics control signal 601 and external control signal 618 can produce in response to by the order of remote control system predefine.Such as, remote subscriber can handle a rocking bar, this rocking bar produces a series of coded commands explained by camera processes device 604, thus represent " in horizontal plane deasil rotary camera 90 degree ", and processor 604 responds by a series of pulse being sent to the stepping motor axially installed relative to video camera 60, this series of pulses makes video camera 60 carry out the rotation expected around its vertical axis thus.The focus of similar order adjustable optical device 600, convergent-divergent and aperture.

In another example, can provide instruction and data in the control information of upstream, this upstream control information can be used for the function of control processor 604 and/or transducer 602.This instruction and data can be used for selecting a certain region in video camera 60 ken to encode in one or more downstream visual signal.In certain embodiments, processor and transducer cooperation are to provide one or more virtual video camera, these virtual video cameras can be remotely steerable to specify in the ken will by the part of encoding, and selects these parts thus by the empty translation that operates in the practicality visual that can determine at the optics by video camera 60, convergent-divergent and tilt function.In certain embodiments, processor 604 can make video camera physics move extraly, expands the scope of translation, inclination and zoom function thus.

In at least some embodiments, conception makes CVBS and digital signal carry the part of the image of being caught by imageing sensor 602 separately.These image sections can be overlapping or carry out the zones of different that formed in the comfortable ken provided by camera lens 600.In addition, in certain embodiments, additional cameras 60 and/or additional image sensor 602 can be used to expand the available ken.Such as, the multiple video camera of desired configuration is to obtain panorama (360 °) view in region.One or more processor 604 can provide the analog and digital signal representing view or a part of view.In one embodiment, can provide complete panoramic view in the digital signal being recorded to DVR, CVBS signal can provide selectable view in panorama simultaneously.Selectable view can use convergent-divergent, translation and other control to be controlled.In another example, CVBS and digital signal can provide the public of panoramic view or different piece and these parts independently can be controlled by remote observers.

Fig. 7 illustrates the example using the SLOC-R700 similar with the SLOC-R35 that describes in Fig. 3 in secure digital video recording system 70.System 70 comprises SLOC-R700, is connected to the DVR processor 702 of ancillary equipment 710,712 and 714, analog video decoder device 704, digital video decoder 708 and HD digital display processor 706.As mentioned above, SLOC-R700 receives the decode the signal from coaxial cable 72, and this signal generally includes simulation SD vision signal and HD digital video signal.SLOC-R700 also transmits upstream audio frequency and control signal by coaxial cable 72.SLOC-R usually will simulate CVBS signal and separate with HD digital video signal in input signal 72, digital video signal 703 is supplied to processor 702 and CVBS signal 701 is supplied to SD monitor 74 as the real-time feeding from the video camera 60 shown in Fig. 6.Analog Baseband vision signal 701 is supplied to analog video decoder device 704 by SLOC-R700 alternatively, and this analog video decoder device 704 processing signals is to produce digital SD vision signal 705.The multiplexed digital standard definition signal 705 of video-stream processor 706 and the signal 707 of deriving from stored HD digital video playback also make one's options between these two kinds of signals.Video-stream processor can provide by signal selected by HD TV or the displayable form of monitor 76.

DVR processor 702 receive digital HD video signal 703 and alternatively storage signal at least partially as the recording of the video of being caught by video camera 60.Recording can be stored in local hard drive 714, is stored on networking memory (not shown) or is stored in other optical, electrical magnetic or semiconductor storage of being connected by network interface 710 and/or USB/ live wire or other local bus 712.The video recorded can compress to save memory space further.DVR processor can use digital video decoder 708 retrieve the video of recording and provide playback signal 707.

Fig. 8 illustrates the example using the SLOC-R800 similar with SLOC-R440 shown in Fig. 4 in networking safety means 80.Equipment 80 is comprised SLOC-R800 and is usually connected to the network exchange processor 802 of IP video server 86 by network.As mentioned above, SLOC-R800 receives the decode the signal from coaxial cable 82, and this signal generally includes simulation SD vision signal and HD digital video signal.SLOC-R800 transmits upstream audio frequency and control signal by coaxial cable 82 alternatively.Simulation CVBS signal is separated with HD digital video signal by SLOC-R usually in input signal 82, digital video signal 803 is supplied to processor 802 and CVBS signal 801 is supplied to SD monitor 84 as the real-time feeding from video camera 60 shown in Fig. 6.In certain embodiments, SLOC-R80 can comprise parts 804,806 etc., with digitlization CVBS signal 801 with the digital display coupling with such as high-clear display 85, same as the real-time feeding from the video camera 60 shown in Fig. 6.But will understand, the display device of suitably configuration or computing equipment can receive the digitlization of CVBS signal 801 executive signal.Network exchange processor 802 receives digital HD video signal 803 and transmits signals to network video server 86 alternatively, and then this network video server 86 can maintain the recording of the video of being caught by video camera 60.Digital HD video signal 803 can be further compressed before transferring to video server 86.

Refer again to Fig. 5 and Fig. 6, some embodiment of the present invention allows the content selecting base-band analog signal 50 and downstream signal 52 as required.In one example, both baseband signal 50 and downstream signal 52 all comprise same image, and the former is analog form and the latter is through digitally coded.Digital picture can optionally and selectively with compression and uncompressed form, with SD and high definition and with the frame rate transmission of full motion or reduction.In another example, baseband signal 50 provides a part for the complete image of being caught by imageing sensor 602, and downstream signal 52 carries complete image simultaneously.In another example, baseband signal 50 provides the complete image provided by imageing sensor, and downstream comprises a part for complete image simultaneously.As a result, visualize a kind of system of fine configuration, its allow digital camera user from for show, record and transmit video image wide region option make one's options.

the analog balanced of baseband signal

Some embodiment of the present invention comprises the system and method for improving the high frequency shuffle effect in cable, and aforementioned high frequency shuffle effect can cause higher frequency decay along with the increase of cable length.By cable introduce this crooked make baseband analog video and passband digital video signal deterioration, this deterioration along with cable length increase and further serious.But some embodiment of the present invention provides equalizer usually in digital receiver, it is crooked that this equalizer is removed on digital passband signal, allows the reliable decoding sending code element.

Some embodiment of the present invention improves and comprises the system of those systems aforementioned and the performance of device, wherein baseband video signal can combine with the digital representation of baseband video signal and control signal, allows thus to transmit on the unit cable of such as coaxial cable (" coaxially ").Fig. 3 and Fig. 4 illustrates the embodiment providing SLOC system, and Fig. 5 illustrates a kind of feasible modulation scheme of SLOC system.For Fig. 3, HD video camera 30 providing package containing the output 332 of compressed digital HD video and the auxiliary camera output 330 comprising simulation SD (SD) CVBS.Compressed HD vision signal 332 utilizes SLOC camera side modulator-demodulator 31 to be modulated to passband 52, this SLOC camera side modulator-demodulator 31 comprises QAM modulation device, this QAM modulation device provides modulated signal, and this signal and baseband analog CVBS signal 330 combine.Signal through combination transmits to downstream on coaxial cable 33, usually reaches 300 meters or longer distance.In monitor side, the signal representing base-band CVBS signal 330 is isolated by SLOC monitor side modulator-demodulator 35 from the signal representing passband downstream visual signal 332.The signal characterizing CVBS is fed to SD display 34 for watching in real time without time delay.The qam demodulator demodulation of upper passband downstream signal, the output of qam demodulator is fed to host-processor and DVR32, DVR32 and supports that real-time (although may slightly time delay) HD viewing on monitor 34 and non real-time HD playback are for watching in the future.

In this example, according to such as by IP agreement need upstream communication is provided.Upstream communication can be used extraly so that audio frequency and camera control signal 334 are sent to video camera 30 from monitor side.As a rule, the bit rate of stream signal---and therefore required bandwidth---is usually far below the bit rate needed for the passband signal of downstream and bandwidth.Monitor side SLOC modulator-demodulator 35 comprises QAM modulation device, this QAM modulation device by IP signal madulation to upstream passband 54.As depicted in figures 5, upstream passband 54 and downstream passband 52 are positioned at different spectral position.In camera side, SLOC modulator-demodulator 31 comprises the qam demodulator for receiving stream signal.The method provides the some advantages being better than existing system and method, comprising:

(1) working range-increase distance is increased.

(2) existing infrastructure can be used and re-use coaxial cable and carry out deployment system.

(3) low delay availability, in real time (fact) video.

(4) live CVBS video and HD video can watched dividually.

Figure 21 is the rough schematic view of the additional detail of the SLOC camera side modulator-demodulator 49 that Fig. 4 is shown.IP connection to HD video camera 2100 is interfaced to QAM modulation device 212 and qam demodulator 214 by Media Independent Interface (MII) module 210.In one example, MII210 follows IEEE802.3 standard.QAM modulation device 212 uses principles well-known work to convert baseband I P data flow 2100 to passband QAM code element 2120.These code elements and base-band CVBS signal 2160 are sued for peace 216 and is fed to duplexer 218 subsequently.Duplexer 218 can be bidirectional analog equipment, and the base band of combination and lower passband downstream signal 2162 are passed to coaxial cable and receive upper passband stream signal 2140 from coaxial cable and be fed to qam demodulator 214 by it.The upper passband stream signal 2140 that qam demodulator 214 usually uses principles well-known work to receive from monitor side joint with demodulation also exports base band data to MII interface 210.

Figure 22 is the rough schematic view of the additional detail of the SLOC monitor side modulator-demodulator 45 that Fig. 4 is shown.Duplexer 220 receives the base band CVBS of downstream combination and lower passband IP signal 2200 and by low pass (LP) and high pass (HP) filtering by signal segmentation component amount element 2201-2203 from coaxial cable.CVBS signal 2201 can be transmitted directly onto SD monitor or other display device.Lower passband signal 220 can be fed to qam demodulator 222, and this qam demodulator 222 is fed to MII interface module 226.Duplexer also can receive upper passband signal 2203 from QAM modulation device 224 and this stream signal can be passed to coaxial cable.QAM modulation device 222 obtains its input from MII interface 226 usually, and MII interface 226 can be connected to the main frame/DVR supporting IP agreement.

Coaxial cable shows significant high frequency slip characteristics usually, and this high frequency slip characteristics can cause higher frequency decay along with cable length increase.This " crooked " is significant and may causes significant intersymbol interference (ISI) in the frequency range of passband signal.Digital equalising may be needed correctly to recover to allow qam demodulator 222 data transmitted.

Base band is modulated to passband

Figure 23 more specifically illustrates that camera side base band is to passband QAM modulation device 212 (Figure 21).Data from MII210 are received by FEC encoder/mapper 2300, and this FEC encoder/mapper 2300 uses the Reed-Solomon of such as cascade coding, byte-interleaved and/or trellis coding error protection data to be added into the data flow received from MII210.Data multiplex is resolved into stream 2300 and 2302 by mapper/encoder 2300, and wherein the bit group of sizing of giving of each stream represents QAM symbol amplitude level respectively at real axis and imaginary axis direction.Transmission QAM pulse through isolation is provided by following formula:

$s_m\left(t\right)=d_{R,m}q\left(t\right)\cos \left(2\mathrm{\π}{f}_{c}t\right)-d_{I,m}q\left(t\right)\sin \left(2\mathrm{\π}{f}_{c}t\right)=\mathrm{Re}\left\{d_{m}q\left(t\right)e^{j2\mathrm{\π}{f}_{c}t}\right\},$

Wherein d _{r, m}and d _{i, m}be determined by two independent message streams and represent real part and the imaginary part of multiple QAM code element respectively, wherein m=1...M is as the index of two-dimentional QAM radix constellation, and wherein M is modulated carrier frequencies and q (t) is square-root raised-cosine impulse function.

Transmission QAM pulse s (t) of one continuous series is with speed F _s=1/T _sby being with the multipath channel of noise.Therefore, provided by r (t)=s (t) * c (t)+v (t) at the Received signal strength of the input side to QAM receiver, wherein * represents convolution, and c (t) is channel impulse response, and v (t) is additive white Gaussian noise.Therefore:

$r\left(t\right)=\mathrm{Re}\left\{e^{j2\mathrm{\π}(f_{\mathrm{LO}}+f_o)t+{\mathrm{\θ}}_o}\underset{n=-\∞}{\overset{+\∞}{\mathrm{\Σ}}}\right[d\left[n\right]*q\left(t\right)\left]c(t-n{T}_s)\right\}+v\left(t\right),$

Wherein d [n] is multiple transmission code element, f ₀and θ ₀receiver pass-band respectively to base band demodulator local oscillator relative to the frequency shift (FS) of transmitter and phase deviation, thus f _lO=f _c-f ₀.

Passband is to base band demodulator

Figure 24 A illustrates in greater detail monitor side passband to base band qam demodulator 222 (Figure 22).Signal r (t) can receive from coaxial cable, such as, with the polydispersity index (see 240) higher than chip rate, obtains the signal r (nT through sampling thus _samp).After sampling:

$r\left(n{T}_{\mathrm{samp}}\right)=\mathrm{Re}\left\{e^{j2\mathrm{\π}(f_{\mathrm{LO}}+f_o)n{T}_{\mathrm{samp}}+{\mathrm{\θ}}_o}\underset{m=-\∞}{\overset{+\∞}{\mathrm{\Σ}}}\right[d\left[m\right]*q\left(n{T}_{\mathrm{samp}}\right)\left]c({\mathrm{nT}}_{\mathrm{samp}}-m{T}_s)\right\}+v\left(n{T}_{\mathrm{samp}}\right).$

Then, after down conversion, with chip rate 1/T _sagain sampling and matched filtering obtain:

$x\left({\mathrm{kT}}_s\right)=x\left[k\right]=e^{j2\mathrm{\π}{f}_{o}k{T}_s+{\mathrm{\θ}}_o}\underset{m=-\∞}{\overset{+\∞}{\mathrm{\Σ}}}d\left[m\right]c[k-m]+v^{\′}\left[k\right],$

Wherein v ' [k] is the multiple filter noise through sampling, supposes that any ISI is attributable simply to channel impulse response c, because pulse shaping and matched filtering q and perfect chip rate sampling timing combine.

Equalizer and carrier phase/frequency ring

Digital equalizer and the carrier phase/frequency loop of Figure 24 A is discussed in more detail referring to Figure 25.Signal x [k] enters adaptive digital equalizer 250, and this adaptive digital equalizer 250 can comprise for compensating the crooked linear digital filter caused by channel impulse response c.One or more known methods comprising LMS algorithm can be used to realize tap-weights adjustment.Equalizer is exported y [k] and is adjudicated with two dimension (2D) sheer the error signal of making comparisons the renewal collection created for calculating filter tap-weights through phase rotating version.LMS algorithm can operate as follows:

Order: x [k] represents the equalizer input vector of N length, and

Y [k] represents equalizer output vector g ^h[k] x [k],

Wherein g ^h[k] is the equalizer tap weights vector of N length and subscript H represents that conjugation exchanges (close conjugation in distress).

$e\left[k\right]=\hat{d}\left[k\right]-y\left[k\right]$

g[k+1]＝g[k]-2μx[k]e ^*[k]，

Wherein μ is little step parameter and subscript * represents complex conjugate.

In order to eliminate the crooked impact of passband cable, after convergence, LMS equalizer tap can be similar to the inverse of channel impulse response c.

2-D sheer 252 real part of cutting z [k] and imaginary part export independently it is the estimation of the d [k] of original transmission.Phase error detection module 258 receive z [k] and and form phase error signal low pass (LP) filter 256 can be integration scale filter, and this integration scale filter allows the skew of loop phase calibration and frequency shift (FS).The output of low pass filter 256 is fed to subdivision and founds voltage controlled oscillator (VCO) 254, this voltage controlled oscillator 254 output calibration θ _oand f _oboth complex phase position/frequency correction factor e ^{-j θ [k]}.VCO254 also provides output (e ^{+ j θ [k]}), this output " does not correct " section and exports can be used for make it deriving the error signal upgraded for equalizer tap.Because equalizer is operated in x [k], therefore this normally needs.See also Figure 24 A, equalizer exports z [k] and is fed to symbol de-mapper, and the real number detected is become bit group with imaginary number level translation by this symbol de-mapper.Then fec decoder device performs Veterbi decoding, and byte deinterleaves and/or Reed-Solomon decodes to correct the error of reception bit and the data obtained is delivered to MII interface.

The impact of cable length

The vision signal received can experience decay because of the frequency become in some feature being attributable to cable.For discussion object, describe the example of coaxial cable.The order of severity of decay---is often referred to as crooked---usually to depend on type of cable and cable length.Figure 26 A and 26B illustrates because becoming in the decay for the type of cable RG6 of various length and the frequency of RG59.Can find out, crookedly be equivalent to multipath distortion, wherein additional path and main path have minimum delay expansion.Along with crooked increase, number and the respective gain thereof of important multipath component also increase.Multipath distortion causes ISI in Received signal strength and therefore may seriously degrading transmission reliability.In a digital signal, equalizer can be used in receivers to remove these infringements.Figure 27 A and 27B illustrates the amplitude response of the equalizer tap of the power spectral density (PSD) that equalizer inputs and convergence respectively.Specifically, Figure 27 A illustrates the PSD (band connection frequency and relative base band frequency are shown) that the equalizer after the RG6 cable by 2000 feet transmits under 15.98MHz carrier frequency inputs, and Figure 27 B illustrates the amplitude response of the digital equalizer tap of convergence.

Some embodiment of the present invention comprises digital equalizer, and this digital equalizer can eliminate the reliable decoding that crooked, the ISI removed in passband signal that are introduced by cable also realize transmitting data.Along with cable length increases, digital equalizer and known numerical data forward error guard method (the Reed-Solomon coding of such as cascade and trellis coding) can be used reliably to be received in the digital passband signal of monitor side.But the crooked high-frequency that also adversely can have influence on baseband analog CVBS signal of cable, this can be reduced in the acutance of the picture that monitor side is watched and the intensity of color.Therefore, some embodiment provides sef-adapting filter, such as analog equalizer, and this sef-adapting filter can be applied to CVBS signal in monitor side crooked with the cable compensating base band place.Some embodiment utilizes passband digital equalizer to estimate the crooked amount at base band place, and select subsequently suitable in one group of baseband analog filter one to be applied to received CVBS signal.

The efficient estimation that passband is crooked

During crooked in estimating signal frequency range, that can select in the PSD of the input signal when quantizing with dB is crooked by close to linear frequency range.Therefore, in the input of base-band digital equalizer, therefore the frequency of-2.67MHz to 2.67MHz will correspond to the frequency of 13.31MHz and 18.65MHz in passband input signal, provides a suitable scope.As shown in fig. 26, be about 3.7dB from 13.31MHz to 18.65MHz to the crooked of the RG-6 of 2000 feet.In order to estimate crooked in dB of the digital equalizer filter tap from convergence, lower column count can be performed:

${\widehat{\mathrm{\Δ}}}_{\mathrm{dB}}=G_{\mathrm{dB}}\left[k_1\right]-G_{\mathrm{dB}}\left[k_2\right]=10{\log }_{10}\left(\frac{{\left|G\right[k_1\left]\right|}^2}{{\left|G\right[k_2\left]\right|}^2}\right),$ (equation 1)

Wherein G [k] is the DFT of the equalizer filter tap of time domain convergence, and k ₁and k ₂corresponding to the certain tones gap of DFT.Because the digital equalising of Figure 25 realizes by convolution, therefore in order to estimate given k ₁and k ₂crooked, usually need FFT (maybe may do N time to two points take advantage of again and be added with).That is,

$G\left[k_1\right]=G_R\left[k_1\right]+i{G}_I\left[k_1\right]=\underset{n=0}{\overset{N-1}{\mathrm{\Σ}}}g\left[n\right]e^{-i2\mathrm{\πn}{k}_1/N},$

(equation 2)

Wherein g (n)=g _r(n)+ig _in (), n=0,1...N-1 are N time-domain equalizer tap (being left in the basket with the interdependence of time index).Note, 1/N scalar is unwanted in this calculates.Can for G (k ₂) perform similar calculating.But, significantly reduce calculating by selecting gap frequently carefully.By making k ₁=N/4, this corresponds to the frequency of 2.67MHz, and the complex exponential in equation (2) is significantly simplified:

(equation 3)

The real part of filter freguency response and imaginary part can use read group total to go out:

$G_R\left[k_1\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R\left[4n\right]+\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+2]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+3].$

(equation 4)

$G_I\left[k_1\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I\left[4n\right]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+2]+\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+3].$

(equation 5)

Finally, the power under this frequency gap is:

${\left|G\right[k_1\left]\right|}^2=G_R^2\left[k_1\right]+G_I^2\left[k_1\right].$ (equation 6)

By making k ₁=N/4, power calculation is significantly simplified.Similarly, if k ₁=3N/4, it corresponds to the frequency of-2.67MHz, then complex exponential is significantly simplified again.

(equation 7)

Real part and imaginary part are calculated as:

$G_R\left[k_2\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R\left[4n\right]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+2]+\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+3],$ (equation 8)

$G_I\left[k2\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I\left[4n\right]+\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_I[4n+2]-\underset{n=0}{\overset{N/4-1}{\mathrm{\Σ}}}{g}_R[4n+3],$ (equation 9)

And rated output as described above | G [k ₁] | ².In figure 2b, the optimum SNR evenly approximately linear of upwards crooked (in units of dB) in the amplitude response of the filter tap of convergence and 64QAM signal and tap noise.In addition, when calculating in this way, this crooked 3.7dB of reality closely in this frequency range.

To the crooked estimation of the passband of base band CVBS skew corrected

After crooked for this digital video signal estimation passband, the baseband analog filter be suitable for can be selected from M different filter.Can find out, the passband estimated of digital video signal frequency range crooked will be crooked in instruction base-band CVBS signal the order of severity, then available analog filter corrects that this is crooked roughly.In Figure 28 A, what illustrate in those electricity for RG-6, RG-11, RG-59 and RG-174 and similar length digital video signal frequency range from 13.31MHz to 18.65MHz is crooked.Figure 28 A illustrates for RG-6, RG-11, RG-59 and RG-174 type of cable relative to the loss under the crooked 3.58MHz in passband digital video signal.Figure 28 B illustrates the loss under 6MHz.Can observe, the loss under 3.58MHz and 6MHz is roughly the same for given crooked all four kinds of type of cables.Figure 29 A illustrates for RG-6, RG-11, RG-59 and RG-174 type of cable relative to the loss under the crooked 3.58MHz in passband digital video signal.Figure 29 B illustrates the loss under 6MHz.Can observe, the loss under 3.58MHz and 6MHz is roughly the same for given crooked all four kinds of type of cables.

It is the unique available information about cable frequencies response because the passband estimated is crooked, therefore desirable scene is crooked relevant in a known way with passband digital signal of the frequency response of cable under base band (CVBS signal frequency range), and no matter how are type of cable or length.Figure 28 B, 29A and 29B confirm this situation in the frequency response under DC, 3.58MHz and 6MHz.Such as, the crooked place of the 1.5dB in passband digital video signal, is approximately 0.68dB, 4.1dB and 5.3dB respectively for all four kinds of cable losses under DC, the loss under color carrier (3.58MHz) and the loss under 6MHz.Therefore, no matter be cause the passband of 1.5dB crooked from the RG-11 of the RG-174 of 275 feet, the RG-59 of 750 feet, 825 feet RG-6 or 1825 foot, same analog filter is by crooked for the base band eliminating CVBS signal.

For selecting an example of the algorithm of suitable analog filter as follows from one group of M filter:

Note α ₀=1; α _nother value < 1 and select them to be enough to calculate R to make displacement addition _n.Therefore, the monitor side qam demodulator of change Figure 24 A provides a signal to make the digital equalizer of passband qam demodulator usually, and this signal behavior M simulates one in CVBS filter response.Figure 24 B illustrates the change part of monitor side qam demodulator, and wherein analog filter is selected to export from according to the digital equalizer of aforementioned algorism work.Figure 30 illustrates complete monitor side modulator-demodulator, and filter selects signal 305 to be supplied to CVBS analog equalizer 302 by the digital equalizer in qam demodulator 304.

The example being suitable for the Active Analog Filter of equalization base band CVBS signal is illustrated in Figure 31.In this example, M=3, the filtering that existence 4 kinds is possible is thus selected.The each RC be attached thereto by the switch of the M+1 in Closing Switch module 310 making thus selects to close the filter response needed to ground connection successively.Filter response possible shown in Figure 32.

Those skilled in that art will understand, and the present invention can be applicable to the digital communication system adopting the modulation of other passband and forward error correction.Those skilled in that art also will recognize, plural point in the FFT of passband digital equalizer tap-weights vector g [n] can be used to come for CVBS signal behavior analog filter, and other type numeral equalizer design that can adopt passband signal, comprises frequency-domain equalizer, wherein G ₁[k] and G ₂the value of [k] calculates as a part for equalization process.In addition, the known equalizer tap weights computational methods except LMS can be adopted, such as RLS.

In certain embodiments, have and the CVBS analog filter of Response to selection can adopt form other than the above.Equally, the equalizer of CVBS signal can adopt the form of digital filter, samples and digitlization before the equalization in this case to CVBS.In this case, according to describing for selecting the identical algorithms of in M analog filter response to concentrate from the predetermined of M tap-weights vector the tap-weights selecting digital filter.

framing in digital communication system

Digit data stream has the frame structure of some types usually, with the bit group or the byte group that make these data be organized into homogeneous size.Any system of block-based forward error correction (FEC) is used to have the frame be organized in about error correcting code word length.Equally, if system uses intertexture to resist impulsive noise, then consideration interleaver parameter is configured by frame structure.If the randomization of system usage data obtains Flat Spectra, then utilized pseudo random sequence can be synchronous with frame structure, restarts when every frame starts.

For RF digital communication system, receiver usually must first obtain carrier wave and chip clock is synchronous and balanced.Then just transmitted data can be recovered.But thisly import data flow into understand, receiver must be also synchronous with this frame structure.In other words, receiver must know that error correction code word terminates wherein from where.The receiver module of such as deinterleaver also must be able to be made synchronously to mate the interleaver operation of transmitter, the bit deinterleaved that obtains or byte is made correctly to be sorted thus, and remove randomizer, become smooth with the starting point of mating for the pseudo random sequence in transmitter to make frequency spectrum.

Legacy system is attached to the head of frame or afterbody frequently by the known pattern of the code element by regular length and provides receiver frame synchronous.Repeatedly, and it is often made up of 2 grades of (i.e. binary system) pseudo random sequences with favourable auto correlation characteristic on this identical patterns every frame ground.Although this means that sequence and itself auto-correlation under zero offset produce a higher value, if skew is non-zero, this autocorrelation value (secondary lobe) is very little.In addition, this frame synchronization sequence and random code element is relevant by generation one smaller value.Therefore, if the receiver storage version of frame synchronization pattern performs the association of input symbols, then should expect only to produce a higher value in the accurate starting position of each frame.Then receiver easily can determine the initial point of each frame.

The example of frame structure

See Fig. 9, the system that ATSC Digital Television (DTV) ground transmission standard adopted for 1996 provides data to transmit with frame.Each frame 90 comprises 313 sections, and each section comprises 832 code elements, and therefore each frame has 260416 code elements altogether.Beginning in each section four code elements are segment sync symbols 92, and these code elements form sequence [+5 ,-5 ,-5 ,+5].The first section in each frame is the frame synchronization section 94 with 312 data segments 96,98.Referring now to Figure 10, frame synchronization section 94 has segment sync 100,511 code element pseudo noise (PN511) sequence 101,63 code element pseudo noise (PN63) sequence the 102, the 2nd PN63 sequence 203 and the 3rd PN63 sequence 104.The 24 pattern code elements 105 of to be pointing-type be afterwards 8VSB.Pre-coded symbol 107 and reserved code element 106 configuration frame sync segment 94.Segment sync 100 and PN511101 code element are known to receiver priori and can be used to by correlation technique to obtain frame synchronization.All aforementioned code elements are derived from collection {+5 ,-5}.Last 12 code elements of this section are from collection {-7-5-3-1+1+3+5+7}, and the copy being last 12 code elements of last data field.They are called as pre-coded symbol (will not discuss) here.

In addition see Figure 11,828 code elements 32 of each follow-up 312 sections---they are referred to as data segments---after four segment sync symbols 30 for field are formed from independent 207 bytes (1657 bit) Reed-Solomon (RS) code word, this be by getting 2 bits at every turn, their trellis codings are become 3 bits and subsequently by per unit 3 bit mapping to from collection, { 8 grades of code elements of-7-5-3-1+1+3+5+7} have been come.

Another example of framing is found in ISDB-T system in digital communication systems.Different from single carrier ATSC system, ISDB-T is the multicarrier system utilizing Coded Orthogonal Frequency Division Multiplexing (COFDM) (COFDM).Such as, the pattern 1 of ISDB-T uses 1404 carrier waves.A frame comprises 204 COFDM code elements and forms, and each COFDM code element can think the combination of 1404 independent QAM code elements, has one for each carrier wave.Therefore, this frame is made up of the combination of 204x1404=286,416QAM code element.Wherein, 254,592 code elements are data, and the pattern information that 31,824 code elements comprise pilot frequency information (can be used for frame synchronization) and are dispersed in known pattern in frame.

The simplicity of illustration of this frame configuration is for Figure 12.Can find out, pilot tone and pattern information are dispersed in around frame with known pattern.This system has and utilizes three kinds of different qam constellation---patterns of QPSK, 16QAM and 64QAM.It also supports five kinds of different trellis coding speed (1/2,2/3,3/4,5/6,7/8) based on female code of single column.This known technology makes to build single Viterbi decoder in receivers and becomes very economical, and described Viterbi decoder is easy to adjustment to decode to all five kinds of codes in designated code.

Before transmitting pusher side carries out trellis coding, data are formed as the long RS block of 204 bytes (1632 bit).Although the COFDM number of symbols in every frame is always constant, but the number of RS block in every frame changes, most significantly, always this number integer with institute's lectotype.Once set up frame synchronization and known trellis coding speed, it is synchronous that this just allows to be easy to RS block in receiver.For realizing this, for all patterns, before trellis coding, the number of the data bit of every frame must be able to be divided equally by 1632.

Table 1 illustrates the number of the data bit of the every frame for all patterns (combination of qam constellation and trellis coding speed).In each case a, the number of every frame data bit can be divided equally (data bit means the bit before trellis coding) by 1632.

Table 1: for every frame data bit of ISDB-T

Some embodiment of the present invention provides a kind of framing structure for the modulating system for digital communication system.Specifically, provide can be applied to comprise above-mentioned those the signaling system of safety system and method.Convolutional byte interleaver interleaving data frame, wherein interleaver and randomizer synchronous with frame structure can be configured to produce through randomized Frame from the Frame interweaved.In one example, under the trellis coded modulation device of perforation is operated in optional code rate, this modulator is from the Frame producing trellis coding through randomized Frame.Bit group in the Frame of trellis coding is mapped to modulated symbol by QAM mapper, provide thus through map frame and synchronizer will synchronously grouping add to through map frame.The trellis coded modulation device of perforated can be bypassed as required with obtaining the optimization net bit rate under various white noise condition, allows the performance optimization of system thus.

In certain embodiments, in carrier wave communication system, novel frame structure is provided.The auto-correlation being in the known pattern of the top of frame or the regular length code element of end in zero offset produces a higher value, if this skew is non-zero, then correlation (secondary lobe) is very little.But this frame synchronization sequence is relevant by formation one smaller value to random code element.Therefore, the storage version of receiver available frame synchronization pattern performs and imports the relevant of code element into, and to obtain the higher value of the accurate starting position at each frame, this makes receiver determine the starting point of each frame.Under communication system can be operated in the arbitrary pattern in various modes, and the various combinations of symbol constellations, trellis code and interlacing pattern can be used.Receiver must identify and understanding pattern successfully to recover transmitted data.For this reason, additional pattern code element can be added into frame synchronization pattern.These pattern code elements can use correlation technique reliably to receive, and repeat to send because they are every frames.Make them more strong by using block code to these mode code primitive encodings.

Those perforation trellis codings and qam constellation combination of being similar to and using in ISDB-T are utilized according to a kind of frame structure of some aspect of the present invention.The code element number of each frame can be depend on the variable integer of pattern and the RS grouping number of each frame no matter why pattern is constant integer.This arrangement simplifies the design of the receiver processing module of such as removing randomizer and deinterleaver, because the RS grouping number of each frame is always fixing.In the legacy system of such as ISDB-T, the code element number of every frame is constant, and the RS grouping number of every frame is the variable integer depending on pattern.The example descriptor frame of the transmitter architecture described with reference to Figure 13, described transmitter architecture builds according to some aspect of the present invention.

RS encoder 1300 accepts byte data 1301 and the outside frame synchronizing signal produced, and the beginning of 315 Reed-Solomon groupings 1322 is often organized in the instruction of this frame synchronizing signal.As shown in figure 14, each grouping 140 comprises 207 bytes, and wherein 20 bytes are parity bytes 142.These 315 Reed-Solomon groupings form forward error correction (FEC) Frame 1322 comprising 62205 bytes.

Convolutional byte interleaver 1302 afterwards.Figure 15 explains orally the mode of operation that reply affects the interleaver 1302 of the impulsive noise of signal transmission.Parameter B in path 156,158 is set as 207, and the parameter M in path 152,154,156 and 158 is set as 1.Input and output change over switch 150,151 is pushed into tip position 1500 by frame synchronizing signal 1303, makes intertexture synchronous with frame structure thus.Along with byte enters interleaver and different bytes exits interleaver, input and output change over switch 150,151 moves down a position 1502.When change over switch 150,151 arrives bottom 1508, their switchings back on top 1500.Each in B parallel route 1506,1508 comprises shift register 156,158, and described shift register 156,158 has the length (path 1506 has length (B-2) M and path 1508 has length (B-1) M) shown in Figure 15.

Randomizer 1306 is by being operated in 65 of FEC Frame 1324,205x8=521,640 bits also produce randomized FEC Frame 328 by performing XOR to those bits of PN (pseudo noise) sequence with length 219-1, by resetting in each frame synchronization time the length that PN sequencer shortens this PN sequence.

The example of trellis coded modulation (PTCM) module 1308 that bit rate can be selected to bore a hole is illustrated in greater detail in Figure 16.The method that PTCM1308 uses those skilled in that art known.The method starts from the encoder of 64 state 1/2 speed, and performs perforation to obtain any one in 5 kinds of different bit rates.In certain embodiments, PTCM1308 also can by complete bypass (bit rate=1).This allows selectable balance between the net bit rate and white noise performance of system.Similar trellis coding technology is used to ISDB-T and DVB-T system.PTCM produces two bits 1332 in the output of each bit being supplied to input 1328.But, according to selected bit rate and respective perforations pattern, some output in bit 1332 are abandoned.QAM mapper 1313 is fetched own coding device and is exported the bit in 1332 2,4 or 6 groups and they mapped to respectively in QPSK, 16QAM or 64QAM code element.The example that these map is given in Figure 17.

Frame synchronization/pattern code element packet (all code elements are QPSK) is added to the beginning of each FEC Frame 1334 by module 1312.See Figure 18, the Part I 180 of this grouping comprises 127 code elements and comprises the identical binary system PN sequence of the real part for code element and imaginary part.Other PN sequence length is also possible, and real part and imaginary part can have contrary symbol.The Part II 182 of this grouping comprises the data of indicating transmission mode, namely selected qam constellation and selected trellis code speed.This mode data can use block error correcting code to encode, for increasing the reliability at receiver place.Adoptable method comprises Bose-Chaudhuri-Hocquenghem Code and other block code.In one example, the trellis code speed comprising 6 kinds of bypass possible is possible.In addition, three constellations may draw 18 kinds of patterns.Therefore, needs 5 bits characterize each in possible model selection.These 5 bits can use the BCH code of expansion to be encoded into 16 bit codewords.Because each QPSK code element comprises 2 bits, therefore need 8 pattern code elements.

Figure 19 explanation is supplied to the frame structure 1336 (see Figure 13) of passband modulation (PB modulation) 1314.Pay(useful) load 190 comprises 315 RS grouping (521640 bits).The number of 315 RS packet map QAM code element extremely can change along with model selection.PB modulation module 1314 uses those skilled in that art's known any proper method that base band QAM code element is modulated to passband subsequently.

Some defect and the shortcoming of conventional frame is usefully overcome according to the frame structure of some aspect of the present invention.Specifically, this frame structure provides all patterns:

The RS of every frame divides into groups constant integer, and no matter pattern is how, and

For all patterns, the QAM code element number of every frame is variable integer

For all patterns, the perforation pattern cycle of every frame is integer.

The QAM code element that the integer of the every frame of attention hypothesis is individual is important realization, because FEC Frame accurately must comprise I × 207 data byte, wherein I is that selected integer has a fixed integer RS grouping to make every frame.Therefore, before trellis coding, the data bit number of every frame must be not only integer, and this data bit number must be able to be divided equally by 207 × 8=1656 for all patterns.In addition, the grid encoder of every QAM code element exports bit number and is respectively 2,4 and 6 bits (see table 2, table 2 illustrates bit rate=1 for trellis code bypass) for QPSK, 16QAM and 64QAM.In addition, trellis coding adds bit.Before trellis coding, the data bit number of each code element is illustrated in table 2, and wherein each entry is calculated as:

The data bit (the QAM code element input bit of each mapping is to grid encoder) of each code element of table 2-

The data bit number of each code element can be RS packet size and the RS grouping number that the fact of mark requires accurately to select each frame.For every frame 207 and 315 RS packet sizes divided into groups, obtain every frame integer code element.As shown in table 3, each entry can be calculated as:

The code element of each frame of table 3-

This frame provides extra advantage, and the perforation pattern cycle (pp/ frame) namely for each frame of all patterns is integer.In order to the trellis coding data of perforated of correctly decoding, the decoder in receiver must know perforation pattern as how about with data alignment.Be applied to being illustrated in the secondary series of the form in Figure 16 by bit perforation pattern of female code grid encoder output.In each perforation pattern several 1 is perforation pattern length.In the system of proposing, perforation pattern is aimed at the starting point of FEC Frame always.This allows to use frame synchronization to aim at bit stream suitably to make the perforator (de-puncturer) that goes in receiver Viterbi decoder in receiver.Indicate the aligning of requirement in table 4, it illustrates the integer number of the pp/ frame for all patterns.Perforation pattern (pp/ code element) entry of each code element can be calculated as:

Pp/ frame entry can be calculated as:

The perforation pattern of the every frame of table 4-

To understand, other combination of RS packet size and every frame packet count can be used to need result to obtain identical conjunction.Here the number provided is only and explains orally object and describe.

As shown in figure 20, some embodiment of the present invention provides a receiver, and this receiving mechanism builds up the frame disposed according to some aspect of the present invention structure.The data transmitted in module 2000 received passband signal also convert thereof into base band QAM code element.The operation performed by module 2000 can comprise chip clock synchronous, balanced (to remove intersymbol interference) and carrier auxiliary, usually uses submodule.Therefore, module 2000 can comprise the equalizer of the base band QAM code element 2001 exported through recovering.Base band QAM signal 2001 is provided to secondary sheer 2018 to do cutting to real axis and imaginary axis both direction, forms the sequence a being supplied to frame synchronization module 2020 thus _r[k] ∈ [-1 ,+1] and a _i[k] ∈ [-1 ,+1] 2019.The stored copies of frame synchronization module 2020 to the QAM code element 2019 binary frame synchronic PN sequence importing cutting into performs continuous cross-correlation operation for real part and imaginary part separately.Each member of stored copies has value-1 or+1.This operation is expressed as:

$b_R\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_R[n-k]$ And $b_I\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_I[n-k],$ (equation 10)

Wherein s is the copy stored in 127 long frame synchronization PN sequences.B _ror b _imaximum amplitude represent the beginning of FEC Frame.

Once the synchronous starting position of locating frame, the position of the code word comprising mode bit (constellation and trellis code rate) just can be known.By such as BCH decoder or by received code word is relevant to all possible code word and select the code word producing the highest end value reliably to decode to code word.Because this information is repeatedly transmitted, therefore by requiring to make identical result repeatedly obtain extra reliability before acceptance.

This frame synchronizing signal 2021 derived out be used to refer to code element is fed to soft remove mapper 2006 before intend removing which code element in " removing frame-synchronous/pattern code element " module 2004.In one example, 127 frame synchronization code elements and 8 pattern code elements are removed from stream, be delivered to corresponding code element of guaranteeing only to divide into groups with RS and softly remove mapper 2006.The soft mapper 2006 that goes uses algorithm known in the art (algorithm such as described by Akay and Tosato) to calculate soft bit metric.In order to correct work, softly go mapper 2006 must know in transmitter to use which kind of perforation pattern (which kind of trellis code speed) and aiming at of this pattern and received bit will be known.This information 2021 is provided by frame synchronization module 2020, this frame synchronization module 2020 decoding schema information the repeating frame synchronizing signal that simultaneously provides perforation pattern to aim at it, and no matter present mode why.These soft bit metric are fed to Viterbi decoder 2008, this Viterbi decoder 2008 with mode work known in the art with draw the PTCM encoder inputed in transmitter bit estimation.

All by synchronous the going randomizer 2013, byte deinterleaver 2014 and RS decoder 2016 to go randomization respectively, to deinterleave and decoded word joint number obtains the data of the RS encoder entered at first in transmitter according to this of frame synchronizing signal 2021.

carrier phase offset corrects

Some embodiment of the present invention adopts carrier phase offset corrective system and method.In certain embodiments, receiver comprises: phase offset correction device, and this phase offset correction device receives the equalizing signal as the sign of quadrature amplitude modulated (QAM) signal and obtains the signal through phasing from equalizing signal; Secondary sheer, this secondary sheer cutting equalizing signal is to obtain real number and imaginary number sequence; Frame synchronizer, this frame synchronizer performs real number and imaginary number sequence and stored frame synchronization pseudo random sequence and the corresponding real part of phase correction signal and the relevant of imaginary part that are supplied to phase offset correction device by frame synchronizer.Phase correction signal is based on relevant maximum real number and imaginary value.Frame synchronizer performs continuous print crosscorrelation to the cutting quadrature amplitude modulation code element imported into.Continuous crosscorrelation carries out for real number and imaginary number sequence separately by the stored copies of the synchronous PN (pseudo noise) sequence of binary frame.

Base band is modulated to passband

Comprise broadcast, some radio digital communication system of WLAN and wide area mobile system adopts QAM with some form.QAM is also for using North America and the European digital cable television standard of quadrature-carrier multiplexing, and quadrature-carrier multiplexing allows the carrier modulation ripple of two double suppress sidebands to occupy identical channel width, and each ripple is modulated by independent message.As mentioned above, Figure 23 describes the simple QAM modulation device of the PB modulation 1314 can served as in Figure 13 example.The QAM pulse of isolated transmission is expressed as follows:

$s_m\left(t\right)=d_{R,m}q\left(t\right)\cos \left(2\mathrm{\&pi;}{f}_{c}t\right)-d_{I,m}q\left(t\right)\sin \left(2\mathrm{\&pi;}{f}_{c}t\right)=\mathrm{Re}\left\{d_{m}q\left(t\right)e^{j2\mathrm{\&pi;}{f}_{c}t}\right\},$

Wherein d _{r, m}and d _{i, m}the real part and the imaginary part (see such as Figure 17) that also represent multiple QAM code element respectively determined by two independent message streams, wherein 2 dimension qam constellations of m=1...M index radix, wherein M is modulated carrier frequencies, and q (t) is square-root raised-cosine impulse function.

Transmission QAM pulse s (t) of one continuous series is with speed F _s=1/T _sby being with the multi-path channel transmission of noise.Therefore, r (t)=s (t) * c (t)+v (t) is expressed as at the Received signal strength of the input of QAM receiver, wherein * represents convolution, and c (t) is channel impulse response, and v (t) is additive white Gaussian noise.Therefore:

$r\left(t\right)=\mathrm{Re}\left\{e^{j2\mathrm{\&pi;}(f_{\mathrm{LO}}+f_o)t+{\mathrm{\&theta;}}_o}\underset{n=-\&infin;}{\overset{+\&infin;}{\mathrm{\&Sigma;}}}\right[d\left[n\right]*q\left(t\right)\left]c(t-n{T}_s)\right\}+v\left(t\right),$

Wherein d [n] is multiple transmitted symbol, f _oand θ _obe respectively receiver pass-band relative to transmitter to the frequency shift (FS) of base band demodulator local oscillator and phase deviation, thus f _lO=f _c-f _o.

Passband is to base band demodulator

Figure 35 illustrates in greater detail that PB to BB in Figure 20, chip clock are synchronous, an example of equalizer/carrier recovery block 2000.Signal r (t) received is sampled 350 with the speed higher than chip rate, produces the signal r (nT through sampling _samp).After sampling:

$r\left(n{T}_{\mathrm{samp}}\right)=\mathrm{Re}\left\{e^{j2\mathrm{\&pi;}(f_{\mathrm{LO}}+f_o)n{T}_{\mathrm{samp}}+{\mathrm{\&theta;}}_o}\underset{m=-\&infin;}{\overset{+\&infin;}{\mathrm{\&Sigma;}}}\right[d\left[m\right]*q\left(n{T}_{\mathrm{samp}}\right)\left]c({\mathrm{nT}}_{\mathrm{samp}}-m{T}_s)\right\}+v\left(n{T}_{\mathrm{samp}}\right).$

Then, after demodulation, with chip rate 1/T _sagain sampling and matched filtering obtain:

$x\left({\mathrm{kT}}_s\right)=x\left[k\right]=e^{j2\mathrm{\&pi;}{f}_{o}k{T}_s+{\mathrm{\&theta;}}_o}\underset{m=-\&infin;}{\overset{+\&infin;}{\mathrm{\&Sigma;}}}d\left[m\right]c[k-m]+v^{\&prime;}\left[k\right],$

Wherein v ' [k] is the multiple filter noise through sampling.This hypothesis combines due to pulse shaping and matched filtering q and perfect chip rate sampling timing, and any ISI is attributable simply to channel impulse response c.After demodulation, suppose perfect equilibrium, then the nearly base band complex sequences z [k] of equalizer output is expressed as:

$z\left[k\right]=\hat{d}\left[k\right]e^{j2\mathrm{\&pi;}{f}_{o}k{T}_S+{\mathrm{\&theta;}}_o}+v^{\&prime;}\left[k\right]$

Therefore, the constellation that the nearly baseband sequence representative through recovering sends, it has with frequency f _othe phase deviation θ rotated _o.In order to reliably recover the d sent _rand d _i, use such as two-dimentional sheer, equalizer and phase place and frequency shift (FS) to recover loop and combine the frequency shift (FS) f that must eliminate and cause constellation rotation _o, this and receiver must eliminate θ _oremaining static phase offset, otherwise the position that constellation rotates for static state will be made.

Recovering to understand phase/frequency, it must be understood that the qam constellation at base band place.In the simple examples of Figure 33 A, modulate for the 4QAM also referred to as QPSK, constellation is made up of four code elements.In described example, the real part of d [k] and imaginary part can respectively be asked for 2 different values (such as ± 3).Phase deviation θ _obe illustrated in Figure 33 B to the effect of the d [k] recovered, Figure 33 B illustrates the rotation in complex plane.Note rotating and carry out in a circle in time, according to f _osymbol counterclockwise or turn clockwise, so just can understand f _oeffect.

Equalizer and carrier phase/frequency ring

In Figure 34, signal x [k] 340 is received by digital equalizer and carrier phase/frequency ring 248 (for example, see Figure 24 A).The parts of equalizer 341 generally include linear digital filter, and use the privately owned or known method of such as lowest mean square (LMS) algorithm, and equalizer 341 is exported y [k] and adjudicates with sheer through the edition comparison of phase rotating to form error signal, this error signal is used for the renewal collection of calculating filter tap-weights.The ISI caused by channel impulse response c removed by this filter.

2-D sheer 342 real part of cutting z [k] and imaginary part export independently it is the estimation of the d [k] of original transmitted.Z [k] and both enter phase error detector module 346 and form phase error signal, and this phase error signal is expressed as integration ratio (IP) filter 345 can comprise any equivalent that the filter of Figure 35 or those skilled in that art know.IP filter 345 allows loop phase calibration to offset and frequency shift (FS).The output of IP filter 345 is fed to multiple voltage controlled oscillator (VCO) 344, and this voltage controlled oscillator 344 exports for correcting θ _oand f _oboth complex phase position/frequency correction factor e ^{-j θ [k]}.VCO344 also exports (e ^{+ j θ [k]}) export with " not correcting " cutting the error signal obtaining upgrading for equalizer tap is can be used to make it.The method is shown, because equalizer is operated in comprise θ _oand f _oon both x [k].

In certain embodiments, by VCO344 being embodied as the delay of an integrator of feeding one complex exponential look-up table (LUT) with absolute version to obtain effect.But, to θ _ocorrection of a final proof can have the ambiguity of pi/2, this ambiguity means that the phase place through recovering may be the skew that correct (skew=0) maybe can have the skew of pi/2, the skew of π or 3 π/4.These results are illustrated in the code element that Figure 36 and Figure 37: Figure 36 illustrates actual transmission, and Figure 37 A-37D illustrates the possible recovery code element with each skew.Typically, receiver cannot know that these four kinds may any in code elements be actually being sent, because 2D sheer 342 performs nearest adjacent operation.Figure 38 illustrates that the code element α wherein sent is received as the example of α ' in equalizer input, as shown in the figure, has angle theta between α and α ' _o.Therefore, phase recovery loop can make signal rotation to compensate θ _o, aim at a to make a '.But the judgement of 2D sheer 162 will be correction code element is b, because it is closer to α '.This may cause phase recovery loop to restrain in alignment with the mode of b to make a ' with rotated constellation.In this case, final phase place from its should position skew-pi/2.

Some embodiment of the present invention provides the method for these problems minimizing and/or eliminate in trellis coding system, is included in the analog of the perforation trellis code used in the embodiment of some current description.As mentioned above, the output of equalizer is split with formation sequence a along real axis and imaginary axis direction by 2D level sheer 342 _r[k] ∈ [-1 ,+1] and a _i[k] ∈ [-1 ,+1], this sequence is fed to frame synchronization module 2020 (see Figure 20).Frame synchronization module 2020 uses the stored copies of binary frame synchronic PN sequence in cutting QAM code element, to perform continuous print cross-correlation operation what import into respectively to real part and imaginary part.Each member of stored copies has value-1 or+1.This computing is characterized as:

$b_R\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_R[n-k]$ and $b_I\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_I[n-k],$

Wherein s is the stored copies in 127 long frame synchronization PN sequences.B _ror b _imaximum amplitude represent the beginning of FEC Frame.

Maximum b RSymbol	Maximum b ISymbol	Required phasing
			+	+	0
-	+	+π/2
			-	-	+π
+	-	-π/2

Table 5

For frame synchronization code element, real part and imaginary part have identical symbol and its constellation is illustrated in Figure 39.Therefore, can understand, maximum amplitude b _ror b _isymbol for zero rotate just be.The rotation of-pi/2 produces negative maximum amplitude b _rwith positive maximum amplitude b _i.For the rotation of π, b _rand b _ibe negative, for the rotation of pi/2, maximum amplitude b _rfor just maximum amplitude b _ibe negative.This summarizes in above table 5.Therefore, maximum amplitude b _rand b _ieach symbol indicate final phase deviation to converge to which quadrant in complex plane together.This allows such as illustrated in fig. 20 additional phase correction to be put on signal.Maximum b _rand b _isymbol be sent to phase offset correction device from based on relevant frame synchronization module.The operation of a phase offset correction device module is illustrated in Figure 40, wherein shows LUT operation 404 in one example.Suppose z [k]=z _r[k]+jz _i[k], this operation can perform simply and be:

When φ=+ θ: z ' [k]=-z _r[k]-jz _i[k]

When φ=+ pi/2: z ' [k]=-z _i[k]+jz _r[k]

When φ=-pi/2: z ' [k]=+ z _i[k]-jz _r[k]

Figure 40 is the block diagram obtaining the phase offset correction device through phase correction signal according to the look-up table having a symbol of maximum real number and imaginary number correlation by index of some aspect of the present invention.

multimode qam constellation detects

Some embodiment is provided for the system and method determining a unknown qam constellation from one group of possible reception qam constellation.A kind of method use the constant modulus algorithm of correction (CMA) equalizer minimize intersymbol interference (ISI) after but before carrier frequency and phase place are recovered completely, utilize the distribution map of signal power.Then unknown constellation is determined from this distribution map.Then again start balancing procedure based on nowadays known constellation standard C MA to minimize to make ISI.Can correctly export by convergent-divergent equalizer, the constellation carrier auxiliary (RCCR) and decision-directed carrier auxiliary stage of simplifying can be performed after this, result through the carrier frequency of equalizer carrier frequency/phase loop and the recovery of phase place of combination thus.In the other method for determining unknown qam constellation, equalizer uses the CMA through revising to work to minimize ISI at the beginning.May not by correctly convergent-divergent at this time point of this process although equalizer exports, but equalizer carrier frequency/phase loop can use RCCR to come reinsertion of carrier frequency and phase place and without the need to why knowing constellation.The phase place recovered may be band noise.Receiver can read the information be embedded in signal frame, and which qam constellation the instruction of this information is transmitting.Then restarting equalizer operation based on known constellation standard C MA, is RCCR and decision-directed carrier auxiliary afterwards.

Some embodiment of the present invention adopt be similar to use in ISDB-T with those perforation trellis codings aforesaid and qam constellation combination.As used herein, constellation is understood to represent the mapping in the complex plane being in possibility code element in modulation scheme.The code element number of each frame is the variable integer depending on pattern, and the RS packet count of each frame is constant integer, and no matter pattern why.This be configured in before illustrated in greater detail and simplified the design of receiver.

Refer again to Figure 20, frame synchronization module 2020 uses the stored copies of binary frame synchronic PN sequence to perform continuous print cross-correlation operation to real part and imaginary part respectively in the cutting QAM code element 1219 imported into.Each member of stored copies has value-1 or+1.This computing provided by equation 10 (above) is here repeated:

$b_R\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_R[n-k]$ And $b_I\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_I[n-k],$ (equation 10)

Wherein s is the stored copies in 127 long frame synchronization PN sequences.B _ror b _imaximum amplitude represent the beginning of FEC Frame.

As described in greater detail below, in exalted carrier phase place, there is the ambiguity of pi/2.This causes 0, the recovery phase deviation additional arbitrarily of ± pi/2 or π.For frame synchronization code element, real part and imaginary part have identical symbol, and the constellation therefore sent them as shown in figure 39.Therefore, will understand, maximum amplitude b _ror b _isymbol for zero phase skew be just.As summarized in the form 404 of Figure 40, the skew of-pi/2 will produce negative maximum amplitude b _rwith positive maximum amplitude b _i; For the skew of π, b _rand b _iboth are negative, and for the skew of pi/2, maximum amplitude b _rto be just maximum amplitude b _ito be negative.Therefore, maximum amplitude b _rand b _ieach symbol indicate final phase deviation to converge to which quadrant in complex plane together.This allows the signal additional phasing being put on phase offset correction device module 2002.Maximum b _rand b _isymbol can be sent to phase offset correction device 2002 from based on relevant frame synchronization module 2020.

See also Figure 40, the operation in some of the phase offset correction device 2002 in Figure 20 example can be better understood.LUT400 produces based on maximum amplitude b _rand b _ithe output (element 404 see in Figure 40) of symbol.Suppose z [k]=z _r[k]+jz _i[k], operation 402 performs as follows:

1) for the situation of φ=+ π: z ' [k]=-z _r[k]-jz _i[k]

2) for situation: z ' [k]=-z _i[k]+jz _r[k]

3) for situation: z ' [k]=+ z _i[k]-jz _r[k]

Once the synchronous starting position of locating frame and correct the phase deviation of m pi/2, the position of the code word comprising mode bit (constellation and trellis code rate) just can be known.By such as Bose-Chaudhuri-Hocquenghem Code device or by received code word is relevant to all possible code word and select the code word producing the highest end value to decode reliably to code word.Because this information is repeatedly transmitted, therefore by requiring to make identical result repeatedly obtain extra reliability before acceptance.

Figure 41 illustrates the example that this process can be performed by frame synchronization module 2020.In response to frame synchronizing signal 2021, in step 4100, the constellation code word received and all effective code word crosscorrelations.Crosscorrelation produces the value that can be used to select most probable coupling.In one example, select in step 4102 the effective code word producing maximum related value.The code word of this selection then can be used to identify current constellation.In step 4104, by the identity of current constellation and record or otherwise store before identify that the identity of constellation is made comparisons.If in step 4104, current constellation and the constellation identified before are same constellations, then confidence counter increases progressively.If the constellation identified before step 4104 is determined is different from current constellation, then confidence counter also successively decreases in step 4017 by the constellation identified before current constellation is recorded as by step 4107, and waits for another synchronization frame in step 4109.After confidence counter increases progressively by step 4106, check confidence counter in step 4108, if determine that confidence counter has exceeded threshold value that is predetermined or configuration in step 4108, then make the judgement of signal constellation (in digital modulation) in step 4110.The iteration of this process can be performed until confidence counter exceeds threshold value that is predetermined or configuration.

Equalizer and carrier phase/frequency ring

See Figure 42, by the equalizer of description Figure 24 A and some aspect of carrier phase/frequency loop 248.Signal x [k] enters digital equalizer and carrier phase/frequency loop 248, and this digital equalizer and carrier phase/frequency loop 248 can comprise the equalizer 420 with linear digital filter.Error Calculator module 422 error signal e [k], this error signal e [k] can use any proper method well known by persons skilled in the art to carry out the renewal collection of calculating filter tap-weights.In one example, LMS algorithm can be used.The ISI caused by channel impulse response c removed by filter.The output y [k] of equalizer 420 makes phase rotating to reduce any remaining carrier phase and frequency shift (FS) 421 subsequently.Then processed by sheer and phase error detector module 427 through the output z [k] of phase rotating, described phase error detector module 427 calculates the phase error e being fed to integration ratio (IP) filter 426 _θ[k].The output of IP filter 426 is fed to integrator and complex exponential look-up table (LUT) 424, and the complex exponential value used in this complex exponential look-up table (LUT) 424 calculating loop is to correct carrier phase and frequency shift (FS).Sheer and phase error detector module 427 also export the two-dimentional sliced symbols judgement of nearest neighbor, its phase place 425 by with e ^{+ j θ [k]}be multiplied and " not being corrected " Error Calculator module 422 will be used to subsequently.Error Calculator module 422 utilizes this input and x [k] to carry out error signal e [k].As depicted, the current generation of operation (1,2 or 3) is depended in the built-in function of Error Calculator module 422 and sheer and phase error detector module 427, and these stages are determined by phase controller 423.

In certain embodiments, use lowest mean square (LMS) algorithm to calculate equalizer filter tap weight and do following operation:

Make x [k] represent the equalizer input vector of L length, y [k] represents equalizer output vector, wherein y [k]=g ^h[k] x [k], wherein g ^h[k] is the linear equalizer tap-weights vector of L length and H subscript represents that conjugation exchanges (close conjugation in distress).Then, use such as method described below carrys out the e [k] through upgrading in error of calculation calculator modules 422:

G [k+1]=g [k]-2 μ x [k] e ^*[k], (equation 11)

Wherein μ is little step parameter and subscript * represents complex conjugate.

In this example, phase controller 423 obtains equalizer and carrier phase/frequency loops 428 by three stages operatings, takes this to perform switching from 2 to the stage of 1 to stage stage 3 based on the simple count threshold value of input data sampling x [k].Note, it is also possible that the more complicated stage based on the error estimation of equalizer outlet side switches.This three phases is summarized in table 6.

Table 6: equalizer and carrier phase/frequency loop stage

Sheer and phase error detector module 427 illustrate in greater detail in Figure 43.According to a configuration switch 430 in the three phases 434 of operation.During the stage 1, switch 430 is in extreme higher position, therefore e _θ[k]=0.This is disconnected carrier loop effectively, thus during this stage without any carrier phase correction.During the stage 2, switch 430 mediates and loop uses the work of simplification constellation carrier auxiliary (RCCR) algorithm.If with | z [k] | ²the power of code element z [k] represented exceeds threshold xi, then suppose that z [k] is in the corner code element of constellation one and RCCR enables by the second switch 432 of description is set in a high position, obtain otherwise, if | z [k] | ²≤ ξ, then second switch 432 is in the low level of the forbidding carrier loop of description.Therefore, a subset is only had during the stage 2, benifit can be had to carrier auxiliary in code element.Threshold xi can be reduced to make to comprise height more multiple-symbol in the region of constellation near corners, but the phase correction terms e of gained _θ[k] will with more noises.During the stage 3, switch 430 is in the extreme lower position of description, obtains wherein 2 dimension sliced symbols judgements of nearest neighbor complex conjugate.During the stage 3, suppose through time enough to make equalizer tap restrain and carrier phase basic correction, the bit decision thus through cutting is reliable.Especially, relational expression with effectively be operated in the single quadrant of complex plane.This causes the ambiguity of m pi/2 in exalted carrier phase place as described above.

The example of IP filter 426 (see Figure 42) illustrates in greater detail in Figure 35.IP filter 426 allows loop phase calibration and frequency shift (FS).The output of IP filter 426 is fed to integrator and complex exponential LUT module 424, Figure 45 specifically illustrates.The input of integrator/LUT424 is added step time delay 442 version of 440 (Figure 44) to input by 2 π moulds, thus form phase error signal θ [k], this phase error signal θ [k] is fed to look-up table (LUT) 444, and this look-up table 444 exports phase correction factor 445e ^{-j θ [k]}, it is used for correcting θ _oand f _oboth.LUT444 also provides output 446 (e ^{+ j θ [k]}), this output " does not correct " sheer and exports can be used for make it deriving the error signal upgraded for equalizer tap.The method is necessary, because equalizer is operated in comprise θ _oand f _oon both x [k].

Error Calculator module and stage computing are summarized

Error Calculator 422 can adopt distinct methods to calculate e [k] according to the stage.Stage 1 and stage 2, e [k] process typically used based on constant modulus algorithm (CMA) is calculated:

e[k]＝y[k](|y[k]| ²-R)，

Wherein R is predetermined constant, and it is expressed as:

$R=\frac{E\left\{{\left|d\right[k\left]\right|}^4\right\}}{E\left\{{\left|d\right[k\left]\right|}^2\right\}},$ (equation 12)

Wherein E is expectation operator and d [k] is code element (see Figure 17).This e [k] that the tap noting triggering above equation 11 upgrades is the phase place independent of bit decision and x [k], and only depend on that equalizer exports, the statistics of equalizer input and constellation.Can find out, during stage 1 and stage 2, use CDMA error to trigger equation 11 and be equivalent to and ISI minimized, even if constellation rotates due to carrier frequency and phase deviation.

Therefore, during the stage 1, it is forbidding that phase/frequency recovers loop, and equalizer uses CMA error function that ISI is minimized.After making ISI minimize, the stage 2 start and loop for RCCR conducting; Carrier phase/frequency is recovered only use the corner code element of constellation and start, as illustrated in conjunction with Figure 43 above.At the end of the stage 2, carrier phase and frequency are fully recovered to make 2 of Figure 43 dimension sheer 436 start to export reliable bit decision

Decision-directed (DD) error can use in the stage 3.DD error can be calculated as for purpose of explanation, which in three constellations of Figure 17 what suppose that receiver determined just to send here is, because for each in these constellations, R is different.In addition, RCCR requires to know constellation, especially knows the power of the corner code element of constellation.

There is the CMA of unknown constellation

In example described herein, in three different Q AM constellations can be sent, and above-mentioned equilibrium and phase/frequency are recovered to need to know sent constellation.Although constellation is selected to encode in pattern code element, but equilibrium and phase/frequency are recovered to lead over frame synchronization (see Figure 20), and now this information is directly decoded (for example, see Figure 18, Figure 20 and Figure 41) as described above.As a result, in certain embodiments, in equalizer and carrier recovery algorithm itself, constellation is determined.

Notice that R (as provided in equation 12) is that constellation is interdependent.In certain embodiments and to continue see the real part of the code element of Figure 17,64-QAM and imaginary part be from collection ± { 1,3, select in 5,7}, the real part of the code element of 16-QAM and imaginary part are from collection ± { 2, select in 6}, and the real part of QPSK code element and imaginary part are selected from collection ± 4.Value according to equation 12, R will be:

For any one in three constellations of Figure 17, can find out, use CMA error calculation through scale value α R make equalizer filter tap converge to by the same class value of convergent-divergent, wherein equalizer exports scaled equally.Can find out, ISI is still minimized.In an example of constellation the unknown, R can be set as 58 and the constellation that no matter sends why, ISI is minimized during the stage 1.For described example, can any value R in scope of application 32-58.But the selection of maximum (namely 58) prevents from compressing at the most dense constellation (being here 64-QAM) of equalizer output and alleviating the burden of equalizer performance.

Use through the CMA parameter R of convergent-divergent causes the upwards convergent-divergent exported by the equilibrium of the filter tap of restraining, and the statistical value that therefore equalizer exports will be:

E{|y[k]| ⁴}/E{|y[k]| ²}＝58，

Suppose that perfection eliminates ISI and no matter constellation is why.Therefore, for QPSK, during the stage 1, equalizer exports will carry out convergent-divergent as follows after ISI minimizes:

$y\left[k\right]=e^{j2\mathrm{\&pi;}{f}_{0}k{T}_S+{\mathrm{\&theta;}}_0}\sqrt{\frac{58}{32}(\&PlusMinus;4\&PlusMinus;j4)}=e^{j2\mathrm{\&pi;}{f}_{0}k{T}_S+{\mathrm{\&theta;}}_0}(\&PlusMinus;5.385\&PlusMinus;j5.385).$

Figure 45 A explains orally at θ ₀=f ₀the real part that the equilibrium of the system of QPSK exports is used when=0.Can see because value R=58, when equalizer convergence is to when eliminating the solution of ISI, export by convergent-divergent.Figure 45 B explains orally at θ ₀=f ₀the real part that the equilibrium of the system of 16-QAM exports is used when=0.Due to relatively closer to 1, the real part that therefore equalizer exports looks only slightly scaled.Therefore, during equalizer convergence, actual convergent-divergent is obvious.

Constellation detection method

In certain embodiments, can before entering the stage 2, use profiling method to determine constellation.Even if carrier phase and frequency are not yet resumed, also constellation can be determined.Consider equalizer power output η [k]=y [k] y ^*the distribution map of [k], Figure 46 A, 46B and 46C illustrate QPSK constellation, 16-QAM constellation and 64-QAM constellation respectively.These distribution graphs be shown in equalizer restrain after power, wherein R=58.The power exported due to equalizer independent of phase place and the distribution map of each constellation is substantially different, therefore can determine from equalizer power output distribution map the constellation that sends in receivers.

Do not have additivity or tap noise, for QPSK constellation, the power that each equalizer exports sampling is η [k]=58.For 16-QAM constellation, the probability mass function of the power that equalizer exports is:

Equally, for 64-QAM constellation, the probability mass function of equalizer power output is:

Because the tap on input signal upgrades noise and additive noise v ', even if concerning the substantive SNR of such as 30dB, also have certain expansion in the distribution map of these values.The noise modeling exported by equalizer is additivity and independent of code element, and supposes that output does not have ISI, so

| y [k] | ²=| d [k]+n [k] | ²=| d [k]+n [k] | ²=| d [k] | ²+ | n [k] | ²+ 2Re{d [k] n ^*[k] } (equation 16)

The variance that given code element adjusts---itself and 2Re{d [k] n ^*[k] } item association---increase along with symbol power and increase.In distribution map, this phenomenon is expressed as the expansion around given constellation power, i.e. variance, and it increases along with symbol power and increases.Under 16-QAM situation, around code element ± 2.1 ± expansion of the constellation power of j2.1 is less than around code element ± 6.3 ± expansion of the constellation power of j6.3.

Some other relation can be observed from the distribution map of equalizer power output:

Region T in QPSK distribution map ₁drop on second and the 3rd region R of 16-QAM distribution map roughly respectively ₂and R ₃between.Therefore, the region of which kind of symbol power of declaration transmission is nonoverlapping for QPSK and 16-QAM constellation.

QPSK distribution map discloses for 64-QAM's to the comparison of 64-QAM distribution map therefore, for η [k] and region T ₁comparison, more likely η [k] drops on outside this region.

Do not exist in the example of noise in 64-QAM, η [k] is from collecting { with probability 9/16 value in 2,18,26,34,58,98}.Therefore, when potential constellation is 64-QAM, noise is ignored:

Pr{ (η [k] ∈ R ₁) ∪ (η [k] ∈ R ₂) ∪ (η [k] ∈ R ₃) < 1/2, (equation 17)

Wherein ∪ represent OR (or).Therefore, if the constellation sent is 64-QAM and by η [k] and region R ₁, R ₂and R ₃relatively, then η [k] likely drops on outside these regions.

Some embodiment adopts the algorithm observed based on these:

This algorithm can start after equalizer is restrained, and if equalizer power output is in region T ₁in, then in N number of equalizer output sampling, make QPSK counter λ at Part I ₄[k] increases progressively.If equalizer power output is not in region T ₁in, then by counter decrement.Equally, if η [k] drops on region R ₁, R ₂and R ₃in, make 16-QAM counter λ ₁₆[k] increases progressively, otherwise just makes it successively decrease.

After N equalizer exports sampling, can suppose that distribution map is special by correct earth's surface.If potential constellation is 64-QAM, because more power capability estimates that η [k] will drop on outside QPSK and 16-QAM region, therefore QPSK and 16-QAM counter will be quite little.If the constellation sent is QPSK or 16-QAM, then the counter sending constellation will be significantly larger.Therefore,

Threshold value M can determine by rule of thumb, but it should be less relative to N.This algorithm is very strong, when transmitting QPSK, 16-QAM or 64-QAM, reliably selects to correct constellation for low signal-to-noise ratio (SNR).After reliably determining constellation, R can be set as that correct equation 13 is worth and the stage 1 can tend to finish.Equalizer export by by suitably convergent-divergent and the stage 2 can from the threshold xi known required by RCCR.

Be described in now the another kind of method determining constellation before equalizer enters the stage 3.In the method, execution phase 1 allow to finish at R=58.Therefore, as described, all three constellations are scaled at equalizer output, and this obtains the y [k] as shown in three constellations of Figure 47, although these constellations may be rotate.As described in conjunction with Figure 43, the key in stage 2 is, RCCR is with | z [k] | ²uniquely considering of code element when the power of code element z [k] represented exceeds threshold xi.Then can suppose that z [k] is one in the corner code element of constellation.Equivalently, corner code element can be indicated.As explained orally for 64-QAM constellation in Figure 48 (A), the value selecting ξ when constellation is known is relatively easy.Figure 48 illustrates all three constellations recovering loop module input in equalizer output and carrier phase/frequency.Can find out, for these corner points | z [k] |=9.90.Such as, the threshold value represented by broken circle 484 guarantee only to select corner point.Equally, circle 482 Hes circle 480 can use respectively together with the abundant allowance of 16-QAM with QPSK.

Figure 49 illustrates the coverage diagram of the right upper quadrant of all three constellations.Can find out, if the corner point then only dropping on QPSK and 16-QAM outside broken circle can be utilized by RCCR.But if receive 64-QAM, then five constellation point (four non-corners) are dropped on outside circle and also will be utilized by RCCR.Because the phase place recovered has less noise, RCCR generally behaves oneself best when using corner constellation point.But even if use some annex points, although can cause the increase of phase noise, RCCR also successfully will recover phase place.Therefore, the stage 2 can be operated at the beginning this allows the fully initial carrier auxiliary of all three constellations while constellation keeps the unknown to receiver.

As described in conjunction with Figure 20 above, equalizer 2000 is fed to 2 grades of sheer 2018, and 2 grades of sheer 2018 are then fed to frame synchronization 2020 successively.Frame synchronization 2020 can use the storage version of binary frame synchronic PN sequence to the continuous cross-correlation operation of semiology analysis of the QAM code element through cutting imported into, described by equation 10.Continuous cross-correlation operation can perform real part and imaginary part independently.Each member of storage version has the value of-1 or+1.B _rand b _ithe beginning of maximum amplitude instruction FEC Frame.Nowadays unique difference is, for 64-QAM constellation, 2 grades of sheer 2018 be operated in there is a certain additional phase noise signal on.But this additional phase noise is to 2 grades of cuttings and next have very little negative effect, even if it is also very strong when there is phase noise based on the frame synchronization of crosscorrelation.As previously mentioned, the decoding of constellation code word is also very strong for phase noise.

Figure 50 explanation is for determining the operation of this possible alternative of constellation, and this way can be summarized as follows:

(1) equalizer and phase/frequency loop make the stage 1 finish at R=58, then enter the stage 2.

(2) different from loitering phase 3, be but accept input data during the stage 2 based on relevant frame synchronization 2020, find frame synchronization, and constellation code word is decoded.

(3) constellation information 2021 determined is sent back to equalizer 2000 and phase/frequency loop, this phase/frequency loop uses R value to turn back to the stage 1, and this R value suitably corresponds to fixed constellation.

(4) then ending phase 1,2 and 3 as before.

To understand, the initial differences between the system that Figure 50 and Figure 20 describes is the additional connection 5000 from frame synchronization 2020 to the equalizer/carrier auxiliary 2000 of carrying constellation information.

sPOT in coaxial cable safety chain monitors

Some embodiment of the present invention improves and comprises the system of those systems aforementioned and the performance of device, wherein baseband video signal can combine with the digital representation of baseband video signal and with control signal, allows thus to transmit on the unit cable of such as coaxial cable (" coaxially ").Refer again to Fig. 4, one embodiment of the present of invention provide safety chain on coaxial cable (SLOC) system.Fig. 5 illustrates a kind of possible modulation scheme of SLOC system.In this example, HD video camera 30 provides the IP output 41 comprising compressed digital HD video image and the auxiliary camera signal comprising simulation SDCVBS330.Compressed HD video IP signal 332 utilizes SLOC camera side modulator-demodulator 49 to be modulated to passband 52, SLOC camera side modulator-demodulator 49 to comprise QAM modulation device (modulator 212 see in the modulator-demodulator 32 of Figure 21).Modulator 212 provide can with baseband analog CVBS signal 330 combine through modulation signal.Signal " to downstream " transmission on coaxial cable 41 through combination may extend to 300 meters or longer distance usually.In monitor side, base-band CVBS signal 330 is separated with passband downstream IP signal 332 by SLOC monitor side modulator-demodulator 45.The CVBS signal 330 be separated is fed to SD display 43 in real time, without the viewing of time delay.Passband downstream IP signal 332 is demodulated with qam demodulator (demodulator 222 see in Figure 22), and signal is exported to mainframe network switch 44 or processor/DVR (not shown in Fig. 4) by this demodulator 222.

In this embodiment, upstream communication is provided according to IP protocol requirement.Upstream communication 334 can be used extraly so that audio frequency and camera control signal are delivered to video camera 40 from monitor side.As a rule, the bit rate of stream signal---and therefore required bandwidth---is far below the bit rate required by the passband signal of downstream and bandwidth.Monitor side SLOC modulator-demodulator 45 comprises QAM modulation device (modulator 224 see in Figure 22), this QAM modulation device by IP signal madulation to upstream passband 44.As depicted in figures 5, upstream passband 54 and downstream passband 52 are positioned at different spectrum positions.In camera side, SLOC modulator-demodulator 49 comprises the qam demodulator (demodulator 214 see in the modulator-demodulator of Figure 21) for receiving stream signal.This way provides the some advantages being better than existing system and method, comprises the working range of increase, is easy to use existing coaxial cable infrastructure to dispose and obtain low delay, real-time video.The simplified schematic of Figure 21 and Figure 22 illustrates the further details of the SLOC camera side modulator-demodulator 49 of Fig. 4 and the SLOC monitor side modulator-demodulator 45 of Fig. 4, and describes in detail above.

Figure 51 A illustrates the SLOC system based on Fig. 4 institute interpretation system, and the tap 519 wherein through filtering is arranged between coaxial cable segmentation 512 and 514 to make tap 513 and cable segmentation 512,514 do in order to camera side equipment is connected to monitor side component.Tap 513 through filtering is generally used for base-band CVBS signal 5100 to be extracted into camera side SD display 5130 at least partially.Display 5130 can be arranged near video camera 510 for test, configuration and/or local monitoring.Tap 513 through filtering generally includes low pass filter, and this low pass filter stops the undesired signal of such as modulated numeral, IP and/or the control signal may disturbing Presentation Function 5130.Tap 513 also can comprise filter or the switch of the Signal transmissions stopped between modulator-demodulator 511 and 515.Such as, test modulator-demodulator can be 5131, it connects by tap 513 trouble shoot or the initial configuration that realize camera side modulator-demodulator 511, and display side modulator-demodulator 515 can be disconnected connection with interference and/or the deterioration of avoiding signal.As shown in Figure 5, SLOC camera side modulator-demodulator 511 also exports the lower passband QAM signal of the video camera generating portion based on signal 5102 usually except base-band CVBS signal 5100, and SLOC monitor side modulator-demodulator 515 exports the upper passband QAM signal based on the control signal in signal 5170.There is provided one or more filter to avoid visible not conforming on SD display 5130 and/or 516 need disturb and stop IP and control signal by tap 513.To understand, some displays and monitor lack higher frequency signals (relative to base band VCBS signal 5100) the necessary filtering stopped in passband signal.

Figure 51 B illustrates the SLOC system based on Fig. 3 institute interpretation system, cable 514 wherein between camera side and monitor side is temporarily disconnected in camera side, and SD display device or display 5130 are directly connected in SLOC camera side modulator-demodulator 511 in cable segmentation 519.Test modulator-demodulator 5131 connects alternatively for test/configuration purpose.SD display device 5130 shows base-band CVBS signal and provides supervision from the video near the video camera physical location of video camera 510, and may require that reconstructing these connections is beneficial to configuration and trouble shoot.In Figure 51 B, lower passband QAM signal 5102 may cause not conforming to the visual interference needed on the SD display 5130 lacking High frequency filter.

In the example shown in Figure 51 A and 51B, the part or all of disconnection of signal may be there is between modulator-demodulator 511 and 515.The part of signal disconnects and QAM signal transmission path can be made to keep complete.But some reconstruct of connection cause the disconnection of QAM signaling between camera side modulator-demodulator 511 and monitor side SLOC modulator-demodulator 515.Some embodiment of the present invention provides the mechanism making camera side modulator-demodulator 511 stop passband QAM to transmit, and when the connection between modulator-demodulator 511 and 515 disconnects, only exports CVBS signal.To understand, with test modulator-demodulator 5131 pairs of display side modulator-demodulators interim substitute generally include a series of, comprise: separated at modulator-demodulator 511 and 515, set up the connection between modulator-demodulator 511 and 5131, in the separated connection of newly setting up between modulator-demodulator 511 and 515 of laying equal stress on of modulator-demodulator 511 and 5131.The various functional parts of modulator-demodulator 511 can be used to detect the disconnection of QAM signal.Therefore, the operation of SLOC system is described in detail as follows.

The QAM modulation device framework of SLOC system

As mentioned above, Figure 19 explanation is supplied to the frame structure 1336 of passband modulation (PB modulation) module 1314 (see Figure 13).The trellis coding of Figure 16 adds bit; The data bit number of the QAM code element of each mapping is before trellis coding (as shown in table 2).The number of QAM code element that 315RS grouping (521640 bit) of Figure 14 maps to changes along with model selection.For every frame 207 and 315 RS packet sizes divided into groups, obtain whole every frame integer code element, as shown in table 3.PB modulation module 1314 uses those skilled in that art's known any proper method base band QAM code element to be modulated to passband (see the such as above explanation in conjunction with Figure 24) subsequently.

As mentioned above, in conjunction with Figure 20, the qam demodulator of Figure 21 and Figure 22 is described further.The data transmitted in module 2000 received passband signal also convert thereof into base band QAM code element.The operation performed by module 2000 generally includes chip clock synchronous, balanced (to remove intersymbol interference) and carrier auxiliary, and logical is use submodule.Correspondingly, module 2000 can comprise the equalizer of the base band QAM code element 2001 exported through recovering.Base band QAM signal 2001 is provided to secondary sheer 2018 to do cutting at real axis and imaginary axis direction, forms the sequence a being provided to frame synchronization module 2020 thus _r[k] ∈ [-1 ,+1] and a _i[k] ∈ [-1 ,+1] 2019.

Frame synchronization module 2020 performs continuous print cross-correlation operation to real part and imaginary part by the stored copies of binary frame synchronic PN sequence respectively in stacked cutting QAM code element 2019.Each member of stored copies has value-1 or+1.This computing provided by equation 1 is here repeated:

$b_R\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_R[n-k]$ And $b_I\left[k\right]=\underset{n=0}{\overset{126}{\mathrm{\&Sigma;}}}s\left[n\right]a_I[n-k]$ Equation 10

Wherein s is the stored copies in 127 long frame synchronization PN sequences.B _ror b _imaximum amplitude represent the beginning of FEC Frame.When FEC Frame initial point being detected in a stream, the one or more communications in frame-synchronizing impulse or other synchronizing signal and receiver module.

Figure 52 A and 52B illustrates the processes element that reliably can produce frame-synchronizing impulse when receiving band noise signal.Figure 52 A illustrates a part for the process determining frame length.Frame length can change according to selected transmission mode (table 3).The process starting from step 5200 is repeatedly executed along with symbol reception, and symbol counter keeps following the tracks of the code element number caused between each time of the value exceeding predetermined threshold execution.In step 5201, crosscorrelation is performed to each code element arrived, and symbol counter increases progressively, until determine to exceed predetermined threshold in step 5202.Symbol counter increases progressively 5203 to each code element, until exceed this threshold value.When exceeding this threshold value in step 5202, then symbol counter reset 5204 and perform crosscorrelation step 5205, symbol counter is increased progressively 5207, and new code element 5208 is received in multiple connection of laying equal stress on, until determine to exceed this threshold value in step 5206.Record intermediate symbol counting in step 5208 and reset symbol counter in step 5209.Crosscorrelation step 5210, the step 5212 that symbol counter is increased progressively and the step 5213 receiving new code element repeat, until determine to exceed this threshold value in step 5211.If in step 5214, when symbol counter counts identical with the intermediate symbol recorded in step 5208, then frame length returned value as symbol counter 5215.To understand, in described example, frame length can be determined after twice consistent continuously counting.But, required consecutive identical counts can be selected as required.

Figure 52 B illustrates a process, even if take this frame synchronization module 2020 to work as the frame-synchronizing impulse also producing correct timing when received signal has unusual large noise.This process changes the acquisition causing and also to provide during frame_size (frame sign) corresponding change new frame synchronization position when the temporary break of signal occurs or at transmitter sending mode.The symbol counter of free-running operation uses mould frame_size algorithm to the cell count received, and the step that wherein frame_size has been described by composition graphs 52A is determined.Estimate when the result of equation 10 crosscorrelation exceeds selected threshold value, symbol counter value will always have identical value.When this value is consistent, confidence counter is upwards incremented to selected maximum, such as maximum 16; Otherwise the minimum value of confidence counter towards zero is successively decreased.

Therefore, once receive code element 5250, just perform crosscorrelations 5251, if exceed threshold value in the result of 5252, then current maximum is set to threshold value 5253 and maximum point is set to the currency of symbol counter.In described example, if confidence counter be set to be at least 4 value (5254) and current symbol counting indicator frame synchronization point (5255), then in 5256 output frame synchronizing signals.Then, symbol counter increases progressively 5257, uses mould 4 to add here.Next code element is waited for, unless be confirmed as zero in step 5270 symbol counter in step 5277.If symbol counter is zero, then 5271, current maximum is reset.Then, if equal frame synchronization point at 5272 current maximum points, confidence counter increased progressively 5273 and waits for next code element in step 5277; Otherwise, to successively decrease confidence counter 5274.In the example illustrated at present, if determine that confidence falls less than 2 in step 5275, then in step 5276, frame synchronization point is set to current maximum point.Under arbitrary situation, wait for next code element in step 5277.

Generally speaking, according to described process, when confidence counter exceeds predetermined value (this value is 4 in this example), frame synchronization is confirmed as reliably obtaining.Then frame synchronization module can be cleared in the correct time to provide frame-synchronizing impulse.If confidence counter is more than 4, frame-synchronizing impulse is output---usually corresponding to the beginning of frame---in orthochronous, even if noise makes equation 10 produce a low value once in a while.

If sending mode changes, then confidence counter is back to zero by finally counting.This can be used to trigger determines that the frame length of new frame length calculates again (such as using the process of Figure 52 A).As discussed below in conjunction with carrier auxiliary, the ambiguity of pi/2 may be there is in exalted carrier phase place, this will cause 0, the recovery phase deviation additional arbitrarily of ± pi/2 or π.For frame synchronization code element, real part and imaginary part have identical symbol and send constellation and be illustrated in Figure 39.

Therefore, will understand, for zero phase skew, maximum amplitude b _rand b _isymbol be just.As summarized in table 5, the skew of-pi/2 will produce negative maximum amplitude b _rwith positive maximum amplitude b _i; For the skew of π, b _rand b _iboth are negative, and for the skew of pi/2, maximum amplitude b _rto be just maximum amplitude b _ito be negative.Therefore, maximum amplitude b _rand b _ieach symbol final phase deviation can be indicated together to have converged to which quadrant in complex plane.Additional phase correction is put on signal by this permission in phase offset correction device module 2002 (Figure 20).Maximum b _rand b _isymbol be sent to phase deviation connector 2002 by from based on relevant frame synchronization module 2020.

See also Figure 40, the operation in some of the phase offset correction device 2002 in Figure 20 example can be better understood.LUT400 produces based on maximum amplitude b _rand b _ithe output (see table 5) of symbol.Suppose z [k]=z _r[k]+jz _i[k], operation 142 performs as follows:

1) for the situation of φ=+ π: z ' [k]=-z _r[k]-jz _i[k]

2) for situation: z ' [k]=-z _i[k]+jz _r[k]

3) for situation: z ' [k]=+ z _i[k]-jz _r[k]

Once the synchronous starting position of locating frame and correct m pi/2 phase skew, just can know the position of the code word comprising mode bit (constellation and trellis code rate).By such as BCH decoder or by received code word is relevant to all possible code word and select the code word producing the highest end value reliably to decode to code word.Because this information is repeatedly transmitted, therefore by requiring that before acceptance same result repeatedly obtains additional reliability.Figure 41 illustrates an example of this kind of process that can be performed by frame synchronization module 2020.

Continue the system of Figure 20, the frame synchronizing signal 2021 exported from frame synchronization module 2020 can be used to instruction code element is fed to soft remove mapper before to remove which code element module 2004.In one example, from stream, remove 127 frame synchronization code elements and 8 pattern code elements, this guarantees only to be passed to softly to remove mapper 2006 by with RS corresponding code element of dividing into groups.The soft mapper 2006 that goes uses algorithm known in the art to calculate soft bit metric, and described algorithm comprises the algorithm such as described by Akay and Tosato.In order to correct computing, softly go mapper 2006 must know in transmitter to use which kind of perforation pattern (which kind of trellis code speed) and aiming at of this pattern and received bit will be known.Regardless of present mode why, this information 2021 is provided by frame synchronization module 2020, this frame synchronization module 2020 decoding schema information provide the repeating frame aimed at perforation pattern synchronizing signal in addition.These soft bit metric are fed to Viterbi decoder 2008, this Viterbi decoder 2008 with mode work known in the art with draw the PTCM encoder inputed in transmitter bit estimation.All by synchronous the going randomizer 2010, byte deinterleaver 2014 and RS decoder 2016 to go randomization respectively, to deinterleave and decoded word joint number obtains the data of the RS encoder entered at first in transmitter according to this of frame synchronizing signal 2010.

Stage switches

Some embodiment utilizes the stage based on the estimation of the mean square error of equalizer output to switch.A series of error e [k] that the accurate estimation of the mean square error (MSE) that equalizer exports can calculate from the Error Calculator module 422 by Figure 42 obtain.Such as, following formula can be used to obtain estimation:

MSE [k]=(1-β) e ²[k]+β MSE [k-1], (equation 18)

Wherein β < 1 is forgetting factor.Other method be averaging e [k] is known and can uses.Equation 18 produces a result, this result can compare with predetermined threshold and by the phase controller module 423 of Figure 42 use with when MSE [k] drops on below threshold value from 1 to stage of stage 2 handover operation.MSE [k] can be compared, when MSE [k] falls below Second Threshold, operation is switched to the stage 3 from the stage 2 with the second predetermined threshold.

Detect and disconnect and be connected again

Some embodiment is provided for detecting the system and method for disconnection in the camera side of communication link and connection event again.Refer again to Figure 51 A and 51B, the part or all of disconnection of the signal between modulator-demodulator 511 and 515 may occur in normal work.Some disconnects the QAM signaling affected between camera side modulator-demodulator 511 and monitor side SLOC modulator-demodulator 515.Specifically, the signal carrying the image of being caught by HD video camera 510 is encoded by modulator-demodulator 511 and/or modulates on cable 514, transfer to display side modulator-demodulator 515.The multiple method of the disconnection that detection associates with coaxial cable 514 and again connection event can be performed by camera side SLOC modulator-demodulator 511.In response to disconnection or connection event again, modulator-demodulator 511 can stop, starts or restart downstream passband QAM and transmit.In certain embodiments, " the coaxially connect " signal being sent to QAM modulation device from qam demodulator can be used to the transmission of control connection dependent event.

See Figure 53, such as camera side QAM modulation device 530 can be configured to only just send downstream passband signal 533 when the coaxial signal 531 connected is stated by camera side qam demodulator 532.Camera side qam demodulator 532 can determine the existence of input signal 534, and this input signal 534 uses multiple method to transmit by monitor side QAM modulation device (not shown).Usually, verifying when the reception of input signal 534 is reliably confirmed, when confirmation constellation mark and/or once obtain frame synchronization, stating the coaxial signal 531 connected by camera side qam demodulator 532.

A kind of method that detection input signal 534 exists comprises the method based on automatic growth control (AGC) ring.In the communication control processor comprising qam demodulator, the common AGC found is used for controlling the signal level of different phase in receiver and point.The example described in Figure 27, it illustrates the AGC loop 540 of the receiver front end being added into Figure 24.In this AGC loop 540, determine the amplitude of complex signal 541 and 542, it deducted from predetermined reference level 543.By low pass filter (LPF) 544, filtering is carried out with restraint speckle and short term variations to result.LPF544 provides output, and this output is fed to the accumulator comprising adder 545 and delay cell 546.Accumulator exports and is used as gain control signal 547, and this gain control signal 547 is fed the gain block 548 that the system of getting back to inputs 549 places.In one example, gain control signal 547 is used as gain factor or multiplier, and the gain being used for determining to be provided by gain block 548 increases in predetermined threshold to make the gain provided by gain block 548 control 547 increases along with gain.When input 549 disconnects (such as coaxial cable disconnection), the output of amplitude block 541 is often very low.Usually, the coaxial signal 531 connected only exports at amplitude block and is just declared higher than when predetermined threshold.In addition, when input 549 disconnects, gain control signal 547 is usually very high.Therefore, the coaxial signal 531 connected only just is declared lower than when predetermined threshold at gain control signal.AGC loop 540 can be used to the connection status monitoring input 549, even if other places find loops in qam demodulator 532.

Detect the other method of input signal 534 existence based on the equalizer shown in Figure 43 and carrier phase/frequency loop order section (separately seeing equation 18).Specifically, when the QAM modulation device phase controller 434 (at the beginning in the stage 1) of qam demodulator 532 switches to the stage 2 based on the result of equation 18, the coaxial signal 531 connected can be stated.Only at connecting coaxial cable and qam demodulator 532 initiatively receives stream signal from monitor side QAM modulation device time, the transition in 1 to stage of stage 2 occurs.Any disconnection afterwards of coaxial cable by the increase of MSE causing the loss of signal, calculated by equation 18, and can cause to the stage 1 repeatedly.When qam demodulator 532 is in the stage 1, the signal 531 coaxially connected can be reset or otherwise removes statement.In certain embodiments, may require that camera side qam demodulator 532 had arrived the stage 3 before the coaxial signal 531 connected of statement.

The another kind of method detecting input signal 534 existence is the demodulator frame synchronization confidence counter discussed based on composition graphs 52B.Specifically, only when confidence counter deposit one be greater than the value of predetermined threshold time, the coaxial signal 531 connected just can be stated by camera side qam demodulator 532.In one example, threshold value can be 4.Therefore, only when connecting coaxial cable and SLOC frame is sent to video camera by monitor side modulator-demodulator time, the coaxial signal 531 connected just is declared.Even without receiving code element, if frame synchronizing process continues freely to carry out, then disconnection can make confidence counter counting in reverse and finally make it lower than 4, and removes the statement to coaxial connection signal 531.

The other method detecting input signal 534 existence is based on higher level protocol.Refer again to Figure 51 A, use gateway protocol communicates with monitor side host computer system 38 by HD video camera 30.For this discussion object, ubiquitous Internet protocol (IP) will be used as an example of gateway protocol.Some patterns of IP are intrinsic two-way and cause upstream sending data with downstream.If cable disconnects, the network controller in HD video camera 30 and/or modulator-demodulator 32 or processor find that the IP grouping do not returned just arrives from monitor side also can notify that camera side SLOC modulator-demodulator 32 stops pass band transfer.In one example, these notices can comprise, by the MII interface 536 such as shown in Figure 53, packet predetermined specially transferred to modulator-demodulator 32 from HD video camera 30.

The supplemental instruction of some aspect of the present invention

Above-mentioned explanation of the present invention is intended to be illustrative but not determinate.Such as, those skilled in that art will understand, and the present invention realizes by the various combinations of above-mentioned functions and ability, and can comprise than above-mentioned parts less or more.Set forth some additional aspect of the present invention and feature below, and these additional aspect and feature can use above function in greater detail and parts to obtain, as can be understood after being subject to disclosure teaching in those skilled in that art.

Some embodiment of the present invention provides the system and method associated with video camera.Some in these embodiments comprise: processor, and this processor receives picture signal from imageing sensor and produces multiple vision signals of token image signal; And encoder, baseband video signal and digital video signal are combined into the output signal transmitted on cable by this encoder.In in these embodiments some, vision signal comprises baseband video signal and digital video signal.In in these embodiments some, through the base band of combination and digital video signal be substantially etc. time.In in these embodiments some, video camera is closed circuit HDTV (High-Definition Television) video camera.In in these embodiments some, baseband video signal comprises SD analog video signal.In in these embodiments some, digital video signal is modulated before combining with baseband video signal.In in these embodiments some, digital video signal comprises compressed digital video.In in these embodiments some, digital video signal is high-definition digital video signal.In in these embodiments some, the frame rate of digital video signal is lower than the frame rate of picture signal.In in these embodiments some, modulated digital signal is provided to video recorder.

Some in these embodiments comprise the decoder being configured to the stream signal from cable reception be carried out to demodulation.In in these embodiments some, the stream signal through demodulation comprises control signal.In in these embodiments some, control signal comprise control video camera position and towards signal.In in these embodiments some, control signal comprises the signal being controlled baseband video signal and digital video signal generation by processor.In in these embodiments some, control signal comprises the signal selecting a part for picture signal to encode as baseband video signal.In in these embodiments some, control signal comprises the signal of a part as encoding digital video signals of selection picture signal.In in these embodiments some, the stream signal through demodulation comprises the audio signal that the audio frequency for actuated camera exports.

Some embodiment of the present invention provides the system and method for transmitting video image.Some in these embodiments comprise: make frequency division multiplexing to obtain modulated digital signal to the vision signal received from high resolution imaging equipment; By modulated digital signal and the base-band analog signal characterizing vision signal are combined to produce output signal; And output signal is transferred to display system and digital video capture and/or memory device simultaneously.In in these embodiments some, display system display characterizes the image of deriving from the baseband analog of vision signal.In in these embodiments some, Digital video storage uses digital video recordings device to record from a series of high definition frames extracted through modulated digital signal.

Some in these embodiments comprise compressed video signal.In in these embodiments some, the step of frequency division multiplexing digital video signal comprises compressed video signal before modulation.In in these embodiments some, send output signal and comprise output signal is supplied to coaxial cable.Some in these embodiments comprise input signal that demodulation receives from coaxial cable to obtain control signal.Some in these embodiments comprise by a part of Video signal encoding is produced base-band analog signal in composite video signal.Some in these embodiments comprise the part vision signal using control signal selection plan to be coded in composite video signal.Some in these embodiments comprise the position using control signal to control video camera.In in these embodiments some, demodulation input signal comprises from input signal extraction audio signal.

Some embodiment of the present invention provides the system and method for running video camera.Some in these embodiments comprise: processor, and this processor receives picture signal from imageing sensor and produces multiple vision signal; Control logic, the control signal that this control logic is configured to being received by video camera responds; And modulator, this modulator is configured to modulation digital vision signal as modulated signal.In in these embodiments some, multiple vision signal comprises baseband video signal and digital video signal.In in these embodiments some, in each the sign camera field in multiple vision signal at least partially.In in these embodiments some, control signal controls the content of base band and digital video signal.In in these embodiments some, transmitted by video camera through modulation signal and baseband video signal simultaneously.

In in these embodiments some, base band and digital video signal be substantially etc. time.Some in these embodiments comprise encoder, and this encoder is by baseband video signal and be combined in the output signal that cable transmits through modulation signal.In in these embodiments some, control signal is received as wireless signal.In in these embodiments some, modulated signal is wirelessly transmitted.In in these embodiments some, digital video signal is high-definition digital video signal.In in these embodiments some, digital video signal comprises compressed digital video.In in these embodiments some, control signal moves by the ken part characterized of in vision signal.

Some embodiment of the present invention is provided for the equalizer of digital signal and base-band analog signal, and described digital signal and base-band analog signal are by frequency separation and carried by cable.Some in these embodiments comprise the digital equalizer removing distortion the digital signal received from receiver side.Some in these embodiments comprise the analog equalizer compensating the analog signal decay caused by cable.In in these embodiments some, analog equalizer applies in one group of baseband analog filter one with compensate for attenuation.In in these embodiments some, the baseband analog filter applied selects based on the estimation of the decay difference at different frequencies calculated by digital equalizer.

In in these embodiments some, digital signal and analog signal are transmitted between the transmitter and receiver being presented as video camera, and wherein the equalizing signal characterized as analog signal is supplied to monitor by receiver.In in these embodiments some, cable comprises coaxial cable.In in these embodiments some, distortion increases along with cable length.In in these embodiments some, distortion comprises multipath distortion.In in these embodiments some, the estimation of decay calculates from the frequency range with crooked power spectral density roughly linearly.In in these embodiments some, crooked is use to calculate the fast fourier transform of multiple filter tap.In in these embodiments some, select the frequency gap in frequency range to allow to use summation to calculate the frequency response of the filter of digital equalizer:

$\begin{array}{c}{G}_R\left[k_1\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_R\left[4n\right]+\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_I[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_R[4n+2]-\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_I[4n+3]\\ G_I\left[k_1\right]=\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_I\left[4n\right]-\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_R[4n+1]-\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_I[4n+2]+\underset{n=0}{\overset{N/4-1}{\mathrm{\&Sigma;}}}{g}_R[4n+3],\end{array}$ Wherein

G [k] is the discrete Fourier transform (DFT) of the equalizer filter tap of time domain convergence, and k ₁corresponding to the certain tones gap of DFT.In in these embodiments some, the high definition that digital signal comprises the video image of being caught by video camera characterizes, and the SD that analog signal comprises video image characterizes.

Some embodiment of the present invention provides and makes to be carried through the frequency method balanced with the analog signal in the cable of the digital signal that analog signal is separated equally.In in these embodiments some, the method is by modulator-demodulator confidence, and this modulator-demodulator receives analog and digital signal and exports baseband video signal.Some in these embodiments comprise that to calculate in digital signal crooked.In in these embodiments some, the decay that crooked sign is function with the frequency owing to cable.Some crooked digital signals that make comprised based on calculating in these embodiments are balanced.Some in these embodiments comprise and use the crooked analog equalizer that configures calculated to select in one group of baseband analog filter.Some baseband analog filter comprised selected by use in these embodiments carry out balance simulation signal.

In in these embodiments some, analog signal comprises baseband video signal and digital signal comprises the high definition version of baseband video signal.In in these embodiments some, cable comprises coaxial cable, and wherein crookedly changes along with cable length.In in these embodiments some, be crookedly derived from multipath distortion.In in these embodiments some, calculate the crooked decay estimated in the frequency range with crooked power spectral density roughly linearly that comprises.In in these embodiments some, estimation decay comprises the fast fourier transform used for multiple filter tap.In in these embodiments some, estimation decay comprises the several frequency gaps selected in frequency range.In in these embodiments some, the efficiency optimization of the step that selected frequency gap makes calculating crooked.

Some embodiment of the present invention provides the digital communication system using novel framing structure.Some in these embodiments comprise the convolutional byte interleaver of interleaving data frame, and wherein this interleaver is synchronous with frame structure.Some in these embodiments comprise the randomizer being configured to produce randomized Frame from the Frame through interweaving.Some in these embodiments comprise the perforation trellis coded modulation device be operated under optional bit rate, and described perforation trellis coded modulation device is from the Frame produced through randomized Frame through trellis coding.Some in these embodiments comprise QAM mapper, and the many groups bit mapping in the Frame of trellis coding to modulated symbol, is provided the frame through mapping by this QAM mapper thus.Some in these embodiments comprise the synchronizer be added into synchronously dividing into groups through mapping frame.

In in these embodiments some, perforation trellis coded modulation device is bypassed to obtain the optimization net bit rate based on the system white noise performance recorded.In in these embodiments some, identical synchronous grouping is added into each in a series of successor map frame.In in these embodiments some, identical synchronous grouping is added into each frame through mapping.In in these embodiments some, the synchronous grouping of a part comprises 127 code elements.In in these embodiments some, a part is synchronously divided into groups the different binary sequences of the real part that comprises for modulated symbol and imaginary part.In in these embodiments some, the synchronous grouping of a part comprises the identical binary sequence of the real part for modulated symbol and imaginary part.In in these embodiments some, synchronous grouping comprises instruction through mapping the data of the transmission mode of frame.In in these embodiments some, the instruction of transmission mode comprises selected qam constellation and selected trellis code speed.In in these embodiments some, system is regardless of transmission mode why each Frame produces constant integer Reed-Solomon grouping and.In in these embodiments some, system is regardless of transmission mode why each Frame produces a variable integer modulated symbol and.In in these embodiments some, regardless of transmission mode why system produces integer perforation pattern cycle and to each Frame.

Some embodiment of the present invention provides the framing method of variable net bit rate digital communication system.Some in these embodiments comprise provides a different set of quadrature amplitude modulation (QAM) constellation.Some in these embodiments comprise the frame using the combination of the trellis code of perforation to produce packet, and often kind of combination is corresponding to the pattern be associated.Some in these embodiments comprise the frame providing and have a variable integer QAM code element.In in these embodiments some, QAM code element number corresponds to selected pattern.In in these embodiments some, the byte that every frame is associated and Reed-Solomon grouping number are constant.In in these embodiments some, the frame using the combination of perforation trellis code to produce packet comprised each Frame generation integer perforation pattern cycle, and the pattern no matter be associated why.In in these embodiments some, the data bit number of each QAM code element is marks for one or more patterns.In in these embodiments some, for all patterns, the number in the grid encoder perforation pattern cycle of every frame is an integer.

The system that some embodiment of the present invention provides phase calibration to offset.Some in these embodiments comprise phase offset correction device, this phase offset correction device receive characterize through quadrature amplitude modulation signal derive the signal of phasing through equalizing signal from through equalizing signal.Some in these embodiments comprise by through equalizing signal cutting to obtain the secondary sheer of real number and imaginary number sequence.Some in these embodiments comprise frame synchronizer, and it is relevant that this frame synchronizer performs the appropriate section of real number and imaginary number sequence and stored frame synchronization pseudo random sequence.Some in these embodiments comprise the phase correction signal being supplied to phase offset correction device by frame synchronizer.In in these embodiments some, phase correction signal is based on relevant maximum real number and imaginary value.In in these embodiments some, frame synchronizer performs continuous print crosscorrelation to the quadrature amplitude modulation code element through cutting imported into.

In in these embodiments some, continuous crosscorrelation uses the stored copies of the synchronous PN (pseudo noise) sequence of binary frame to carry out for real number and imaginary number sequence respectively.In in these embodiments some, the signal through quadrature amplitude modulation uses perforation trellis coded modulation.In in these embodiments some, the signal through quadrature amplitude modulation uses Quadrature Phase Shift Keying modulation to modulate.In in these embodiments some, the signal through quadrature amplitude modulation (QAM) uses 16-QAM modulation.In in these embodiments some, the signal through quadrature amplitude modulation (QAM) uses 64-QAM modulation.In in these embodiments some, the frame synchronization code element through the signal of quadrature amplitude modulation has identical symbol, and the symbol instruction of relevant maximum real number and imaginary value is through the phase rotating of equalizing signal.In in these embodiments some, the phase correction signal provided by frame synchronizer comprises relevant maximum real number and the symbol of imaginary value.In in these embodiments some, phase offset correction device has the symbol of relevant maximum real number and imaginary value look-up table by index is derived through phase correction signal, thus determines phase correcting value.

Some embodiment of the present invention provides the method for the carrier phase offset corrected in receiver in the signal of quadrature amplitude modulation.Some in these embodiments comprise carries out equilibrium to signal.Some in these embodiments comprise the signal of cutting through equilibrium, thus from through the signal acquisition real number of equilibrium and imaginary number sequence.Some in these embodiments comprise the frame synchronization sequence identified in real number and imaginary number sequence.In in these embodiments some, identify that frame synchronization sequence comprises stored pseudo random sequence relevant with imaginary number sequence to real number.In in these embodiments some, identify that frame synchronization sequence comprises from the maximum related value determination frame be associated with real number and imaginary number sequence.Some in these embodiments comprise the phase error corrected in equalizing signal based on maximum correction.

In in these embodiments some, the storage version that correlation step comprises the synchronous PN (pseudo noise) sequence of use binary frame performs continuous crosscorrelation to a succession of quadrature amplitude modulation code element through cutting.In in these embodiments some, correlation step comprises and performs continuous crosscorrelation with the storage version of real number and imaginary number sequence pair frame synchronization sequence respectively.In in these embodiments some, the frame synchronization code element of frame synchronization sequence has identical symbol.In in these embodiments some, the symbol that phase calibration error comprises based on maximum related value determines the phase rotating in equalizing signal.In in these embodiments some, the phase error corrected in equalizing signal comprises carrys out index look-up table with the symbol of real number and imaginary number maximum related value.

Some embodiment of the present invention is provided for the method for the carrier phase offset in correction of orthogonal amplitude-modulated signal.In in these embodiments some, these methods can realize comprising one or more being configured in the system of the processor performing instruction.Some in these embodiments comprise the instruction performing on the one or more processors and be configured to make signal equalization.Some in these embodiments comprise performing on the one or more processors and are configured to cutting through equalizing signal thus from the instruction obtaining real number and imaginary number sequence through equalizing signal.Some in these embodiments comprise the instruction performing on the one or more processors and be configured to the frame synchronization sequence identified in real number and imaginary number sequence.In in these embodiments some, identify that frame synchronization sequence comprises and perform continuous print crosscorrelation with the storage version of real number and imaginary number sequence pair frame synchronization sequence respectively.In in these embodiments some, identify that frame synchronization sequence comprises from the maximum related value determination frame be associated with real number and imaginary number sequence.Some in these embodiments comprise the instruction performing on the one or more processors and be configured to correct the phase error in equalizing signal based on maximum related value.In in these embodiments some, the frame synchronization code element of frame synchronization sequence has identical symbol.In in these embodiments some, the symbol that phase calibration error comprises based on maximum related value determines the phase rotating in equalizing signal.

Some embodiment of the present invention provides the method for the constellation identifying code element.In in these embodiments some, the method is performed by one or more processors of the communication system of multimode quadrature amplitude modulation.Some in these embodiments comprise the instruction performing and the power in one or more processor characterization signal is distributed.In in these embodiments some, the generation of the power level detected in power distribution statistics ground tracking signal.Some in these embodiments comprise the instruction performing and one or more peak values of the power level in the distribution of one or more processor determination power are occurred.Some in these embodiments comprise performing makes one or more processor determine the instruction of constellation based on the distribution that peak value occurs.

In in these embodiments some, constellation is determined in the expansion that one or more processor also occurs based on one or more peak value.In in these embodiments some, signal is through equalizing signal and described one or more processor determines constellation by the multiple sections in the distribution map of inspection power distribution.In in these embodiments some, each section corresponds to in multiple constellation candidate but the power level range that is associated of not all constellation candidate.In in these embodiments some, multiple constellation candidate comprises quadrature phase keying constellation and quadrature amplitude modulation (QAM) constellation.In in these embodiments some, multiple constellation candidate comprises 16-QAM and 64-QAM constellation.In in these embodiments some, multiple constellation candidate comprises 256-QAM constellation.

Some in these embodiments comprise performing makes one or more processor set up the instruction of the reliability of identified constellation by the step of each judgement in performing a Continual constellation and judging.In in these embodiments some, these steps comprise makes counter increment when judgement subsequently confirms the identity of constellation.In in these embodiments some, these steps comprise makes counter decrement when judgement identification various constellations subsequently.In in these embodiments some, these steps value comprised based on counter provides the measurement of reliability.In in these embodiments some, when counter exceeds a threshold value, constellation is identified.In in these embodiments some, for each in multiple constellation candidate provides a counter, and identify this constellation when its corresponding counter exceeds a threshold value.In in these embodiments some, there is the corner code element corresponding to constellation in the peak value of power level.In in these embodiments some, before equalizing signal, identify constellation.

Some embodiment of the present invention is provided for the method for the constellation of the code element identified in the communication system of multimode quadrature amplitude modulation.In in these embodiments some, these methods are performed by the processor in the modulator-demodulator of communication system.Some in these embodiments comprise the instruction performing and make processor extract pattern information from Frame in response to the beginning detected at the Frame of modulator-demodulator place reception.Some in these embodiments comprise performing makes processor pass through to select the code of the corresponding code in closest match pattern bit from multiple potential constellation code and the instruction of determining current constellation.If some in these embodiments comprise current constellation and mate with the constellation determined before, perform and make processor increase the instruction of the confidence measure be associated with the constellation identified before.If some in these embodiments comprise, to perform current constellation different from the constellation identified before, perform and make processor minimizing confidence measure and the instruction identifying constellation before being recorded as by current constellation.Some in these embodiments comprise repeating to be made processor extract pattern information, selects current constellation and for follow-up Frame adjustment confidence measure until confidence measure exceeds the step of predetermined threshold.In in these embodiments some, when confidence measure exceeds predetermined threshold value, constellation is identified.

In in these embodiments some, constellation code is selected to comprise to make processor to perform the cross correlation of each and corresponding code bit in multiple potential constellation code.In in these embodiments some, in non-equalizing signal, identify constellation, this non-equalizing signal carries this Frame and follow-up Frame.In in these embodiments some, at processor from while signal recuperation carrier wave, constellation is identified.Some in these embodiments comprise execution makes processor use constant modulo n arithmetic (CMA) to carry out error signal to make equalizer filter tap restrain thus to allow the instruction of signal equalization.In in these embodiments some, the CMA parameter through convergent-divergent is used to carry out error signal to improve equalization performance.In in these embodiments some, the equilibrium of executive signal comprises the distribution map analyzed through the power of equalizing signal.In in these embodiments some, analysis distribution figure comprises probability of use mass function.In in these embodiments some, executive signal equilibrium comprises performing makes processor calculate and the instruction of the multiple code elements in equalizing signal to the power associated.In in these embodiments some, executive signal equilibrium comprises performing makes the instruction of processor by using threshold power level to identify the corner code element of constellation.In in these embodiments some, the identity of threshold power level instruction constellation.

Some embodiment of the present invention is provided for the system of transmission video signal, this system comprises camera side modulator-demodulator, this camera side modem configuration becomes to receive two signals from video camera, the image sequence that each characterization is caught by video camera, this camera side modulator-demodulator is configured to two signal Zhong mono-tunnels another signal is modulated as with the nonoverlapping passband vision signal of baseband signal and transmitted number as composite baseband video Signal transmissions further.In in these embodiments some, camera side modulator-demodulator comprise combination base band and passband vision signal to provide the blender of signal transmission.In in these embodiments some, camera side modulator-demodulator comprises and is configured to send signal transmission on transmission line and extracts the duplexer of the passband signal received from transmission line.In in these embodiments some, camera side modulator-demodulator comprises surveillance camera side modulator-demodulator and produces the detector enabling signal when the passband signal received is identified.In in these embodiments some, enable the transmission of at least one in signal controlling baseband video signal and passband vision signal.

In in these embodiments some, passband vision signal is only just transmitted when signal is enabled in generation.In in these embodiments some, the passband signal received is by quadrature amplitude modulation.In in these embodiments some, the estimation of the mean square error in detector monitors quadrature amplitude demodulation device, and signal is enabled in generation when this estimation exceeds threshold value.In in these embodiments some, detector monitors constellation detector.In in these embodiments some, enabling signal is produce based on the reliability measurement provided by constellation detector.In in these embodiments some, reliability measurement is based on frame synchronization sequence.In in these embodiments some, the valuation of the mean square error in detector monitors equalizer.In in these embodiments some, when valuation exceeds threshold value, produce and enable signal.In in these embodiments some, the gain factor in the automatic growth control module of detector monitors camera side modulator-demodulator.In in these embodiments some, when gain factor has the value being less than threshold value, produce and enable signal.In in these embodiments some, the amplitude of the passband signal that detector monitors receives.In in these embodiments some, when amplitude has the value exceeding threshold value, produce and enable signal.In in these embodiments some, the passband signal received comprises the data according to Internet protocol coding.

Some embodiment of the present invention is provided for the method for the signaling controlled in safety system.Some in these embodiments are included in the existence that upstream modem place determines the composite signal middle and upper reaches QAM signal of transmission over coaxial cable.Some in these embodiments comprise when determining that upstream QAM signal makes upstream modem send composite baseband video signal and passband vision signal on coax when existing.In in these embodiments some, composite baseband video signal and passband vision signal are the parallel signs of a sequence image of being caught by video camera.Some in these embodiments comprise when determining that upstream QAM signal makes upstream modem send composite baseband video signal on coax when not existing and stops the transmission of passband vision signal.

In in these embodiments some, when the yield value in automatic gaining controling signal exceeds threshold value, upstream QAM signal is confirmed as to be existed.In in these embodiments some, when the amplitude measurement of upstream QAM signal is less than threshold value, upstream QAM signal is confirmed as to be existed.In in these embodiments some, when the estimation of the mean square error in equalizer exceeds threshold value, upstream QAM signal is confirmed as not to be existed.In in these embodiments some, when identifying that internet protocol datagram divides into groups in the QAM signal of upstream, upstream QAM signal is confirmed as not to be existed.

Some embodiment of the present invention is provided for the automatic reconfiguration system of transmission video signal.Some in these embodiments comprise the upstream modem being configured to receive two signals from video camera.In in these embodiments some, the image sequence that each characterization is caught by video camera.In in these embodiments some, upstream modem is configured to a signal in two signals as composite baseband video Signal transmissions another signal to be modulated as with the nonoverlapping passband vision signal of baseband signal and sent.Some in these embodiments comprise downstream modulator-demodulator, and this downstream modem configuration becomes from upstream modem reception composite baseband video signal and passband vision signal and is configured to upstream passband signal to transfer to upstream modem further.In in these embodiments some, when upstream passband signal deteriorated being detected, upstream modem stops the transmission of at least one signal in two signals.

Although with reference to certain exemplary embodiments, invention has been described, but obviously, for those of ordinary skill in the art, can make various modifications and variations to these embodiments and do not depart from the wider spirit and scope of the present invention.Such as, described compressed digital HD video and the system of baseband analog video signals are provided.Other embodiments of the invention provide the feeding of SD Digital and analog simultaneously.Other embodiment is providing full motion digital HD video together with baseband analog video.Therefore, this specification and accompanying drawing are considered to illustrative but not restrictive, sense.

Claims (28)

1. a video camera, comprising:

Imageing sensor;

Processor, described processor receives picture signal from described imageing sensor and produces the multiple vision signals characterizing described picture signal, and described vision signal comprises analog video signal and digital video signal; And

Modulator-demodulator, described digital video signal modulated by described modulator-demodulator, to obtain the modulated digital video signal centered by carrier frequency,

Wherein modulated digital video signal transmits on coax,

Wherein said analog video signal transmits in the base band of described coaxial cable as baseband video signal, and the display system that described baseband video signal can be connected to coaxial cable receives and utilized low pass filter to extract by described display system, and

Wherein the frequency of modulated digital video signal exceedes the highest frequency of baseband video signal.

2. video camera as claimed in claim 1, is characterized in that, described baseband video signal and digital video signal be substantially etc. time.

3. video camera as claimed in claim 1, it is characterized in that, described baseband video signal comprises SD analog video signal.

4. video camera as claimed in claim 1, it is characterized in that, digital video signal is modulated in the frequency range be separated with the highest frequency of baseband video signal by described modulator-demodulator, and wherein video camera also comprises encoder, modulated digital video signal and baseband video signal combine to transmit on coax by described encoder.

5. video camera as claimed in claim 4, it is characterized in that, described digital video signal is high-definition digital video signal.

6. video camera as claimed in claim 4, is characterized in that, the frame rate of described digital video signal is lower than the frame rate of described picture signal.

7. video camera as claimed in claim 4, it is characterized in that, modulated digital video signal is provided to video capture device.

8. video camera as claimed in claim 1, is characterized in that, also comprise decoder that the stream signal that described decoder configurations becomes demodulation to receive from described coaxial cable wherein comprises control signal through demodulation stream signal.

9. video camera as claimed in claim 8, is characterized in that, described control signal comprise position for controlling described video camera and towards signal.

10. video camera as claimed in claim 8, is characterized in that, described control signal comprises the signal for being controlled described baseband video signal and the generation of described digital video signal by described processor.

11. video cameras as claimed in claim 10, it is characterized in that, described control signal comprises the signal for selecting a part for described picture signal to encode as described baseband video signal.

12. video cameras as claimed in claim 10, is characterized in that, described control signal comprises for selecting a part for described picture signal as the signal of described encoding digital video signals.

13. video cameras as claimed in claim 8, is characterized in that, describedly comprise audio signal for driving the audio frequency of described video camera to export through demodulation stream signal.

14. 1 kinds, for sending the method for video image, comprising:

By the vision signal frequency division multiplexing received from video imaging apparatus to obtain through modulated digital signal;

By by described through modulated digital signal with characterize the base-band analog signal of described vision signal and combine and produce and output signal; And

What described output signal transferred in digital video capture device and display system is one or more,

Wherein transmit described output signal to comprise described output signal is supplied to coaxial cable,

The video image of wherein deriving from the baseband portion of transmitted output signal directly can be shown by display system.

15. methods as claimed in claim 14, is characterized in that, described digital video capture device is recorded from the described frame sequence extracted through modulated digital signal.

16. methods as claimed in claim 14, it is characterized in that, described digital video capture device comprises video server.

17. methods as claimed in claim 14, comprise input signal that demodulation receives from described coaxial cable further to obtain control signal.

18. methods as claimed in claim 17, comprise further:

Described base-band analog signal is produced by a part for described vision signal being coded in composite video signal; And

Described control signal is used to select to be coded in the part vision signal in described composite video signal.

19. methods as claimed in claim 17, comprise further and use described control signal to control the position of video camera.

20. methods as claimed in claim 17, is characterized in that, input signal described in demodulation comprises extract audio signal from described input signal.

21. methods as claimed in claim 14, it is characterized in that, described in frequency division multiplexing, digital video signal comprises: compress described vision signal; And on carrier wave, modulate compressed video signal.

22. methods as claimed in claim 14, is characterized in that, described baseband portion utilizes low pass filter to extract to remove all signals beyond baseband signal.

23. 1 kinds of video cameras, comprising:

Processor, described processor receives picture signal from imageing sensor and produces multiple vision signal, each in described multiple vision signal characterizes described camera field at least partially, and described multiple vision signal comprises baseband video signal and digital video signal;

Control logic, described control logic is configured to the control signal in response to being received by described video camera, and wherein said control signal controls the content of described base band and digital video signal; And

Modulator, described modulator is configured to modulate described digital video signal, wherein modulated digital video signal transmits on cable, and baseband video signal transmits in a base band on cable, wherein the frequency of modulated digital video signal exceedes the highest frequency of baseband video signal, wherein modulated digital video signal and described baseband video signal simultaneously by described video camera by cable transmission to display device, recording arrangement and video server.

24. video cameras as claimed in claim 23, is characterized in that, described control signal moves the ken part represented by least one in multiple vision signal.

25. video cameras as claimed in claim 23, is characterized in that, described control signal is received as wireless signal.

26. video cameras as claimed in claim 23, it is characterized in that, described modulated digital video signal is wirelessly transmitted.

27. video cameras as claimed in claim 23, is characterized in that, described base band and digital video signal be substantially etc. time.

28. video cameras as claimed in claim 23, is characterized in that, also comprise encoder, described baseband video signal and described modulated digital video signal are combined as the output signal transmitted on cable by described encoder.