KR101297453B1 - Motor control device and air conditioner - Google Patents

Abstract

An object of the present invention is to provide a motor control device that can accurately perform high efficiency operation.
When the three-phase inverter circuit 17A having the IGBT on the upper arm and the MOSFET on the lower arm performs PWM control, when the modulation rate of the three-phase inverter circuit 17A is lower than the predetermined threshold value, two of the lower arms are used. Phase modulation (lower two-phase modulation) is performed. If the modulation rate is higher than a predetermined threshold, two-phase modulation (upper two-phase modulation) of the upper arm is performed. If the modulation rate is at an intermediate level, Two-phase modulation (up and down two-phase modulation) is performed. As a result, when the modulation rate is low, the lower two-phase modulation allows current to flow through the MOSFET of the lower arm, so that the conduction loss can be reduced and the operation efficiency of the three-phase inverter circuit 17A can be improved. In addition, when the modulation rate is higher than the predetermined threshold value, by performing upper two-phase modulation, an increase in conduction loss can be suppressed and a decrease in the operating efficiency of the three-phase inverter circuit 17A can be suppressed.

Description

Motor Controls and Air Conditioners {MOTOR CONTROL DEVICE AND AIR CONDITIONER}

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a motor control apparatus and an air conditioner for driving control of a three-phase synchronous motor using a three-phase inverter circuit for converting DC power into three-phase AC power.

In a device equipped with a motor as a load, resource saving and energy saving have been strongly demanded from recent requests for global environmental conservation. In order to meet such a demand, the technique of carrying out the drive control of a three-phase synchronous motor using the three-phase inverter circuit which converts DC power into three-phase AC power is used widely.

Patent Document 1 has a three-phase series circuit of IGBTs and MOSFETs as switching elements, and includes a switching circuit in which interconnection points of IGBTs and MOSFETs in these series circuits are connected to a load, and a control unit for PWM control of the switching circuit. A three-phase inverter device is disclosed.

In the technique related to Patent Document 1, the control unit is a two-phase in which one switching element of each of the two series circuits of each of the three-phase series circuits is turned on and off, and the other switching element of the remaining one series circuit is turned on. Any one of energization or three-phase energization in which one switching element of each series circuit for three phases is turned on and off in a different phase and the other switching element is turned on and off in reverse phase Optionally, follow the instructions below.

According to the technique of patent document 1, even if IGBT and MOSFET are used as a switching element of the upper arm and lower arm which comprise a three-phase inverter circuit, either two-phase energization or three-phase energization can be used. By selectively switching and executing according to the above, the operation efficiency can be improved.

Japanese Patent Application Publication No. 2008-104282

However, in the technique concerning patent document 1, either two-phase electricity supply or three-phase electricity supply is selectively performed according to the height of a load. For this reason, even when low load is high, there exists a possibility that the performance of high efficiency operation may be impaired when ambient temperature is high.

In addition, in the technique according to Patent Document 1, no consideration is given to the modulation rate indicating the ratio of the amplitude value of the AC output voltage to the DC input voltage of the three-phase inverter circuit performing PWM control. For this reason, for example, when the modulation rate is large, there is a fear that the switching loss of the switching element is increased, which may interfere with the performance of high efficiency operation.

The present invention has been made to solve the above problems, and an object of the present invention is to provide a motor control device and an air conditioner capable of accurately performing high efficiency operation.

MEANS TO SOLVE THE PROBLEM In order to solve the said subject, the motor control apparatus which concerns on this invention is a motor control apparatus which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power. A temperature detector for detecting a temperature of the switching element, a current detector for detecting a circuit current flowing through the three-phase inverter circuit, a temperature of the element detected by the temperature detector and the circuit current detected by the current detector. It is a main feature to have a modulation scheme control section for performing control using an based modulation scheme.

Moreover, the motor control apparatus which concerns on this invention is a motor control apparatus which carries out drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power, The said three-phase inverter A current detector for detecting a circuit current flowing through the circuit, a voltage detector for detecting a DC voltage applied to an input side of the three-phase inverter circuit, the circuit current detected by the current detector and the DC voltage detected by the voltage detector. A modulation rate calculation unit for calculating a modulation rate indicating a ratio of an amplitude value of an AC voltage applied to the three-phase motor with respect to the DC voltage, and a modulation scheme based on the modulation rate calculated by the modulation rate calculation unit. It is a main feature to have a modulation system control unit that performs control using.

Moreover, the motor control apparatus which concerns on this invention is a motor control apparatus which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into three-phase AC power, A temperature detector for detecting a temperature, a current detector for detecting a circuit current flowing through the three-phase inverter circuit, a voltage detector for detecting a DC voltage applied to an input side of the three-phase inverter circuit, and the current detector detected the A modulation rate calculator for calculating a modulation rate indicating a ratio of an amplitude value of an AC voltage applied to the three-phase motor to the three-phase motor based on a circuit current and the DC voltage detected by the voltage detector; and the temperature detector A modulation scheme based on the temperature of the element detected by the controller and the modulation rate calculated by the modulation rate calculator; In that it comprises a modulation scheme and a control section which performs the essential characteristics thereof.

According to the motor control apparatus according to the present invention, high efficiency operation can be performed accurately.

BRIEF DESCRIPTION OF THE DRAWINGS The block diagram which shows the whole structure of the motor control apparatus which concerns on 1st Embodiment of this invention.
Fig. 2 is a diagram showing output voltage waveforms when the three-phase inverter circuit 17A is driven using three-phase modulation.
Fig. 3 is a diagram showing output voltage waveforms when the three-phase inverter circuit 17A is driven using up and down 60 degree fixed two-phase modulation.
4 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the upper fixed 120-degree two-phase modulation.
Fig. 5 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120 degree two-phase modulation.
Fig. 6 is a characteristic diagram showing the relationship between the voltage of the IGBT and the current to the MOSFET.
Fig. 7 is a characteristic diagram showing the relationship between the losses in the current of the IGBT and the MOSFET.
8 is a characteristic diagram showing the relationship between the junction temperature (channel temperature) and the circuit loss of a MOSFET in each modulation method in a three-phase inverter circuit in which an IGBT is disposed on an upper arm and a MOSFET is disposed on a lower arm.
Fig. 9 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is small.
Fig. 10 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is medium.
Fig. 11 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is large.
12 is a block diagram showing an overall configuration of a motor control device according to a second embodiment of the present invention.
FIG. 13 is a table showing a relationship of a modulation method with respect to a modulation rate in the three-phase inverter circuit 17A. FIG.
Fig. 14 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using three-phase modulation when the modulation rate is one.
Fig. 15 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120 degree two-phase modulation when the modulation rate is one.
Fig. 16 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using up and down 60-degree fixed two-phase modulation during overmodulation.
FIG. 17 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation during overmodulation.
Fig. 18 is a block diagram showing the structure of a converter circuit for controlling a direct current output voltage, which is a direct current power source used in the three-phase inverter circuit of the motor control apparatus according to the third embodiment of the present invention, and the reactor is provided on the alternating current side.
Fig. 19 is a block diagram showing the structure of a converter circuit which is a DC power supply used in a three-phase inverter circuit of the motor control device according to the third embodiment of the present invention, and includes a reactor on the DC side to control the DC output voltage.
20 is a block diagram showing a configuration of a converter circuit which is a DC power supply used in a three-phase inverter circuit of the motor control device according to the fourth embodiment of the present invention, and which is capable of controlling the electric wave double voltage.
21 is a block diagram showing an overall configuration of a motor control device according to a fifth embodiment of the present invention.
Fig. 22 is a characteristic diagram showing the relationship between device temperatures and on resistances of Si-MOSFETs, SJ-MOSFETs, and SiC-MOSFETs.
Fig. 23 is a block diagram showing the overall configuration of a motor control device according to a tenth embodiment of the present invention.
24 is a circuit configuration diagram in which a SiC-SBD is connected in parallel with a MOSFET in a motor control device according to an eleventh embodiment of the present invention.

EMBODIMENT OF THE INVENTION Hereinafter, the air conditioner equipped with the some embodiment of the motor control apparatus which concerns on this invention, and the motor control apparatus which concerns on this invention is demonstrated in detail, referring drawings.

In addition, in drawing for demonstrating some embodiment, in common, the common component is attached | subjected the common number, and the overlapping description of the content is abbreviate | omitted.

&Quot; First embodiment "

<Overall configuration of the motor control device according to the first embodiment of the present invention>

First, the whole structure of the motor control apparatus which concerns on 1st Embodiment of this invention is demonstrated with reference to FIG. 1 is a block diagram showing the overall configuration of a motor control device 11A according to a first embodiment of the present invention.

The motor control device 11A according to the first embodiment corresponds to the three-phase synchronous motor 15 (the "three-phase motor" of the present invention) using the three-phase inverter circuit 17A. For example, the case where drive control of a DC brushless motor etc. is used is demonstrated as an example.

The motor control device 11A according to the first embodiment includes the junction temperature Tj for the first to sixth switching elements SIup, SMun, SIvp, SMvn, SIwp, SMwn, and the three-phase inverter circuit 17A. By switching the modulation scheme used for driving the three-phase inverter circuit 17A based on the flowing circuit current Io, accurate performance of high efficiency operation is achieved.

In detail, the motor control device 11A according to the first embodiment includes three DC motors 13 and drive control of the three-phase synchronous motor 15 by PWM driving as shown in FIG. 1. The phase inverter circuit 17A, the electromagnetic induction type wire current sensor 19, the junction part temperature detection part 21, the modulation system control part 23, and the inverter drive circuit 25 are comprised.

The DC power supply 13 is a storage battery, for example. However, as the DC power supply 13 described later in detail, for example, a converter circuit 100A (see FIG. 18) or the like may be employed.

The three-phase inverter circuit 17A converts the DC power supplied from the DC power supply 13 into three-phase AC power that is a pseudo sine wave in u phase, v phase, and w phase based on the pulse width modulated wave signal (PWM signal). Then, by supplying the three-phase AC power which is the pseudo sine wave after conversion to the three-phase synchronous motor 15, it has a function to drive control of the three-phase synchronous motor 15. As shown in FIG.

The three-phase inverter circuit 17A has first to sixth switching elements SIup, SMun, SIvp, SMvn, SIwp, SMwn, as shown in FIG.

In the following description, when the first to sixth switching elements SIup, SMun, SIvp, SMvn, SIwp, SMwn are collectively referred to as "first to sixth switching elements SIup to SMwn". do.

As the first, third and fifth switching elements (switch elements of the upper arm) (SIup, SIvp, SIwp), a semiconductor element having an IGBT structure is used. On the other hand, as the second, fourth, and sixth switching elements (switch elements of the lower arm) SMun, SMvn, SMwn, semiconductor elements having a M0SFET structure are used.

The first and second switching elements SIup and SMun are connected in series via the first connection point Nd1. The reflux diode Dup and the parasitic diode Dun are antiparallel connected to each of the first and second switching elements SIup and SMun. The first connection point Nd1 is connected to the u-phase power line of the three-phase synchronous motor 15. In the following description, the first switching element SIup may be referred to as the first upper arm UA1, and the second switching element SMun may be referred to as the first lower arm LA1.

The third and fourth switching elements SIvp and SMvn are connected in series via the second connection point Nd2. In each of the third and fourth switching elements SIvp and SMvn, the reflux diode Dvp and the parasitic diode Dvn are anti-parallel connected. The second connection point Nd2 is connected to the v-phase power line of the three-phase synchronous motor 15. In the following description, the third switching element SIvp may be referred to as the second upper arm UA2, and the fourth switching element SMvn may be referred to as the second lower arm LA2.

The fifth and sixth switching elements SIwp and SMwn are connected in series via the third connection point Nd3. A reflux diode Dwp and a parasitic diode Dwn are antiparallel connected to each of the fifth and sixth switching elements SIwp and SMwn. The third connection point Nd3 is connected to the w-phase power line of the three-phase synchronous motor 15. In the following description, the fifth switching element SIwp may be referred to as the third upper arm UA3, and the sixth switching element SMwn may be referred to as the third lower arm LA3.

Series connection circuits of the first and second switching elements SIup and SMun, series connection circuits of the third and fourth switching elements SIvp and SMvn, and series connection circuits of the fifth and sixth switching elements SIwp and SMwn. Are connected in parallel to each other between the positive DC bus PL and the negative DC bus NL.

The wire current sensor 19 is provided in close proximity to the negative DC bus NL as shown in FIG. 1. The negative DC bus NL is grounded. The wire current sensor 19 has a function of detecting a circuit current Io flowing from the DC power supply 13 to the three-phase inverter circuit 17A. The wire current sensor 19 corresponds to the "current detector" of the present invention. The circuit current Io detected by the wire current sensor 19 is sent to the modulation method determination unit 37 of the junction temperature estimation unit 31 and the modulation method control unit 23 described later, respectively.

The junction temperature detector 21 has a function of detecting junction temperature Tj of the first to sixth switching elements SIup to SMwn. The junction temperature detector 21 corresponds to the "temperature detector" of the present invention.

The junction temperature of the first to sixth switching elements SIup to SMwn may be collectively referred to as "junction temperature for the switching element". The junction temperature detector 21 is composed of a temperature measurement unit 30 and a junction temperature estimation unit 31.

The temperature measuring unit 30 has a function of measuring the temperature of a substrate (not shown) on which the first to sixth switching elements SIup, SMun, SIvp, SMvn, SIwp, and SMwn are mounted. The temperature measurement part 30 is comprised by connecting the pullup resistor 33 and the thermistor 35 in series between DC power supply Vdd and a ground terminal. The pullup resistor 33 and thermistor 35 are mounted directly on the substrate. This is, for example, compared with the case where the pull-up resistor 33 and thermistor 35 are indirectly provided to the substrate in a heat sink or the like, so that the detection accuracy of the substrate temperature is increased, and further, the junction temperature Tj of the switching element with respect to the switching element is increased. This is because the estimation precision increases. The potential (substrate temperature information) of the connection point P1 of the pull-up resistor 33 and the thermistor 35 is sent to the junction temperature estimation unit 31. In addition, the thermistor 35 is supposed to be mounted on the same substrate as the MOSFET in order to detect the temperature of the substrate on which the MOSFET is mounted.

The junction temperature estimation unit 31 has a function of estimating the junction temperature Tj for the switching element. Specifically, the junction temperature estimation unit 31 includes the substrate temperature information (potential of the connection point P1) measured by the temperature measurement unit 30, the substrate obtained in advance, and the first to sixth switching elements SIup. The junction temperature Tj of the first to sixth switching elements SIup to SMwn is estimated using the information on the thermal resistance between each of the to SMwn.

The modulation method control unit 23 is composed of a modulation method determination unit 37 and a modulation method commanding unit 39. The modulation method determination unit 37 stores the modulation method determination information described later. The modulation method determination unit 37 flows to the modulation method information, the junction temperature Tj for the switching element estimated by the junction temperature estimation unit 31 and the three-phase inverter circuit 17A detected by the wire current sensor 19. Based on the circuit current Io, it has a function of determining the modulation method used when PWM driving the three-phase inverter circuit 17A. The modulation method command unit 39 has a function of sending command information to the inverter drive circuit 25 so as to PWM drive the three-phase inverter circuit 17A using the modulation method determined by the modulation method determination unit 37.

In addition, the modulation method control unit 23 and the junction temperature estimating unit 31 include, for example, a microcomputer (not shown) provided with a CPU (Central Processing Unit), a ROM (Read 0nly Memory), a RAM (Random Access Memory) and the like. (Hereinafter referred to as "microcomputer"). The microcomputer reads and executes a program stored in the ROM, and performs execution control of various functional units including a junction temperature estimation unit 31, a modulation method determination unit 37, and a modulation method command unit 39. Function.

The inverter drive circuit 25 controls switching of the first to sixth switching elements SIup to SMwn according to the command information about the modulation method sent from the modulation method command unit 39 of the modulation method control unit 23 (PWM control). Is configured to have a function of driving the three-phase inverter circuit 17A by using a predetermined modulation method.

<2 phase modulation and 3 phase modulation>

Here, to facilitate understanding of the modulation scheme used for driving the three-phase inverter circuit 17A according to the first embodiment of the present invention, reference is made to FIGS. 2 and 3 for two-phase modulation and three-phase modulation. Will be explained.

FIG. 2 is a diagram showing output voltage waveforms when the three-phase inverter circuit 17A is driven using three-phase modulation. FIG. 3 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using up and down 60 degree fixed two-phase modulation. In FIG.2 and FIG.3, the horizontal axis shows the phase [Fig.], And the vertical axis shows the duty [%] of a PWM control waveform.

4 to 5, 9 to 11, and 14 to 17, similarly to Figs. 2 and 3, the horizontal axis shows a phase [Fig.] And the vertical axis shows the duty [%] of the PWM control waveform. It is shown.

When the three-phase synchronous motor 15 is driven by PWM control using the three-phase inverter circuit 17A as shown in FIG. 1, three-phase modulation is generally used. In three-phase modulation, as shown in FIG. 2, PWM control is performed over the phase period of 180 degrees of electrical angle with respect to the voltage of u phase, v phase, and w phase.

On the other hand, in the two-phase modulation, the motor current of the three-phase synchronous motor 15 is determined not by the phase voltage but by the phase voltage, so that each phase voltage is fixed for each predetermined period while ensuring the phase voltage. By turning on each switching element of), as shown in FIG. 3, PWM control is performed so that one phase may be sequentially fixed by the electric angle (pi) / 3 (60 degree | times) by the high power supply level or the low power supply level for every phase. The two-phase modulation as shown in Fig. 3 is specifically called up and down 60 degrees fixed two-phase modulation.

In short, the 60 degree up and down fixed two-phase modulation constitutes the first to third upper arms UA1, UA2 and UA3 and the first to third lower arms LA1, LA2 and LA3 of the three-phase inverter circuit 17A. Two-phase modulation is performed by sequentially fixing the first to sixth switching elements SIup to SMwn at 60 degrees at a high power supply level (100% duty control) or a low power supply level (0% duty control).

When the three-phase inverter circuit 17A is driven using three-phase modulation, as shown in Fig. 2, the duty of each of the three phase voltages (Vu, Vv, Vw) with respect to the change of phase is changed into a substantially sinusoidal shape. . For this reason, the carrier frequency of a PWM control waveform becomes high, the switching loss of the 1st-6th switching elements SIup-SMwn increases, and the operation efficiency of the three-phase inverter circuit 17A is reduced.

On the other hand, when the three-phase inverter circuit 17A is driven using the fixed two-phase modulation up and down 60 degrees, as shown in FIG. 3, in the u-phase voltage, the phase section of 60 to 120 degrees and the w phase voltage are 180 100-degree duty control (high power supply level) is performed in each of the phase section of FIGS. In addition, in the u-phase voltage, 0% duty control (low power supply level) in the phase section of 240 degrees to 300 degrees, the w phase voltage of 0 degrees to 60 degrees, and the v phase voltage of 120 degrees to 180 degrees, respectively. ) Is performed.

When the three-phase inverter circuit 17A is driven using fixed two-phase modulation up to 60 degrees, switching is performed in a phase section corresponding to one third of each of the voltages Vu, Vv, and Vw of each of the three phases. Not performed (100% duty control or 0% duty control). For this reason, the switching loss of the 1st-6th switching elements SIup-SMwn does not generate | occur | produce in this phase section corresponded to this 1/3. As a result, the operating efficiency of the three-phase inverter circuit 17A can be improved.

In addition, for each phase, the high power supply level (100% duty control) or the low power supply level (0% duty control) is sequentially fixed by 2π / 3 (120 degrees) of electric angle, thereby reducing the switching loss of the three-phase inverter circuit. Also, a technique for applying a three-phase voltage to a three-phase motor by stopping the two-phase modulation method when the amplitude of the phase voltage is small is known (for example, Japanese Patent Application Laid-Open No. 2006-217673 or Japanese Patent Application). See publication 2005-229676). Such two-phase modulation is called fixed 120 degree two-phase modulation. Among these, the thing which fixed the fixed phase of 2-phase modulation at the high power supply level (100% duty control) of DC voltage is called upper fixed 120 degree two-phase modulation, and the fixed phase of 2-phase modulation is called the low power supply level of DC voltage ( 0% duty control) is called lower fixed 120 degree two-phase modulation.

In other words, the upper fixed 120 degree two-phase modulation constitutes the first to third upper arms UA1, UA2, UA3 and the first to third lower arms LA1, LA2, LA3 of the three-phase inverter circuit 17A. Two-phase modulation is performed by sequentially fixing the first to sixth switching elements SIup to SMwn by 120 degrees. In addition, the lower fixed 120 degree two-phase modulation constitutes the first to third upper arms UA1, UA2, UA3 and the first to third lower arms LA1, LA2, LA3 of the three-phase inverter circuit 17A. Two-phase modulation is performed by sequentially fixing the first to sixth switching elements SIup to SMwn by 120 degrees.

4 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the upper fixed 120-degree two-phase modulation. FIG. 5 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation.

When the three-phase inverter circuit 17A is driven using the upper fixed 120-degree two-phase modulation, one-third of each of the three-phase voltages Vu, Vv, and Vw, as shown in FIG. No switching is performed in the phase section corresponding to 100% duty control. Therefore, when PWM control is performed using the upper fixed 120 degree two-phase modulation shown in Fig. 4, the PWM control is performed with a duty in the range of about 15% to about 85% as in the three-phase modulation shown in Fig. 2. In comparison, the switching losses of the first to sixth switching elements SIup to SMwn can be reduced. As a result, the operating efficiency of the three-phase inverter circuit 17A can be improved.

When the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation, as shown in Fig. 5, three minutes for each of the voltages Vu, Vv, and Vw of the three phases are shown. No switching is performed in the phase section corresponding to 1 (0% duty control). Therefore, when the PWM control is performed using the lower fixed 120 degree two-phase modulation, the PWM control is performed with a duty in the range of about 15% to about 85% as in the three-phase modulation shown in FIG. Switching losses of the first to sixth switching elements SIup to SMwn can be reduced. As a result, the operating efficiency of the three-phase inverter circuit 17A can be improved.

<Characteristics of IGBT and MOSFET>

Next, the characteristics of the IGBT and the MOSFET used as the first to sixth switching elements SIup to SMwn will be described. 6 is a characteristic diagram showing the relationship between the voltage of the IGBT and the current to the MOSFET. Fig. 7 is a characteristic diagram showing the relationship between the losses of the IGBT and the MOSFET with respect to the current. In FIG. 6, the horizontal axis shows current, and the vertical axis shows voltage. In Fig. 7, the abscissa shows the current and the ordinate shows the loss.

As shown in FIG. 6, the characteristics of the collector emitter voltage with respect to the collector current of the IGBT rapidly increase in the rising section of the collector current, and then exhibit a moderately linear increase characteristic. On the other hand, the characteristics of the drain-source voltage with respect to the drain current of the MOSFET show a moderately linear increase characteristic in all current sections. The characteristics of the voltage between the collector emitters in relation to the IGBT and the characteristics of the drain and source voltages in relation to the MOSFETs intersect at the critical point as shown in FIG. 6. In other words, in the low load region (low input region) compared to the threshold point, the characteristics of the collector emitter voltage regarding the IGBT exceed the characteristics of the drain source voltage regarding the MOSFET, but in the high load region (high input region) compared to the threshold point. The relationship between the two is reversed.

Due to the characteristic relationship as shown in FIG. 6, as shown in FIG. 7, in the low load region (low input region) as compared with the critical point, the loss characteristic regarding the IGBT exceeds the loss characteristic regarding the MOSFET. In the high load region (high input region) as compared with the critical point, the relationship between the above is reversed. That is, the loss of the MOSFET is smaller than that of the IGBT in the low load region, but larger than the IGBT in the high load region. This is because the loss of the MOSFET increases by the power of the current.

That is, as shown in Fig. 7, the MOSFET has a lower loss than the IGBT in the low load region (low input region), but the loss increases in comparison with the IGBT in the high load region (high input region). This is because the on-resistance of the MOSFET has a positive temperature characteristic, and therefore the on-resistance becomes larger at high loads (at high inputs), and the loss is increased by the power of the current. Therefore, in order to achieve high efficiency operation of the three-phase inverter circuit 17A, the current flow rate to the MOSFET side is increased at low load (low input) and the current flow rate to the IGBT side at high load (high input). It is preferable to switch the modulation scheme in consideration of changing the amount of current flow to each switching element in accordance with the magnitude of the loss of each switching element in each load region.

In view of this, the above-mentioned Patent Document 1 proposes an inverter device having an inverter circuit combining an IGBT and a MOSFET. In this technique, an inverter circuit having an IGBT in the upper arm and a MOSFET in the lower arm of the inverter circuit employs a configuration of switching between three-phase modulation and fixed two-phase modulation up to 60 degrees in accordance with the magnitude of the load. . For the determination of the magnitude of the load, the motor current, the input voltage to the inverter circuit, the magnitude of the on / off duty with respect to the switching element, or the rotational speed of the motor is used.

However, since the on-resistance of the MOSFET has a positive temperature characteristic, the value of the on-resistance changes depending on the temperature of the MOSFET. For this reason, even when the same current is energized using three-phase modulation or two-phase modulation, the conduction loss generated in the MOSFET changes depending on the height of the element temperature. Moreover, in MOSFET, the ratio of temperature rise at the time of high load is large compared with IGBT. Therefore, only by switching three-phase modulation and fixed two-phase modulation up and down 60 degrees in accordance with the magnitude of the load, high-temperature operation cannot be maintained when a temperature change occurs in the MOSFET.

FIG. 8 is a characteristic diagram showing a relationship between a junction temperature (channel temperature) of a MOSFET and a circuit loss when the three-phase inverter circuit 17A is driven using a plurality of modulation schemes. In FIG. 8, channel temperature [degrees Celsius] is shown on the horizontal axis, and circuit loss [W] is shown on the vertical axis.

Moreover, as a characteristic parameter (modulation system) of a circuit loss, three phase modulation, 60 degrees up and down fixed two-phase modulation, upper fixed 120 degree two-phase modulation, and lower fixed 120 degree two-phase modulation are shown.

As shown in FIG. 8, even when the three-phase inverter circuit 17A is driven by using any two-phase modulation method, compared with the case where the three-phase inverter circuit 17A is driven by using three-phase modulation. , The circuit loss is small. In addition, at the low temperature of the MOSFET (about 40 ° C. or less), the case where the three-phase inverter circuit 17A is driven using the lower fixed 120 degree two-phase modulation becomes the lowest loss. On the other hand, when the MOSFET is at a high temperature (greater than about 40 ° C), the case where the three-phase inverter circuit 17A is driven using the upper fixed 120-degree two-phase modulation is the lowest loss.

This is because the on-resistance of the MOSFET at high temperature is larger than that at low temperature, and therefore the conduction loss of the MOSFET increases with the increase in the on-resistance. For example, when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation when the junction temperature (the concept encompassing the channel temperature. As shown in the figure, the circuit loss is larger than in the case where the three-phase inverter circuit 17A is driven using the upper fixed 120-degree two-phase modulation. Therefore, when the junction temperature of a MOSFET is high, even if it is low load (when low input), even if it drives the three-phase inverter circuit 17A using lower fixed 120 degree two-phase modulation, high efficiency operation can be achieved. none.

By the way, in the technique of Patent Document 1, the modulation method is determined only by the magnitude of the load (load current). Therefore, even when the three-phase inverter circuit 17A is driven by using a lower fixed 120 degree two-phase modulation in which the driving efficiency becomes better at low load, when the junction temperature of the MOSFET is high, the MOSFET is turned on. Since the resistance becomes large and the loss of the MOSFET becomes large, the high efficiency operation of the three-phase inverter circuit 17A cannot be performed.

Therefore, in the motor control apparatus 11A according to the first embodiment, the junction temperature Tj for the first to sixth switching elements SIup, SMun, SIvp, SMvn, SIwp, SMwn, and the three-phase inverter circuit ( The modulation method used for driving the three-phase inverter circuit 17A is switched based on the circuit current Io (load magnitude) flowing through the 17A to achieve accurate high efficiency operation.

<Operation of the Motor Control Device 11A According to the First Embodiment>

Next, the operation of the motor control device 11A according to the first embodiment will be described. When the power switch (not shown) of the motor control device 11A according to the first embodiment is turned on, the three-phase inverter circuit 17A is connected to the command information about the modulation method transmitted from the inverter drive circuit 25. By sequentially turning on / off the first to sixth switching elements SIup to SMwn by the based PWM control signal (driving control signal), three-phase alternating current power of the pseudo sine wave by the PWM waveform is generated, and 3 The phase synchronizing motor 15 is driven.

During the driving of the three-phase synchronous motor 15, the junction temperature estimation unit 31 includes the circuit current Io flowing through the three-phase inverter circuit 17A detected by the wire current sensor 19, and the temperature measurement unit 30. The first to sixth switching elements SIup to SMwn by using the substrate temperature at which the thermistor 35 is measured and the thermal resistance between the first to sixth switching elements SIup to SMwn from the substrate. Assume the junction temperature Tj for

Here, the actual junction temperature estimation method by the junction temperature estimation part 31 is demonstrated. It is assumed that IGBTs or MOSFETs are employed as the first to sixth switching elements SIup to SMwn. First, the product of the current Io flowing through the three-phase inverter circuit 17A detected by the wire current sensor 19 and the saturation voltage Vce of the IGBT, or the product of the square of the current Io and the on-resistance of the MOSFET. From this, the conduction loss of the IGBT or MOSFET is calculated. Next, the junction temperature Tj is estimated by adding the product of the calculated conduction loss and the thermal resistance between the switching element and the substrate to the substrate temperature measured by the thermistor 35 of the temperature measurement portion 30.

However, since the on resistance Ron _{( Tj )} of the MOSFET becomes a function of the junction temperature Tj, the conduction loss becomes a function of the junction temperature Tj. Also, since the junction temperature Tj becomes a function of the conduction loss, it is not possible to constantly calculate the conduction loss and the junction temperature Tj. Therefore, as the on resistance Ron in calculating the conduction loss, when the substrate temperature measured by the thermistor 35 of the temperature measurement unit 30 is tc, the estimated temperature difference Δt with the actual junction temperature Tj is shown. It is assumed that Ron (tc + Δt), which is a value when (tc + Δt) is a temperature at which) is estimated. By doing in this way, conduction loss and junction temperature Tj can be computed uniformly.

The junction temperature estimation method of the MOSFET using the above procedure may be represented by Equations 1 and 2 below.

Here, Tj is the junction temperature, tc is the thermal resistance, Δt is the substrate temperature between the substrate temperature, P _{(tc + Δt)} is the conduction loss of the MOSFET, R _jθ is the first to sixth switching elements (SIup to SMwn) from the substrate and The estimated temperature difference of the junction temperature is shown.

The modulation method determination unit 37 of the modulation method control unit 23 performs modulation method determination information for determining the lowest loss modulation method related to the junction temperature Tj and the circuit current Io, and the junction temperature addition. Based on the junction temperature Tj estimated by the step 31 and the circuit current Io of the three-phase inverter circuit 17A detected by the wire current sensor 19, a modulation method that results in the lowest loss is determined, The modulation method is sent to the modulation method command unit 39 of the modulation method control unit 23. Upon receiving this, the modulation method command unit 39 sends command information about the modulation method to the inverter drive circuit 25. In response to this, the inverter drive circuit 25 generates a drive signal of PWM control according to the modulation method determined to be the lowest loss, and drives the first to sixth switching elements SIup to SMwn of the three-phase inverter circuit 17A. do.

The conduction loss at this time is not P _{( tc + Δt)} shown in Equation 1, but the three-phase inverter circuit detected by the on-resistance at the junction temperature estimated by the junction temperature estimation unit 31 and the wire current sensor 19 ( It calculates from the electric current Io which flows into 17A).

Specifically, for example, the value of the circuit current Io of the three-phase inverter circuit 17A detected by the live wire current sensor 19 is small (load is small), and the junction temperature estimation unit 31 estimates it. A case where the junction temperature Tj of the first to sixth switching elements SIup to SMwn is low is considered. In this case, the modulation method determination unit 37 of the modulation method control unit 23 compares each of the IGBTs of the switching elements SIup, SIvp, SIwp belonging to the upper arm, and the switching elements SMun, SMvn, Each MOSFET of SMwn) determines that the on-resistance and conduction loss are small, and uses the lower fixed 120 degree two-phase modulation as the lowest loss modulation method.

Thereby, the inverter drive circuit 25 performs drive control of the three-phase inverter circuit 17A using the lower fixed 120 degree two-phase modulation. As a result, the current flow through the switching elements SMun, SMvn, SMwn belonging to the lower arm of the three-phase inverter circuit 17A to each MOSFET can be increased, so that high efficiency operation of the three-phase inverter circuit 17A can be performed accurately. have.

In addition, for example, the first value of the circuit current Io of the three-phase inverter circuit 17A detected by the live wire current sensor 19 is small (load is small) and the junction temperature estimation unit 31 is estimated. Consider the case where the junction temperature Tj of the sixth to sixth switching elements SIup to SMwn is high. In this case, the modulation method determination unit 37 of the modulation method control unit 23 compares each of the IGBTs of the switching elements SIup, SIvp, SIwp belonging to the upper arm, and the switching elements SMun, SMvn, It is judged that each MOSFET of SMwn) has a large on-resistance and conduction loss, and determines which of the lowest loss loss modulation methods is one of up / down 60 degree fixed two-phase modulation or upper fixed 120 degree two-phase modulation. Down.

As a result, the inverter drive circuit 25 performs drive control of the three-phase inverter circuit 17A by using either up / down 60 degree fixed two-phase modulation or upper fixed 120 degree two-phase modulation. As a result, the efficiency of operation of the three-phase inverter circuit 17A can be precisely increased by increasing the current flow rate to each IGBT of the switching elements SIup, SIvp, and SIwp belonging to the upper arm of the three-phase inverter circuit 17A. Can be done.

For example, the 1st value which the circuit current Io of the three-phase inverter circuit 17A detected by the live wire current sensor 19 is large (load is large), and the junction temperature estimation part 31 estimated A case where the junction temperature Tj of the sixth to sixth switching elements SIup to SMwn is low is considered. In this case, the modulation method determination unit 37 of the modulation method control unit 23 compares each of the IGBTs of the switching elements SIup, SIvp, SIwp belonging to the upper arm, and the switching elements SMun, SMvn, Each MOSFET of SMwn) determines that the on-resistance and conduction loss are small, and uses the lower fixed 120 degree two-phase modulation as the lowest loss modulation method.

Thereby, the inverter drive circuit 25 performs drive control of the three-phase inverter circuit 17A using the lower fixed 120 degree two-phase modulation. As a result, the current flow through the switching elements SMun, SMvn, SMwn belonging to the lower arm of the three-phase inverter circuit 17A to each MOSFET can be increased, so that high efficiency operation of the three-phase inverter circuit 17A can be performed accurately. have.

For example, the 1st value which the circuit current Io of the three-phase inverter circuit 17A detected by the live wire current sensor 19 is large (load is large), and the junction temperature estimation part 31 estimated Consider the case where the junction temperature Tj of the sixth to sixth switching elements SIup to SMwn is also high. In this case, the modulation method determination unit 37 of the modulation method control unit 23 compares each of the IGBTs of the switching elements SIup, SIvp, SIwp belonging to the upper arm, and the switching elements SMun, SMvn, Each MOSFET of SMwn) determines that the on-resistance and conduction loss are large, and determines that the lowest loss is a modulation method using up and down 60 degree fixed two-phase modulation or upper fixed 120 degree two-phase modulation.

As a result, the inverter drive circuit 25 performs drive control of the three-phase inverter circuit 17A using up and down 60 degrees fixed two-phase modulation or upper fixed 120 degrees two-phase modulation. As a result, by increasing the current flow rate to each IGBT of the switching elements SIup, SIvp, SIwp belonging to the upper arm of the three-phase inverter circuit 17A, high efficiency operation of the three-phase inverter circuit 17A can be performed accurately. have.

<Effects of the Motor Control Device 11A According to the First Embodiment>

According to the motor control device 11A according to the first embodiment, the junction temperature of the first to sixth switching elements SIup to SMwn and the value of the circuit current Io and the modulation method of the three-phase inverter circuit 17A are determined. Based on the information, among the modulation schemes prepared in advance (three phase modulation, up and down 60 degree fixed two phase modulation, upper fixed 120 degree two phase modulation and lower fixed 120 degree two phase modulation), the lowest modulation method is determined, By switching and using the modulation method according to this determination result, high efficiency operation of the motor control apparatus 11A including the three-phase inverter circuit 17A can be performed correctly.

As a trigger point for switching the modulation method, for example, as shown in FIG. 8, when the junction temperature of the switching element exceeds the threshold temperature (about 40 ° C. in the example of FIG. 8), the modulation method is changed. Attention is drawn to the fact that the magnitude of the circuit loss changes, and instead of the lower fixed 120 degree two-phase modulation, the upper and lower 60 degree fixed two-phase modulation or the upper fixed 120 degree two-phase modulation is used to change the motor control device 11A. High efficiency operation of) can be performed accurately.

In addition, when the value of the circuit current Io of the three-phase inverter circuit 17A changes, the magnitude of the conduction loss of the switching element also changes. Therefore, you may change suitably the threshold of the temperature regarding a switching element for switching a modulation system according to the value of circuit current Io. In such a configuration, it is possible to switch and use an appropriate modulation method based on the conduction loss of the switching element according to the value of the circuit current Io.

&Quot; Second Embodiment &

<Switching Control of Modulation Method Based on Modulation Rate>

In the first embodiment, the junction temperature Tj and the circuit current Io relating to the switching element in the three-phase inverter circuit 17A are used as technical elements for switching the modulation method. In the second embodiment, The use of a modulation rate is different from the first embodiment.

The switching losses of the first to sixth switching elements SIup to SMwn of the three-phase inverter circuit 17A change not only in the modulation method but also in the modulation rate. Specifically, if the switching operation of the PWM control is performed when the circuit current Io of the three-phase inverter circuit 17A is large, the switching losses of the first to sixth switching elements SIup to SMwn become large. In addition, if the switching operation of the PWM control is performed when the circuit current Io is small, the switching loss of the first to sixth switching elements SIup to SMwn becomes larger than the above (when the circuit current Io is large).

In short, the switching loss of the first to sixth switching elements SIup to SMwn is changed by the magnitude (modulation value) of the circuit current Io when the switching operation of the three-phase inverter circuit 17A is performed. . In other words, the switching loss is greatly changed by the selection of the modulation method and the magnitude order of the modulation rate.

It demonstrates in more detail. For example, when comparing three-phase modulation and two-phase modulation, in three-phase modulation, the number of PWM switching times increases by three-thirds as in the case of two-phase modulation, and the current value including the peak value of the sinusoidal wave is Since the number of switching operations in a large place is large, the switching loss inevitably becomes large.

On the other hand, in two-phase modulation, if the fixed 120-degree two-phase modulation (upper fixed 120-degree two-phase modulation and lower fixed 120-degree two-phase modulation) and the up-down 60-degree fixed two-phase modulation are compared, both of the first The switching losses of the sixth to sixth switching elements SIup to SMwn are approximately the same. In addition, when the modulation rate is large, PWM switching is performed when the circuit current Io is large, so that the lower fixed 120-degree two-phase modulation has a larger switching loss than the up-and-down 60-degree fixed two-phase modulation. This reason will be described with reference to FIGS. 9 to 11.

FIG. 9 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is small. FIG. 10 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is medium. FIG. 11 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed phase 120 degrees two-phase modulation when the modulation rate is large. 9 to 11 show basic technical details for using an appropriate modulation method based on the modulation rate in the motor control apparatus 11B according to the second embodiment of the present invention.

As shown in Fig. 9, when the modulation rate is small, the duty of the PWM control is small and the peak of the output voltage waveform is small. However, as shown in Fig. 11, when the modulation rate is large, the duty of the PWM control is increased and the output voltage waveform is The peak is also increasing. That is, as shown in Figs. 9 to 11, it can be seen that as the modulation rate is increased, the duty of the PWM control is increased and the peak of the output voltage waveform is increased. For this reason, as the modulation rate is increased, the peak of the circuit current Io of the three-phase inverter circuit 17A is increased, so that the switching loss of the PWM control is increased.

For example, when the DC input voltage of the three-phase inverter circuit 17A fluctuates to increase the modulation rate, and the switching loss is increased to affect the operation efficiency of the three-phase inverter circuit 17A, the switching loss becomes small. By selectively switching and using the modulation method, high efficiency operation of the three-phase inverter circuit 17A can be performed accurately.

<Overall structure of motor control device 11B according to the second embodiment of the present invention>

Next, the motor control apparatus 11B which concerns on 2nd Embodiment of this invention is demonstrated with reference to FIG. 12 and FIG. 12 is a block diagram showing an overall configuration of a motor control device 11B according to a second embodiment of the present invention. FIG. 13 is a table showing the relationship of the modulation method to the modulation rate in the three-phase inverter circuit 17A.

11 A of motor control apparatus which concerns on 1st Embodiment, and 11 B of motor control apparatus which concerns on 2nd Embodiment have a common component. Therefore, by focusing attention on the difference between the two, the description of the motor control apparatus 11B according to the second embodiment will be replaced.

The difference between the motor control device 11A according to the first embodiment and the motor control device 11B according to the second embodiment is that the motor control unit 29 is provided in place of the junction temperature detection unit 21 and the voltage detection unit. (27) is added.

The motor control unit 29, in addition to the modulation method control unit 23 (modulation method determination unit 37 and modulation method command unit 39) described above, as shown in Fig. 12, the motor current reproducing unit 41, the motor voltage The calculation part 43 and the modulation rate calculation part 45 are comprised.

The motor current reproducing unit 41 of the motor control unit 29 flows to the three-phase synchronous motor 15 based on the circuit current Io of the three-phase inverter circuit 17A sent from the wire current sensor 19. It has a function of reproducing currents Iu, Iv, and Iw.

The motor voltage calculating section 43 uses the three-phase synchronous motor based on the three-phase alternating currents Iu, Iv, and Iw sent from the motor current reproducing section 41 and the motor command rotation speed f ^* sent from the outside. The three-phase AC command voltages (Vu ^* , Vv ^* , Vw ^* ) to be applied to 15) are calculated and transmitted to the inverter drive circuit 25. Furthermore, the motor voltage calculation unit 43 is a three-phase synchronous motor (15) calculates the amplitude value (Vs ^*) of the sine wave voltage, tuning the amplitude value (Vs ^*) of the sine wave voltage change operation unit (45) to be applied to the It has a function to transmit.

The modulation rate calculator 45 is a DC voltage Vd detected by the voltage detector 27 and an amplitude value Vs ^* of the sinusoidal voltage applied to the three-phase synchronous motor 15 calculated by the motor voltage calculator 43. Has a function of calculating the modulation rate kh and transmitting the modulation rate kh to the modulation method determining unit 37 of the modulation method control unit 23.

In addition, although the modulation method control part 23 provided with the modulation method determination part 37 and the modulation method command part 39 is comprised in the motor control part 29, their function is basically the same as that of 1st Embodiment. Do. However, in the modulation method determination unit 37 according to the first embodiment, the modulation method determination unit according to the second embodiment is compared with the determination of the modulation method based on the junction temperature Tj and the circuit current Io. In 37), the modulation method is determined based on the modulation rate kh sent from the modulation rate calculation unit 45.

The voltage detector 27 has a function of detecting a DC voltage of the DC power supply 13 and transmitting the detected DC voltage Vd to the modulation rate calculator 45 of the motor controller 29.

<Operation of the Motor Control Device 11B According to the Second Embodiment>

Next, operation | movement of the motor control apparatus 11B which concerns on 2nd Embodiment is demonstrated. In addition, description of the operation which overlaps with 1st Embodiment is abbreviate | omitted in principle.

First, the motor current reproducing unit 41 of the motor control unit 29 supplies the three-phase synchronous motor 15 to the three-phase synchronous motor 15 based on the circuit current Io flowing through the three-phase inverter circuit 17A detected by the wire current sensor 19. The three-phase alternating currents Iu, Iv, and Iw flowing are estimated and reproduced. Thereby, the motor voltage calculating part 43 of the motor control part 29 is based on the three-phase alternating currents Iu, Iv, Iw acquired from the motor current reproducing part 41, and the motor command rotational speed f ^* acquired from the exterior. The three-phase AC command voltages (Vu ^* , Vv ^* , Vw ^* ) to be applied to the three-phase synchronous motor 15 are calculated, and the inverter drives the three-phase AC command voltages (Vu ^* , Vv ^* , Vw ^* ). Transmit to circuit 25.

In addition, the method of calculating the three-phase alternating current command voltages Vu ^* , Vv ^* , Vw ^* to be applied to the three-phase synchronous motor 15 by, for example, Japanese Patent Application Laid-Open No. 2002 The general method (the method of dq conversion by vector control) disclosed also in -272194 can be used. In addition to this method, the three-phase AC command voltage may be calculated using three phases or two phases of the motor current.

In addition, changes in the motor voltage calculation unit 43 is a three-phase synchronous motor 15, the amplitude value (Vs ^*) for operation, and the amplitude value (Vs ^*) of the sine wave voltage of the sinusoidal voltage applied to the motor control unit 29 The transmission is sent to the tuning operation unit 45.

The modulation rate calculator 45 of the motor controller 29 includes a DC voltage Vd of the DC power supply 13 detected by the voltage detector 27 and a three-phase synchronous motor 15 calculated by the motor voltage calculator 43. The modulation rate kh is calculated on the basis of the amplitude value Vs ^* of the sinusoidal voltage applied to, and the modulation rate kh is transmitted to the modulation method determination unit 37 of the modulation method control unit 23.

The modulation method determination unit 37 stores a table indicating the relationship of the modulation method to the modulation rate as shown in FIG. The modulation method determination unit 37 determines a modulation method corresponding to the modulation rate kh based on the modulation rate kh calculated by the modulation rate calculation unit 45 and the table shown in FIG. The modulation method is sent to the modulation method command unit 39. Upon receiving this, the modulation method command unit 39 sends command information about the modulation method to the inverter drive circuit 25.

The inverter drive circuit 25 includes three-phase alternating current command voltages Vu ^* , Vv ^* , Vw ^* transmitted from the motor voltage calculator 43, and command information about the modulation method transmitted from the modulation method command unit 39. Based on the above, a drive signal of the PWM control is generated, and the drive signal of the PWM control is transmitted to the first to sixth switching elements SIup to SMwn of the three-phase inverter circuit 17A. As a result, the three-phase inverter circuit 17A performs PWM driving by an appropriate modulation method based on the modulation rate.

Next, a method of calculating the modulation rate performed by the modulation rate calculating unit 45 of the motor control unit 29 will be described. The general modulation rate is the ratio of the amplitude of the signal wave to the amplitude of the carrier wave. Therefore, the signal wave in PWM control of the three-phase inverter circuit 17A is a sine wave applied to the three-phase synchronous motor 15, and the carrier wave is a square in which half of the DC input voltage of the DC power supply 13 is amplitude. It becomes a wave. Therefore, when the amplitude of the sine wave applied to the three-phase synchronous motor 15 is Vs ^* and the direct current input voltage is Vd, the modulation rate kh is calculated by the following expression (3).

Here, the magnitude (threshold) of the modulation rate (kh) used when switching the modulation scheme is a three-phase including the switching losses of the first to sixth switching elements SIup to SMwn of the three-phase inverter circuit 17A. It depends on the magnitude of the losses in the inverter circuit as a whole. Therefore, the threshold value of the modulation rate kh is memorize | stored in the modulation system determination part 45 previously. Therefore, when the modulation rate kh calculated by the modulation rate calculation unit 45 as shown in Equation 3 exceeds a predetermined threshold stored in advance, the modulation method determination unit 45 performs modulation with small switching loss. You can switch between them.

Here, IGBTs and switching elements belonging to the lower arm (SMun, SMvn, SMwn) belonging to the switching elements SIup, SIvp, SIwp belonging to the upper arm of the three-phase inverter circuit 17A shown in FIG. A case where the MOSFET is arranged in the following will be described as an example.

As shown in Fig. 7, the MOSFET has a lower loss than the IGBT during low input (low load), so that high efficiency is achieved by performing a lower fixed 120 degree two-phase modulation. However, in the case of using a device having a large reverse recovery time of a parasitic diode such as a super junction MOSFET as the MOSFET, the lower fixed 120 degree two-phase modulation is performed, so that in the case of a three-phase inverter circuit using IGBTs for the upper arm and the lower arm. In comparison, the switching loss is further worsened.

For this reason, when a superjunction MOSFET is used for the lower arm, and the modulation rate is increased and the switching loss is deteriorated, the modulation method determination unit 37 is a modulation method corresponding to the modulation rate. The decision is made to use fixed two-phase modulation, or upper fixed 120 degree two-phase modulation. In this manner, by switching and using a modulation method with a small switching loss according to the size of the modulation rate, high efficiency operation of the motor control device 11B including the three-phase inverter circuit 17A can be accurately performed.

Here, as a specific trigger point for switching the modulation method, although the modulation rate is usually 1 or less, for example, the modulation rate is changed due to hunting phenomenon between the three-phase inverter circuit 17B and the three-phase synchronous motor 15 as a load. In the case of 1.15 or more, instead of the lower fixed 120 degree two-phase modulation, the upper and lower 60 degree fixed two-phase modulations or the upper fixed 120 degree two-phase modulation may be switched.

Fig. 14 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using three-phase modulation when the modulation rate is one. FIG. 15 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation when the modulation rate is one.

As shown in Fig. 14, in the case of three-phase modulation, when the modulation rate is 1, the peak of the output voltage waveform whose PWM duty is changed from 0 to 100% is maximized, and when the modulation rate is raised beyond that, the waveform is distorted. It becomes. On the other hand, as shown in Fig. 15, in the case of the lower fixed 120 degree two-phase modulation, even when the modulation rate is 1, the PWM duty is about 85% at maximum, and the peak of the output voltage waveform does not reach 100%. . This indicates that in the case of the lower fixed 120 degree two-phase modulation, the modulation rate has more margin than the three-phase modulation.

On the other hand, the region whose modulation rate is, for example, 1.1 or more, preferably 1.15 or more, is an overmodulation area. In other words, if the modulation rate is increased beyond that, the waveform of the line voltage applied to the three-phase synchronous motor 15 cannot be maintained in the sine wave and is distorted. For this reason, the line voltage actually applied to the three-phase synchronous motor 15 does not increase beyond that.

FIG. 16 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using up and down 60 degree fixed two-phase modulation during overmodulation. FIG. 17 is a diagram showing an output voltage waveform when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation during overmodulation.

As shown in Fig. 16, when the three-phase inverter circuit 17A is driven using up and down 60 degree fixed two-phase modulation during over modulation, the duty becomes 100% or 0% near the peak of the current value. No PWM switching is performed. On the other hand, as shown in Fig. 17, when the three-phase inverter circuit 17A is driven using the lower fixed 120-degree two-phase modulation during overmodulation, the duty is slightly lower than 100% near the peak of the current value. Since it is low, switching of PWM is performed with respect to the IGBT arrange | positioned at the upper arm. As a result, the switching loss is worsened.

Although not shown in the figure, PWM switching is performed near the peak even in the case of the upper fixed 120-degree two-phase modulation during overmodulation, but at this time, there is a large ratio of switching of the M0SFETs arranged on the lower arm. Therefore, the switching loss is small compared to the lower fixed 120 degree two-phase modulation with a large ratio of switching of the IGBTs disposed on the upper arm. In addition, even in the case of vertical 60 degree fixed two-phase modulation, the switching loss of the MOSFET disposed in the lower arm is smaller than in the case of the lower fixed 120 degree two-phase modulation. As a result, high efficiency operation of the three-phase inverter device 17A can be performed accurately.

As described above, when the modulation rate is overmodulated, for example, 1.15 or more, instead of the lower fixed 120 degree two-phase modulation, the upper and lower 60 degree fixed two-phase modulations or the upper fixed 120 degree two-phase modulation are switched and used. , The highly efficient operation of the motor control device 11B including the three-phase inverter device 17A can be performed accurately. In addition, it is preferable to actually allow a threshold of the modulation rate for switching the modulation method, for example, to switch the modulation method between the modulation rates of 1.10 to 1.15.

<Effects of the Motor Control Device 11B According to the Second Embodiment>

According to the motor control apparatus 11B which concerns on 2nd Embodiment, based on the modulation rate of the three-phase inverter circuit 17A, the modulation system (three-phase modulation, up-down 60 degree fixed two-phase modulation, upper fixed 120 degree) prepared in advance is carried out. Motor control device including three-phase inverter circuit 17A by determining the lowest loss modulation method among two-phase modulation and lower fixed 120-degree two-phase modulation), and switching the modulation method according to the determination result. High efficiency operation of 11B can be performed accurately. In addition, even when the modulation rate is over-modulated, the configuration in which the modulation scheme is switched to be used so as to reduce the switching loss of the switching element in the three-phase inverter circuit 17A is adopted. High efficiency operation of the motor control device 11B including a can be performed accurately.

&Quot; Third Embodiment &

As shown in FIG. 1 or FIG. 12, although the three-phase inverter circuit 17A was driven by the DC power supply 13 in 1st Embodiment and 2nd Embodiment, this DC power supply 13 is a DC output. It can be replaced by the converter circuit which can control voltage.

FIG. 18 is a block diagram showing a configuration of a converter circuit 100A that is a direct current power source of the three-phase inverter circuit according to the third embodiment of the present invention and includes a reactor on the alternating current side to control the direct current output voltage.

As shown in Fig. 18, the main circuit of the converter circuit 100A includes a reactor 103 connected in series with the commercial power source 101, and a diode bridge for rectifying the AC voltage of the commercial power source 101 to a DC voltage. 105, a smoothing capacitor 107 for smoothing the pulsation component included in the DC voltage rectified in the diode bridge 105, and a bidirectional switch 109 in which the diode bridge and the transistor are connected in anti-parallel. have.

In addition, the control system of the converter circuit 100A includes a zero cross detector 111 that detects a zero cross point (that is, a timing at which the AC voltage passes the zero potential) of the AC voltage of the commercial power supply 101, and the converter circuit 100A. And a direct current voltage detection unit 113 for detecting the direct current output voltage. In addition, the external controller 115 has a built-in converter control unit 117 for controlling the DC output voltage of the converter circuit 100A.

With this configuration, the converter control unit 117 selects a transistor belonging to the bidirectional switch 109 based on the zero cross voltage from the zero cross detector 111 and the DC output voltage from the DC voltage detector 113. To control. As a result, the current flowing through the reactor 103 connected in series with the commercial power supply 101 is controlled in synchronization with the zero cross point of the alternating voltage, so that the diode bridge 105 performs a synchronous rectification, so that the voltage is controlled. The voltage can be output. Therefore, the converter circuit 100 can supply the DC output voltage controlled by the DC voltage and smoothed to the smoothing capacitor 107 to the three-phase inverter circuit 11A shown in FIG. 1 or 12.

In other words, the converter controller 117 short-circuits the bidirectional switch 109 based on the DC output voltage fed back from the DC voltage detector 113 while synchronizing with the AC output voltage waveform detected by the zero cross detector 111. The AC current flowing through the reactor 103 on the commercial power supply 101 side can be controlled. Thereby, the converter circuit 100 can control the DC voltage output from the diode bridge 105, and can also improve power factor and suppress harmonics.

Fig. 19 is a block diagram showing the configuration of a converter circuit that is a direct current power source of the three-phase inverter circuit according to the third embodiment of the present invention and includes a reactor on the direct current side to control the direct current output voltage.

As shown in FIG. 19, the main circuit of the converter circuit 200 is connected in series with a diode bridge 203 for rectifying the AC voltage of the commercial power supply 201 to a DC voltage and a positive output terminal of the diode bridge 203. Is connected in a forward direction between the reactor 205 connected to the reactor 205, the transistor 207 connected in the forward direction between the output terminal of the reactor 205 and the negative output terminal of the diode bridge 203, and the output terminal of the reactor 205. And the smoothing capacitor 211 which smoothes the pulsation component contained in the DC voltage rectified by the diode bridge 203. As shown in FIG.

The control system of the converter circuit 200 includes a DC voltage detector 213 for detecting a DC output voltage of the converter circuit 200 and a DC current detector 215 for detecting a DC output current of the converter circuit 200. It is. In addition, the external controller 217 has a built-in converter control unit 219 for controlling the DC output voltage of the converter circuit 200.

According to the converter circuit 200 of the structure shown in FIG. 19, the converter control part 117 is based on the DC output voltage fed back from the DC voltage detector 113, and the DC output current fed back from the DC current detector 215. FIG. Since the transistor 207 is electrically controlled, the DC current flowing through the reactor 205 on the output side of the diode bridge 203 can be controlled. As a result, the converter circuit 200 can control the DC voltage output from the diode bridge 203. Therefore, the converter circuit 100 can supply the DC output voltage, whose voltage is controlled and smoothed to the smoothing capacitor 211, to the three-phase inverter circuit 17A shown in FIG. 1 or 12.

That is, for example, a method of controlling the DC output voltage of the converter circuit 100A by the reactor 103 on the AC side shown in FIG. 18 and the converter 205 on the DC side shown in FIG. 19. There is a method of controlling the DC output voltage of the circuit 200. Therefore, by using the converter circuit 100A or the converter circuit 200 capable of controlling the DC output voltage as shown in FIG. 18 or 19 for the input power supply of the three-phase inverter circuit 17A, the three-phase inverter circuit ( The DC input voltage of 17A) can be arbitrarily changed. As a result, the advantage of using an appropriate modulation method can be more exerted on the basis of the modulation rate described in the second embodiment.

&Quot; Fourth Embodiment &

As shown in FIG. 1 or FIG. 12, although the three-phase inverter circuit 17A was driven by the DC power supply 13 in 1st Embodiment and 2nd Embodiment, this DC power supply 13 is a radio wave distribution. It can be replaced by a converter circuit capable of voltage control.

20 is a block diagram showing the configuration of a converter circuit that is a direct current power source of the three-phase inverter circuit according to the fourth embodiment of the present invention and capable of controlling the full-wave voltage.

The converter circuit 100B capable of controlling the full-wave voltage distribution shown in FIG. 20 is the same as that of the converter circuit 100A shown in FIG. 18 with the same reference numerals as to components that control the DC output voltage. Therefore, the configuration and the operation will be described only for the components which perform the full-wavelength voltage control, avoiding overlapping explanations.

As shown in FIG. 20, in the converter circuit 100B capable of full-wave voltage control, the double-voltage capacitors 107A and 107B are connected in series instead of the smoothing capacitor 107 of the converter circuit 100A shown in FIG. 18. It is connected to the diode bridge 105 in parallel. In addition, the full-wave voltage switch 119 is connected from the connection point of the double voltage capacitor 107A and the double voltage capacitor 107B to the connection point of the two diodes which comprise one arm of the diode bridge 105. .

Moreover, as a control system, it is built in the external controller 115, and from the converter control part 117 which controls the DC output voltage of the converter circuit 100B to the full-wave voltage switch switch 119, the said full-wave voltage switch switch Control signal lines for turning on and off the 119 are connected.

With such a configuration, the converter circuit 100B controls the DC output voltage in the same manner as the above-described operation of the converter circuit 100A of FIG. 18, and also turns on and off the full-wave voltage switching switch 119. By doing so, the DC output voltage of the full-wave double voltage can be supplied to the three-phase inverter circuit 11A shown in FIG. 1 or FIG. 12, for example.

Here, the outline | summary of the control regarding the full wave double voltage which the converter circuit 100B performs is demonstrated. When the diode bridge 105 is charging the double voltage capacitors 107a and 107b in a positive half cycle of the alternating voltage, the full-wave double voltage switching switch 119 is turned off. Next, when the diode bridge 105 rectifies in a negative half cycle of the alternating voltage, the converter control unit 117 turns on the full-wave voltage double-over switch 119. As a result, only the double voltage capacitor 107b is charged. As a result, a full-wave double voltage is generated at both ends of the double voltage capacitors 107A and 107B connected in series (that is, the output terminals of the converter circuit 100B).

Therefore, the converter circuit 100B capable of controlling the DC output voltage as shown in FIG. 20 and capable of controlling the electric wave double voltage is input to the input power supply of the three-phase inverter circuit 17A of FIG. 1 or 12. By using it as such, the DC input voltage of the three-phase inverter circuit 17A can be greatly changed. As a result, the advantage of using an appropriate modulation method can be more exerted on the basis of the modulation rate described in the second embodiment.

<< fifth embodiment >>

It is a block diagram which shows the whole structure of the motor control apparatus which concerns on 5th Embodiment of this invention.

The motor control device 11C according to the fifth embodiment shown in FIG. 21 includes the motor control device 11A according to the first embodiment shown in FIG. 1 and the motor according to the second embodiment shown in FIG. 12. The control apparatus 11B is combined and the structure which changes and uses an appropriate modulation system is employ | adopted based on junction temperature Tj, circuit current Io, and modulation rate kh which concern on a switching element. Therefore, since the motor control apparatus 11C according to the fifth embodiment shown in FIG. 21 is composed of only the components described with reference to FIGS. 1 and 12, description of overlapping configurations is omitted.

In addition, although the circuit structure of the temperature measuring part 30 is abbreviate | omitted, it is the same as the circuit structure of the temperature measuring part 30 shown in FIG.

However, in the motor control apparatus 11C according to the fifth embodiment shown in FIG. 21, the modulation method determination unit 37 of the modulation method control unit 23 includes the junction temperature Tj of the switching element and the three-phase inverter. The modulation method is determined based on the circuit current Io of the circuit 17A and the modulation rate kh from the modulation rate calculating section 45.

According to the motor control device 11C according to the fifth embodiment, based on the junction temperature Tj of the switching element, the circuit current Io of the three-phase inverter circuit 17A, and the modulation rate kh, By switching and using a modulation method with a small switching loss, high efficiency operation of the motor control device 11C can be performed accurately.

In addition, the method of estimating junction temperature Tj and the calculation method of a modulation rate which concern on a switching element use the method similar to the above-mentioned 1st Embodiment and 2nd Embodiment.

Specifically, for example, when the input current is small and the device temperature is low, the lower fixed 120-degree two-phase modulation is used, and when the modulation rate is increased and the switching loss of the switching device is increased, the operating efficiency is deteriorated. 60 degree fixed two-phase modulation or upper fixed 120 degree two-phase modulation may be switched. Thus, based on the junction temperature Tj of the switching element, the circuit current Io of the three-phase inverter circuit 17A, and the modulation rate kh, the whole of the three-phase inverter circuit including the switching loss. By switching and using a modulation method with low loss, high efficiency operation of the motor control device 11C can be performed accurately.

<< sixth embodiment >>

In the sixth embodiment, the DC power supply 1 of the motor control device 11C according to the fifth embodiment shown in FIG. 21 is controlled by the DC output voltage of the third embodiment shown in FIGS. 18 and 19. It replaces with the possible converter circuit 100A, 200 or the converter circuit 100B which can control the full wave voltage distribution of 4th Embodiment shown in FIG. Thereby, the advantage of using the modulation method based on a modulation rate can be exhibited more favorable.

<< seventh embodiment >>

In the first to sixth embodiments, the IGBT is disposed in the switching elements SIup, SIvp, SIwp belonging to the upper arm of the three-phase inverter circuit 17A, and the switching elements SMun, SMvn, SMwn belonging to the lower arm. MOSFETs were placed. In the seventh embodiment, on the contrary, the MOSFET is disposed in the switching elements SIup, SIvp, and SIwp belonging to the upper arm, and the IGBT is disposed in the switching elements SMun, SMvn, SMwn belonging to the lower arm. Even in this case, a modulation scheme having a low loss of the entire circuit is selected based on a suitable combination of the junction temperature Tj of the switching element, the circuit current Io of the three-phase inverter circuit 17A, or the modulation rate kh. By selectively using, high efficiency operation of the motor control device can be performed accurately.

<< eighth embodiment >>

In the first to seventh embodiments, the MOSFET is used as the switching element of the upper arm or the lower arm of the three-phase inverter circuit 17A. In the eighth embodiment, the superjunction MOSFET [SJ having a lower on-resistance than the MOSFET is used. (Super Junction) -MOSFET]. As a result, a more efficient three-phase inverter circuit can be realized.

<< ninth embodiment >>

In the first to seventh embodiments, the MOSFET is used as the switching element of the upper arm or the lower arm of the three-phase inverter circuit 17A. In the ninth embodiment, the silicon carbide MOSFET [SiC ( Sillicon Carbide) -MOSFET]. As a result, a more efficient three-phase inverter circuit can be realized.

<< tenth embodiment >>

In the first to seventh embodiments, a three-phase inverter circuit in which an IGBT and a MOSFET, an IGBT and an SJ-MOSFET, or an IGBT and a SiC-MOSFET are combined has been described. In the tenth embodiment, a three-phase inverter circuit combining a SJ-MOSFET and a SiC-MOSFET is used. In this case, a more efficient three-phase inverter circuit can be realized. This will be described below with reference to FIGS. 22 and 23.

Fig. 22 is a characteristic diagram showing the relationship between device temperatures and on resistances of Si-MOSFETs, SJ-MOSFETs, and SiC-MOSFETs. In FIG. 22, element temperature is shown on the horizontal axis, and resistance on the vertical axis is shown. As shown in Fig. 22, since the Si-MOSFET and the SJ-MOSFET have positive temperature characteristics, the on-resistance increases as the device temperature increases. However, the SiC-MOSFET has a characteristic that the on-resistance hardly changes even when the device temperature rises. In addition, SiC-MOSFETs and SJ-MOSFETs have lower on-resistance characteristics than Si-MOSFETs. It is preferable to use a combination of SJ-MOSFET and SiC-MOSFET in the switching elements of the upper and lower arms of the three-phase inverter circuit by utilizing such low on-resistance characteristics.

Fig. 23 is a block diagram showing the overall configuration of a motor control device 11D according to the tenth embodiment of the present invention.

In the tenth embodiment, as shown in FIG. 23, the SiC-MOSFET is disposed in the switching elements SIup, SIvp, SIwp belonging to the upper arm of the three-phase inverter circuit 17B, and the switching element SMun belonging to the lower arm. , SMvn, SMwn). In FIG. 23, the same MOSFET symbol is used for both the upper arm and the lower arm. In addition, since it is the same structure as the motor control apparatus 11A which concerns on 1st Embodiment shown in FIG. 1 except the 3-phase inverter circuit 17B, the overlapping description is abbreviate | omitted.

Unlike the device characteristics shown in FIG. 22, when the on resistances of the SJ-MOSFET and the SiC-MOSFET are compared, the SJ-MOSFET may have a smaller on resistance. In such a case, high efficiency is achieved by increasing the current flow rate of the SJ-MOSFET in the lower arm. However, as the device temperature rises, the on-resistance of the SJ-MOSFET increases, and the magnitude of switching loss between the SJ-MOSFET and the SiC-MOSFET is reversed. By increasing the flow rate, high efficiency operation can be maintained.

Thus, in the three-phase inverter circuit 17B shown in FIG. 23, when the temperature of the switching element is low, the lower fixed 120 degree two-phase modulation is performed so as to increase the current flow rate to the lower arm. Moreover, when the temperature concerning a switching element is high, it switches to up-and-down 60 degree fixed two-phase modulation or upper fixed 120 degree two-phase modulation so that the current flow rate to an upper arm may be improved. Thereby, the motor control apparatus 11D which concerns on 10th Embodiment shown in FIG. 23 can maintain high efficiency operation.

Also, as a configuration of the three-phase inverter circuit 17B, the high efficiency operation of the motor control device 11D is also exhibited even when the SJ-MOSFET is disposed on the upper arm and the SiC-MOSFET is disposed on the lower arm. In this case, the upper fixed 120 degree two-phase modulation is used when the temperature of the switching element is low, and the upper fixed 60 degree two-phase modulation or the lower fixed 120 degree two-phase modulation when the temperature of the switching element is high. You can use to switch.

<< eleventh embodiment >>

In the first to tenth embodiments, in order to reduce switching losses, a method of switching and using a modulation method based on the junction temperature of the switching element, the circuit current of the three-phase inverter circuit, or the modulation rate has been described. . This is because the reverse recovery time of the flyback diode (parasitic diode) of the MOSFET used as the switching element is large, so that a large reverse recovery current is generated by switching of the IGBTs in which the MOSFET and the arm are paired. Therefore, in addition to switching the modulation method by circuit current, device temperature, or modulation rate, a SiC-Schottky Barrier Diode (SiC-SBD (Schottky Barrier Diode)), which is a device using silicon carbide as a reflux diode of a MOSFET, is used. This can further reduce switching losses.

This SiC-SBD is an element that improves the reverse recovery characteristics of a diode using Si (silicon), which is represented by a typical fast recovery diode (FRD), and is effective in reducing reverse recovery current. This will be described with reference to a diagram showing a circuit configuration in which SiC-SBDs are connected in anti-parallel to a MOSFET as shown in FIG. That is, as the reflux diode of the MOSFET, as shown in FIG. 24, the SiC-SBD 53 is connected between the drain sources of the MOSFET 51, and the reflux current is compared with the parasitic diode 55. By suppressing the reverse recovery current by flowing a large amount of), it is effective for further reducing the switching loss.

As the reflux diode, the SiC-SBD can be connected to either or both of the drain and the source of the MOSFET, or the collector and the emitter of the IGBT.

<< twelfth embodiment >>

Although the motor control device has been described in the first to tenth embodiments, as a twelfth embodiment, a high efficiency air conditioner can be realized by using the motor control device according to the first to tenth embodiments in an air conditioner. In other words, if the air conditioner configured to perform drive control of the three-phase synchronous motor is adopted by the motor control apparatus according to the first to tenth embodiments, an air conditioner having high efficiency and high energy saving performance can be provided.

Specifically, for example, when these motor control devices are mounted in an air conditioner and the motor control device is applied for the purpose of driving control of an outdoor fan motor of an air conditioner, an air conditioner having high efficiency and high energy saving performance is provided. It can be realized.

The air conditioner can greatly improve APF (Annual Performance Factor), which is an index indicating energy saving performance, by improving the operating efficiency in the low load region (middle / rated region) shown in FIGS. 6 and 7. In the motor control apparatus according to each embodiment of the present invention, the modulation method is optimally switched according to the circuit current of the three-phase inverter circuit, the temperature and the modulation rate related to the switching element. Therefore, the air conditioner with high energy saving performance can be provided through the motor control apparatus which concerns on each embodiment of this invention.

"theorem"

As mentioned above, although embodiment of the motor control apparatus and air conditioner which concerns on this invention was described concretely, this invention is not limited to the content of each embodiment mentioned above, A various change is made in the range which does not deviate from the summary. Of course it is possible.

11A: Motor Control Device According to First Embodiment
11B: Motor Control Device According to Second Embodiment
11C: Motor Control Device According to Third Embodiment
11D: Motor Control Device According to Tenth Embodiment
13: DC power
15: 3-phase synchronous motor (3-phase motor)
17A, 17B: three-phase inverter circuit
19: wire current sensor
21: junction temperature detector (temperature detector)
23: modulation method control unit
25: inverter drive circuit
27: voltage detector
29: motor control unit
30: temperature measurement part
31: junction temperature estimation unit
33: pullup resistor
35: thermistor
37: modulation method determination unit
39: modulation method command
41: motor current reproducing unit
43: motor current calculation unit
45: modulation rate calculator
51: MOSFET
53: SiC-SBD
55: parasitic diode
Dup: free-wheel diode
Dun: Parasitic Diodes
Dvp: free-wheel diode
Dvn: Parasitic Diode
Dwp: free-wheel diode
Dwn: Parasitic Diode
Io: Circuit current
PL: Definition DC bus
NL: negative DC bus
SIup, SMup (UA1): first switching element (switching element)
SIun, SMun (LA1): Second switching element (switching element)
SIvp, SMvp (UA2): third switching element (switching element)
SIvn, SMvn (LA2): fourth switching element (switching element)
SIwp, SMwp (UA3): Fifth switching element (switching element)
SIwn, SMwn (LA3): Sixth switching device (switching device)
100A, 100B, 200: Converter Circuit
101, 201: commercial power
103, 205: reactor
105, 203: Diode bridge
107, 211: smoothing capacitor
107A, 107B: Double Voltage Capacitor
109: bidirectional switch
111: zero cross detector
113: DC voltage detector
115, 217: controller
117, 219: converter control unit
119: radio wave double voltage switch
207: Transistor
209: non-return diode
213: DC voltage detector
215: DC current detector

Claims (19)

It is a motor control device which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power,
A temperature detector for detecting a temperature of the switching element;
A current detector for detecting a circuit current flowing in the three-phase inverter circuit;
And a modulation scheme control unit for performing control using a modulation scheme based on the temperature of the element detected by the temperature detector and the circuit current detected by the current detector;
Among the switching elements constituting the three-phase inverter circuit, the switching element constituting the upper arm is IGBT, the switching element constituting the lower arm is MOSFET,
The modulation method control unit, when the modulation rate is overmodulation of more than 1.1, instead of the lower fixed 120 degrees two-phase modulation for performing two-phase modulation by sequentially fixing the switching element constituting the lower arm by 120 degrees, Up and down 60 degree fixed two-phase modulation for performing two-phase modulation by sequentially fixing the switching elements constituting the upper arm and the lower arm by 60 degrees, or two phases by sequentially fixing the switching elements constituting the upper arm by 120 degrees. A motor control device characterized by performing control using an upper fixed 120 degree two-phase modulation for modulating. It is a motor control device which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power,
A temperature detector for detecting a temperature of the switching element;
A current detector for detecting a circuit current flowing in the three-phase inverter circuit;
And a modulation scheme control unit for performing control using a modulation scheme based on the temperature of the element detected by the temperature detector and the circuit current detected by the current detector;
Of the switching elements constituting the three-phase inverter circuit, the switching element constituting the upper arm is a MOSFET, the switching element constituting the lower arm is an IGBT,
The modulation method control unit, when the modulation rate is overmodulation of more than 1.1, instead of the upper fixed 120 degree two-phase modulation for performing two-phase modulation by sequentially fixing the switching element constituting the upper arm by 120 degrees, Up and down 60 degree fixed two-phase modulation for performing two-phase modulation by sequentially fixing the switching elements constituting the upper arm and the lower arm by 60 degrees, or two phases by sequentially fixing the switching elements constituting the lower arm by 120 degrees. A motor control device characterized by performing control using a lower fixed 120 degree two-phase modulation for modulating. It is a motor control device which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power,
A temperature detector for detecting a temperature of the switching element;
A current detector for detecting a circuit current flowing in the three-phase inverter circuit;
And a modulation scheme control unit for performing control using a modulation scheme based on the temperature of the element detected by the temperature detector and the circuit current detected by the current detector;
Among the switching elements constituting the three-phase inverter circuit, the switching element constituting the upper arm is IGBT, the switching element constituting the lower arm is MOSFET,
The modulation method control unit is a lower fixed 120 degree two-phase modulation for performing two-phase modulation by sequentially fixing the switching element constituting the lower arm by 120 degrees when the junction temperature of the switching element exceeds a predetermined temperature. Instead, up and down 60 degrees fixed two-phase modulation for performing two-phase modulation by sequentially fixing the switching elements constituting the upper arm and the lower arm by 60 degrees, or sequentially fixing the switching elements constituting the upper arm by 120 degrees. And control using upper fixed 120 degree two-phase modulation for two-phase modulation. It is a motor control device which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power,
A temperature detector for detecting a temperature of the switching element;
A current detector for detecting a circuit current flowing in the three-phase inverter circuit;
And a modulation scheme control unit for performing control using a modulation scheme based on the temperature of the element detected by the temperature detector and the circuit current detected by the current detector;
Of the switching elements constituting the three-phase inverter circuit, the switching element constituting the upper arm is a MOSFET, the switching element constituting the lower arm is an IGBT,
The modulation method control unit, when the junction temperature of the switching element exceeds a predetermined temperature, the upper fixed 120 degree two-phase modulation for performing two-phase modulation by sequentially fixing the switching element constituting the upper arm by 120 degrees Instead, up and down 60 degrees fixed two-phase modulation for performing two-phase modulation by sequentially fixing the switching elements constituting the upper arm and the lower arm by 60 degrees, or sequentially fixing the switching elements constituting the lower arm by 120 degrees. And control using the lower fixed 120 degree two-phase modulation for two-phase modulation. It is a motor control device which performs drive control of a three-phase motor using the three-phase inverter circuit which has a some switching element, and converts DC power into 3-phase AC power,
A temperature detector for detecting a temperature of the switching element;
A current detector for detecting a circuit current flowing in the three-phase inverter circuit;
And a modulation scheme control unit for performing control using a modulation scheme based on the temperature of the element detected by the temperature detector and the circuit current detected by the current detector;
Among the switching elements constituting the three-phase inverter circuit, the switching element constituting the upper arm is IGBT, the switching element constituting the lower arm is MOSFET,
In the region where the modulation rate is 0.2 or more and 0.6 or less, the lower fixed 120-degree two-phase modulation for two-phase modulation is performed by sequentially fixing the switching element constituting the lower arm by 120 degrees,
In the region where the modulation rate is 0.6 or more, the upper arm and the lower arm constitute the upper arm instead of the lower fixed 120 degree two-phase modulation in which the switching elements constituting the lower arm are sequentially fixed by 120 degrees to perform two-phase modulation. A motor control device, characterized in that up and down 60 degrees fixed two-phase modulation is performed by sequentially fixing the switching elements by 60 degrees to perform two-phase modulation. delete delete delete delete delete delete delete delete delete delete delete delete delete delete

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