JPWO2012153794A1 - Position sensorless control device for permanent magnet synchronous motor - Google Patents

Abstract

The fundamental wave current detector 8 detects a current in the zero voltage vector period V7 or V0 of PWM control. The current change detection unit 11 calculates the current difference of the actual current corresponding to the current differentiation from the detected current value. The current difference component calculation unit 12 calculates an electromotive force from the current detection value, the voltage drop due to the motor resistance R, the inductance L, and the angular frequency command ω *, and uses the time difference ΔTp between the inductance L and the current difference to generate the voltage component. The current difference component generated by The corrected angular frequency command calculation units 13 to 15 add the current difference and the current difference component of the actual current, multiply the filter by the correction gain, and multiply by the angular frequency weighting function Kω approximated to dividing the angular frequency command component. Thus, the correction angular frequency Δωp * is calculated.

Description

  The present invention relates to a position sensorless control device for a synchronous motor (hereinafter referred to as PM motor) using a permanent magnet as a field source, and in particular, a motor speed when starting control of a PM motor having no magnetic saliency by a current drawing method. Related to vibration suppression.

  The structure for arranging the field magnet of the PM motor includes a structure in which a magnet is attached to the surface of the field core (SPMSM: Surface Permanent Synchronous Motor) and a structure in which the magnet is embedded in the core (IPMSM: Interior Permanent Magnet). It can be divided roughly.

  In the permanent magnet synchronous machine, since the relative permeability of the permanent magnet is as low as about 1, the SPM structure and the IPM structure have different characteristics. In general, the SPM structure often has non-saliency in which the d-axis inductance and the q-axis inductance are almost equal, and the IPM structure often has a reverse salient pole characteristic in which the q-axis inductance is smaller than the d-axis. .

  Here, the case where the inductance components of the d-axis and the q-axis of the stator winding in the orthogonal coordinate system with the field pole (N pole) as a reference are substantially equal is defined as “having magnetic non-saliency”.

Definition of basic symbols d axis: field axis q axis: axis orthogonal to the field axis and moved in the forward rotation direction by 90 ° in electrical angle from the d axis Ld: d axis of the armature winding Inductance Lq: q-axis inductance of armature winding R: Armature winding resistance θ: Phase angle from U-phase winding axis to d-axis (electrical angle, normal rotation direction is positive)
ω: angular velocity (dθ / dt) (electrical angle, forward rotation direction is positive)
Here, the saliency is defined as follows.

“Synchronous machine with magnetic non-saliency” or “non-saliency machine”: Ld≈Lq “Synchronous machine with magnetic saliency” or “saliency machine”: Ld ≠ Lq Is a PM motor that is a non-salient pole machine and has no damper winding.

  As a method for controlling the speed and torque of the PM motor without detecting the magnetic pole position of the PM motor by using a position sensor, the position sensor is not required, such as a method of obtaining magnetic pole position information from the speed electromotive force of the PM motor. A position sensorless control method for controlling the rotation speed and the rotation phase has been proposed. Among these, the method of obtaining magnetic pole position information from the speed electromotive force of the PM motor is such that the speed electromotive force is relatively smaller than the rated voltage of the PM motor when the speed is near 10% or less of the rated speed of the PM motor. Therefore, the influence of the voltage error of the speed electromotive force becomes relatively large. Therefore, in the position sensorless control method using the speed electromotive force information, an accurate magnetic pole phase in the low speed range cannot be obtained. As a countermeasure for the magnetic pole phase accuracy by sensorless control in the low speed region, a magnetic pole phase estimation method in which a high frequency component is superimposed on a motor drive current has been developed for salient pole machines and the like (see, for example, Patent Document 1).

  However, a non-salient pole machine having non-saliency magnetically has a problem that a phase estimation method cannot be applied because a phase difference does not occur between harmonic voltage and current vectors. Therefore, in a non-salient pole machine, a current of an arbitrary amplitude is passed as in the drive control of a stepping motor, and the magnetic phase is forcibly maintained while maintaining the synchronous pull-in state by gradually changing the frequency of this current. Is started (followed by the current phase) (referred to as current drawing method).

  However, the starting method by the current drawing method has the following problems.

  (1) Since the torque fluctuates as a sin function of the phase difference between the magnetic pole axis and the current axis (magnetomotive force phase difference angle ψ), this torque characteristic is similar to that in which an elastic axis exists between the stator and the rotor. Behave. Then, speed vibration (shaft torsional vibration) that resonates with the rotor and load inertia moment occurs due to load torque fluctuation, etc. However, a PM motor without a damper winding has no vibration suppression effect and shaft torsional vibration (hereinafter referred to as “torsional torsion vibration”) , Referred to as axial vibration). This shaft vibration lowers the quality of torque output to the load, and there is a problem that coupling causes metal fatigue due to shaft vibration.

  (2) In the current pull-in method, torque pulsation corresponding to shaft torsion is added to the torque of the output shaft when the above-mentioned shaft vibration occurs, so if the current amplitude is not set large, step-out will occur exceeding the maximum torque. Occur. In order to prevent this step-out, a current that is 1.5 to 2 times larger than that required for normal vector control must be supplied. As a result, a large capacity power converter is required.

  Because of these problems, the scope of application of the current pull-in method is limited to the square reduction load characteristics with a small torque at the time of starting or low speed of a fan / pump.

  If the current drawing method is applied to the salient pole machine, there is a method of applying the high frequency method to vibration suppression as disclosed in Patent Document 2. Further, as a method for estimating the magnetic pole phase without using a high frequency component, there is a method of Non-Patent Document 1.

  The feature of the method of Non-Patent Document 1 is that in the switching mode by PWM control, the three-phase switching element has a period (V7) in which only the upper arm is conductive (ON) for all phases and only the lower arm is conductive for all phases ( ON) Current differential information is obtained during the period (V0) during which the magnetic pole phase is estimated. This period of V7 and V0 is usually called a zero voltage vector period. Since the generation ratio of the zero voltage vector periods V7 and V0 is large at a low speed of 10% or less of the rated speed of the PM motor, the time interval for detecting a plurality of currents to obtain current differential information is also ¼ or more of the PWM carrier period. Current can be detected at two or more times.

JP 2008-295220 A JP 2006-353025 A JP 2000-236694 A

J. et al. -L. Chen, T.A. -H. Uu, and C.I. -L. Chen: "Implementation of a Novel High-Performance Sensor IPMSM Control System", ICIT 2010, pp. 361-366 (2010)

  The magnetic pole phase estimation method in the position sensorless control of the PM motor is roughly classified into the following three types.

(A) Magnetic pole phase estimation method using velocity electromotive force (b) Magnetic pole phase estimation method using magnetic saliency (c) Magnetic pole using the fact that the magnetic saturation characteristics of the iron core are different in the N-pole and S-pole directions Phase estimation method Among these phase estimation methods, Non-Patent Document 1, which uses current differential information in a zero voltage vector period, includes not only the component (b) but also the magnetic saliency. The component of velocity electromotive force is also used for phase estimation.

  The method using the magnetic saliency of (b) has a drawback that even if the d-axis phase can be estimated, the N pole and the S pole cannot be distinguished. If the speed reaches a certain level, it becomes possible to determine the N pole and the S pole based on the electromotive force information in (a). Until then, the PM motor operates abnormally if there is an error in the estimation of the N pole and the S pole. Sometimes

  In the method (c) using the magnetic saturation of the iron core, it is necessary to make the design easy to perform magnetic saturation at the time of designing the motor, and it is necessary to pass an excessive current.

  As described above, the conventional PM motor position sensorless control method cannot accurately estimate the magnetic pole phase only by the magnetic pole phase estimation method using the speed electromotive force of (a). Further, since the method (b) uses magnetic saliency, it cannot be applied as it is to a non-salient pole machine having no magnetic saliency. Further, when the current drawing method is applied, there remains a problem that causes shaft torsional vibration. Further, the method (c) has a problem that a large-capacity power converter is required to prevent step-out.

  It is an object of the present invention to accurately estimate the magnetic pole phase of a PM motor having no magnetic saliency up to a low speed such as 10% or less of the rated speed, and to perform velocity oscillation by a current drawing method based on the estimation of the magnetic pole phase. An object of the present invention is to provide a position sensorless control device for a permanent magnet synchronous motor that can suppress (shaft vibration) and prevent step-out.

  Since the current differential information of the zero voltage vector period used in the present invention is not affected by the dead time, the speed is reduced from a lower speed than the magnetic pole phase estimation method using a voltage command or the like input to the conventional PWM control. Focusing on the possibility that electromotive force information can be obtained, current differential information during zero voltage vector period is detected to obtain speed electromotive force information, and the magnetic pole phase is estimated by the magnetic pole phase estimation method using this speed electromotive force. Based on this estimated magnetic pole phase, velocity vibration (shaft vibration) is suppressed when the current drawing method is applied to a PM motor with no magnetic saliency. And

(1) Phase estimation means for starting the magnetic pole phase θ of a synchronous motor that uses a permanent magnet that is a non-salient pole machine and does not have a damper winding as a field source to follow the current phase and estimates the magnetic pole phase θ A position sensorless control device for a permanent magnet synchronous motor having:
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points during each of the zero voltage vector periods;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the time difference ΔTp is divided by the inductance L. A current difference component calculation unit (12) that converts a current difference component that is a product of the value and the speed electromotive force;
An output obtained by adding the output of the current difference detection unit and the output of the current difference component calculation unit, removing the DC component by a filter, multiplying the sum by a correction gain K _p , and further multiplying the correction gain K _p angular frequency weight multiplied by the function K _omega calculates the correction angle frequency [Delta] [omega _p ^*, the correction angular frequency command calculator for calculating the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this to the ( 13-15)
It is provided with.

(2) Phase estimation means for starting the magnetic pole phase θ of a synchronous motor that uses a permanent magnet that is a non-salient pole machine and has no damper winding as a field source to follow the current phase and estimate the magnetic pole phase θ A position sensorless control device for a permanent magnet synchronous motor having:
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points during the zero voltage vector period;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A correction angular frequency Δω _p ^* is calculated by multiplying an output obtained by removing the DC component from the current difference by a filter and multiplying the correction gain by the filter, and multiplying the output multiplied by the correction gain K _p by an angular frequency weighting function K _ω. the was added to the angular frequency command omega ^* calculates a new angular frequency command omega _p ^* corrected angular frequency command calculation section (13 to 15),
It is provided with.

(3) Phase estimation means for starting the magnetic pole phase θ of a synchronous motor that uses a permanent magnet that is a non-salient pole machine and does not have a damper winding as a field source to follow the current phase and estimates the magnetic pole phase θ A position sensorless control device for a permanent magnet synchronous motor having:
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points symmetrical with respect to the upper and lower vertices of the triangular wave carrier signal during the zero voltage vector period;
The average value of the current difference of the current detection values at time t _2n and t _{(2n + 1)} at two or more points symmetrical with respect to the vertices of the zero voltage vector period (V7, V0) of the upper arm and the lower arm. The speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the time difference ΔTp is calculated by the inductance L. An average value calculation unit (12A) of a current difference component that is converted into a current difference component that is a multiplication value of the divided value divided by the speed electromotive force;
The average value output of the current difference of the current detection unit and the output of the current difference component calculation unit are added (12C), the DC value is removed by a filter, and the correction gain _Kp is multiplied. K _p calculates the correction angle frequency [Delta] [omega _p ^* by multiplying the angular frequency weighting function K _omega output multiplication, it calculates the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this correction angle A frequency command calculation unit (13 to 15);
It is provided with.

(4) Phase estimating means for starting the magnetic pole phase θ of a synchronous motor that uses a permanent magnet that is a non-salient pole machine and does not have a damper winding as a field source to follow the current phase and estimate the magnetic pole phase θ A position sensorless control device for a permanent magnet synchronous motor having:
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points during the zero voltage vector period;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A moving average of current differences at times t _2n and t _{(2n + 1)} at two or more points in two types of detection periods of the upper arm zero voltage vector period (V7) and the lower arm zero voltage vector period (V0). Then, the speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L and the angular frequency command ω ^* , and the time difference ΔTp is calculated by the inductance L. An average value calculation unit (12B) of current difference components to be converted into a current difference component that is a multiplication value of the divided value divided by the speed electromotive force;
Wherein adding the output of the current difference and the current difference component calculation unit of the current detecting portion (12C), the added value by removing the DC component in the filter multiplied by the correction gain K _p, multiplied further the correction gain Kp Output angular frequency weight multiplied by the function K _omega calculates the correction angle frequency [Delta] [omega _p ^*, the correction angular frequency command calculator for calculating the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this to the ( 13-15)
It is provided with.

  As described above, according to the present invention, the speed electromotive force information is obtained using the current differential information in the zero voltage vector period, and the magnetic pole phase is estimated by the magnetic pole phase estimation method using the speed electromotive force. Vibration suppression when applying the current pull-in method to start the PM motor by following the current phase with the magnetic pole phase θ of a PM motor that has no magnetic saliency based on the magnetic pole phase. It is possible to accurately estimate the magnetic pole phase of a non-PM motor to a low speed, and to suppress the speed vibration (shaft vibration) and the step-out by the current drawing method based on the estimation of the magnetic pole phase.

The apparatus block diagram which shows Embodiment 1 of this invention. The apparatus block diagram which shows Embodiment 2 of this invention. The apparatus block diagram which shows Embodiment 3 of this invention. The apparatus block diagram which shows Embodiment 4 of this invention. The example of the short circuit current at the time of V0 voltage vector. The example of the short circuit current at the time of V7 voltage vector. Definition period of zero voltage vectors V0 and V7. The relationship diagram of zero voltage vectors V0 and V7 and current detection. The characteristic example of the frequency weight function in FIG. The simulation waveform figure by the conventional basic current drawing method. The simulation waveform diagram by embodiment of this invention.

(1) Basic description of the invention Before showing an embodiment of the present invention, the basic principle and relational expression of the present invention will be described first. First, the relationship between the voltage, current, and the like exhibited by the synchronous machine can be expressed as an equation (1) in an orthogonal biaxial coordinate system (dq coordinate) having the magnetic pole axis as the d axis.

here,
V _d , V _q : d-axis, q-axis voltage component i _d , i _q : d-axis, q-axis current component φ _d : stator winding interlinkage magnetic flux by field magnetic flux (assuming that field magnetic flux is only d-axis )
R: Winding resistance L _d , L _q : Inductance component on the field axis (d axis) and an axis advanced by 90 ° phase p: Differential operator (= d / dt)
θ: Phase angle (electrical angle) of field axis (d-axis) viewed from U-phase winding axis
ω: angular frequency (electrical angle) dθ / dt
Next, since the present invention is intended for non-salient pole machines, when the inductance is set to L _d = L _q = L, the matrix becomes an element that is symmetric as shown in equation (2).

In order to convert this dq coordinate component from a fixed coordinate to a rotational coordinate (γ-δ coordinate) based on a current vector rotating at a phase angle θ _γ and an angular velocity ω _γ , the current viewed from the d axis The phase angle to the vector generation axis (γ axis) is defined as magnetomotive force phase difference angle ψ = θ _γ −θ, and equations (3) to (5) may be applied.

Further, in the current drawing method, only the current command I _γ ^* is set, and the current control response is also high, so that an actual current exactly equal to I _γ ^* and I _δ ^* = 0 flows according to the angular velocity command ω ^* . Assume that

  Substituting these formulas (3) to (5) into formulas (1) and (2) and multiplying the inverse matrix from the left of both sides to eliminate the conversion matrix of the voltage term, the formula (6) When it is further summarized, the equation (7) is obtained.

  Furthermore, if limited to the zero voltage vector period of the PWM pattern, the voltage component on the left side is replaced with zero as shown in equation (8). Here, the voltage drop component of the semiconductor element of the main circuit such as the PWM inverter is ignored, and it is considered that the change amount of the phase angle between the d axis and the γ axis is small, and the angular velocity difference between the d axis and the γ axis ( Assuming that dψ / dt) is also very small, dψ / dt≈0 is substituted and ignored.

By substituting the relationship of equation (8) into equation (7) and extracting only the γ-axis component, an equation for obtaining φ _d · sin ψ is obtained.

  Since the generated torque is a product of the current and the orthogonal flux linkage component, the torque Torque is obtained by the product of these orthogonal components.

  Here, since the positive torque is a torque component that accelerates the rotor in the forward rotation direction, positive torque is generated when the current is in a direction advanced by 90 ° with respect to the magnetic flux. If this torque can be estimated, the stabilization principle adopted in the position sensorless control method of the V / F type PM motor such as Patent Document 2 and Patent Document 3 can be applied.

  In Patent Document 3, the q-axis current component having the same phase component as that of the voltage vector is used for the feedback signal. However, this is equivalent from the viewpoint of a component proportional to the torque, and substantially the same vibration suppressing effect can be obtained. . In Patent Document 2, the phase difference (load angle δ) between the axis of the input voltage vector and the axis of the voltage vector due to the field magnetic flux is fed back. In both configurations, the detected value related to the motor torque is used as the feedback amount, and the stabilization principle using each information is explained. Here, the principle explanation about the frequency compensation necessary for stabilization is explained. Is omitted.

In conclusion, the frequency in the current drawing method may be corrected as ω _p ^* corrected by the equation (11) proportional to the torque.

In order to express the current differential p · i _γ in the equation (11) as a discrete system, the difference between the detected current values i _γ2n and i _{γ (2n + 1)} at time t _2n and t _{(2n + 1)} and the time difference ΔTp = t Approximate by _{(2n + 1)} −t _2n, and move the coefficient applied to the current difference i _{γ (2n + 1)} −i _{γ (2n)} to another term. Further, if negative signs are also collected, it can be transformed into equation (12). The times t _2n and t _{(2n + l)} and the current detection values i _{γ (2n)} and i _{γ (2n + l)} will be described in detail in FIG. 8 and the description of the trigger signal generator 9 described later. To do.

Since the expression (12) has a term of the angular frequency command ω ^{* in} the denominator, the compensation amount becomes excessive when the frequency is near zero. In order to prevent this division of zero, the polarity of the frequency is taken into consideration and approximated to an angular frequency weight gain K _ω (ω ^* ) that suppresses the magnitude of the compensation amount to zero near the frequency (13) I will treat it as an expression.

  Further, in order to suppress the vibration of the phase and the rotational speed, it is only necessary to compensate for the transient component of the equation (13), so that a bandpass filter function BPF { } Is applied.

(2) Basic Configuration of Apparatus An example of a basic configuration for realizing the present invention is shown in FIG. 1, FIG. 2, FIG. 3, and FIG. The apparatus configuration shown in FIGS. 1 to 4 and the main functions and operations in these configurations will be schematically described below.

Figure 5 is an example of a short-circuit current when the voltage V0 vector, the switching state of the inverter in the zero voltage vector period _{V0, (+ I u, -I} v, -I w) current flow path in the polarity of the three-phase current An example of Thus, it can be seen that the motor terminal current is independent of the DC link voltage source (V _dc ) of the inverter and is equivalent to a state where the three-phase terminals are short-circuited.

FIG. 6 is an example of a short-circuit current at the time of the V7 voltage vector, and the current flow path in the switching state of the inverter in the zero voltage vector period V7 and the three-phase current of (+ I _u , −I _v , −I _w ) polarity. An example of Thus, it can be seen that the motor terminal current is independent of the DC link voltage source (V _dc ) of the inverter and is equivalent to a state where the three-phase terminals are short-circuited.

  FIG. 7 shows the definition periods of the zero voltage vectors V0 and V7, and shows the zero voltage vector periods V0 and V7 in FIGS. 5 and 6 using an example of PWM control pulse generation by the triangular wave carrier comparison method.

  FIG. 8 shows the relationship between zero voltage vector periods V0 and V7 and current detection, and shows the relationship between the PWM pattern of FIG. 7 and current detection timing. Thus, the method of applying the current differentiation of the zero voltage vector period is referred to herein as the Current-Slope method.

  FIG. 1 is an apparatus configuration diagram for suppressing vibration by a current drawing method to which a current-slope method is applied, and shows an overall configuration diagram of a control system using the current sample of FIG.

  FIG. 9 shows an example of the characteristics of the angular frequency weighting function in FIG. 1, which is the example used in FIG.

  FIG. 2 is a configuration diagram of a device that suppresses vibration by a current drawing method to which the current-slope method is applied. In the configuration shown in FIG. It is a structural example.

  FIG. 3 is a configuration diagram of an apparatus that suppresses vibration by a current drawing method using the Current-Slope method. Compared to the configuration of FIG. The structure at the time of correcting to the electric current synthesize | combined from the some detected electric current value for difference calculation is shown.

  FIG. 4 is a configuration diagram of an apparatus that suppresses vibration by a current drawing method to which the Current-Slope method is applied. In contrast to the configuration of FIG. 3, the current difference value is the upper limit vertex (V7) and lower limit vertex of the triangular wave carrier. Since there are two types of (V0), an example in which statistical disturbance suppression is applied by obtaining a moving average of two types of average values is shown.

  FIG. 10 is a time chart example of start-up and reverse rotation operation in the low speed region by the current drawing method when vibration suppression by the Current-Slope method is not applied, and compensation is applied to the control configuration example of FIG. An operation chart is shown. Since there is no compensation, an example in which the speed is oscillating is shown.

  FIG. 11 is a time chart example of starting and reversing operation in a low speed region by a current drawing method when vibration suppression by the Current-Slope method is applied. By applying the control configuration example of FIG. This shows the effect that the continuous vibration component of is suppressed while being attenuated.

(3) Embodiment 1
This embodiment has the apparatus configuration shown in FIG. The figure shows a current control system based on the deviation between the current command of the rotation coordinate (γ-δ coordinate) by the current control unit 1 and its feedback signal, and the voltage command (γ− δ coordinate) is converted into fixed coordinates of voltages v _a ^* and v _b ^* by the coordinate conversion unit 2 using the estimated phase (θ ^), and the two-phase / three-phase conversion unit 3 further converts the three-phase voltage v _u ^* , The PWM waveform is converted into v _v ^* and v _w ^* , and these voltages are used as a three-phase voltage command to generate a PWM waveform by the triangular wave carrier comparison method using the PWM control unit 4 and the triangular wave oscillation unit 5. The PWM current output of the PM motor 7 is obtained in the inverter 6 as each phase gate command.

  Next, detection of a motor current used in the apparatus configuration of FIG. 1 will be described with reference to FIG. FIG. 5 shows an example of a current flow path in the zero voltage vector period V0, which is a motor current detection period. In the figure, in a voltage-type three-phase inverter having a six-arm configuration, the case where the switching side (Sx, Sy, Sz) of the lower arm is turned on (ON) in all three phases is called a V0 period. If the terminal current of the motor is positive in the U phase and negative in the V and W phases, the inductance component inside the motor will continue to flow, so reverse conduction with the switching element as shown by the dotted line A current flows as if the motor terminal is short-circuited via the diode.

  FIG. 6 shows an example of a current path in the zero voltage vector period V7 which is another detection period of the motor current. In the figure, since the switching elements (Su, Sv, Sw) of the upper arm are in the conductive (ON) state, the current paths are different, but the motor terminals are also short-circuited as in FIG. Thereby, it can be understood that the voltage component is approximated to zero in the equation (8).

Next, FIG. 7 shows the period of zero voltage vectors V0 and V7 in this case, taking as an example the case of PWM waveform generation in the triangular wave carrier comparison method. When a U-phase voltage command v _u (solid line), a V-phase voltage command v _v (broken line), and a W-phase voltage command v _w (one-dot chain line) as shown in FIG. An operation example of six gate signals of Sz is shown. In the above three phases, the V0 period in which the lower arm switching elements (Sx, Sy, Sz) are conductive (ON) and the upper arm switching element (Su, Sv, Sw) are conductive (ON) V7. The period is specifically shown in this figure. Since the voltage command is small at low speeds, the three-phase voltage commands are also concentrated almost at the center level with respect to the amplitude of the triangular carrier waveform. Therefore, it can be read that both the V0 period and the V7 period are longer than the other voltage vector periods.

In order to construct a current control system using PWM control based on such a triangular wave carrier comparison method, carrier signals at times t _m , t _{m + 1} , t _{m + 2} and t _{m + 3} in FIG. A method is used in which current is detected in synchronization with the top and bottom vertices.

FIG. 8 shows the relationship between the current detection time for detecting the current differentiation proposed in the present embodiment and the detection result of the synchronous current equivalent to the current component from which the PWM ripple is removed when the PWM waveform shown in FIG. 7 is generated. Is shown. In FIG. 8, times t _m , t _{m + 1} , t _{m + 2} , t _{m + 3} are detection times used for conventional current control as described above, and detect currents such as i _um. ing. In contrast, a symmetrical detection time is added to each vertex. For example, t ₂ and t ₃ for time t _{m + 1} , t ₄ and t ₅ for time t _{m + 2} , and currents i _u2 and i _{u3 at} this time are detected. Let ΔTp be the time difference between the detection times of t ₂ and t ₃ and t ₄ and t ₅ . This time width is determined in consideration of the maximum amount of voltage amplitude.

  FIG. 7 shows an example in which the V0 period and the V7 period necessary for the third and fourth embodiments described later are set, but in the first embodiment and the second embodiment described later, current differentiation is performed. Can be detected, and there is no problem even if some deviation occurs rather than in the middle.

  The waveform of the three-phase current and the current detection point at this time are indicated by circles and squares in the lower part of FIG. This figure is an example of a chart when it is steady and not affected by the dead time. However, the amount of current change during each zero voltage vector period is constant, and the difference between the two current detection values is the current time. It can be seen that it is proportional to the differentiation.

Based on the above description, the basic configuration of the current drawing method in FIG. 1 will be described. The current control unit 1 includes a current command including a current value I _γ ^* = | I ₁ ^* |, I _δ ^* = 0 and an angular frequency ω ^* , and a current i _u detected by the current sampler 8A of the fundamental wave current detection unit 8. , I _v , i _w are converted into two phases by the three-phase / two-phase conversion unit 8B, the coordinates are converted by the fixed / rotation coordinate conversion unit 8C, and further, the fundamental wave detection current i _γ detected synchronously by the current sampler 8D. , I _{δ is used} for current control. Since the current control unit 1 outputs the biaxial voltage components v _γ ^* and v _δ ^* on the rotational coordinates, which are control references, the rotational coordinate conversion unit (e ^−jθ ) 2 ^outputs the voltage values v of the fixed coordinate system. After conversion to _a ^* and v _b ^* , two-phase / three-phase conversion by the two-phase / three-phase conversion unit 3 or zero-phase modulation is applied to apply three-phase voltage commands (v _u ^* , v _v ^* , v _w ^* ), and finally, the PWM controller 4 converts it into a PWM waveform to generate a gate signal for controlling the switching elements of the main circuit of the inverter 6. The dead time, which is a major factor of current error, is added by the PWM control unit 4. Then, the inverter 6 outputs a PWM voltage pattern equivalent to the PWM voltage command in accordance with the gate signal and supplies power to the PM motor 7.

The three-phase current of the PM motor 7 is detected by converting it into a signal with a current detector (HCT). When the current sampler 8 is configured by digital control, such as a microcomputer configuration, the analog current detection signal is sampled and held, and then A / D converted, and then three-phase / two-phase conversion or rotational coordinate conversion e ^jθ. The detected currents i _γ and i _δ of the γ-δ coordinate component are obtained via (exp (jθ)). This detected current is used for the above-described current control.

A triangular wave oscillating unit 5 necessary for PWM generation and a trigger signal generating unit 9 synchronized therewith generate a triangular wave necessary for PWM control and current detection times t _m , t _2n , t _{(2n + 1)} . Further, the phase information θ input to the rotation coordinate transformation is obtained by integrating the angular frequency command ω ^* without correction by the integration unit 10 by time integration and adding an estimated phase θ ^ (in the drawing, a ^ mark is attached to the top of θ. Description) get.

  The above is the configuration of the conventional current drawing method. In the present embodiment, the following functions are newly added thereto.

To generate a trigger signal for detecting a current having a time difference ΔT _p between time t _(2n) and t _{(2n + 1)} , a trigger signal having t _(2n) as the first time to detect the current difference, The trigger signal generation unit 9 generates a trigger signal whose time is t _{(2n + 1)} . gamma-axis current difference detector 11, the current detection value at time t _2n based on the generation of the trigger signal i _γ (t _2n) and the detection time t _{(2n + 1)} current detection value i _gamma of (t _{2n + 1)} The current difference i _γ (t _{2n + 1} ) −i _γ (t _2n ) is obtained. This corresponds to ΔTp (p · i _γ ), which is the product of the differential current component of equation (14) and time.

The current difference component calculation unit 12 based on the resistance and speed electromotive force is a part that calculates the (ΔT _p / L) (R · i _γ −ω ^* · L · i _δ ) component of the equation (14). The speed electromotive force is calculated from the voltage drop due to the motor resistance R, the inductance L, and the frequency command ω ^* from the detected current values i _γ and i _δ , and the time difference ΔTp between the inductance L and the current difference is used. The generated current difference component is converted, and the output of the voltage difference component calculation unit 12 is added to the current difference i _γ (t _{2n + 1} ) −i _γ (t _2n ) output from the γ-axis current difference detection unit 11. 11 is generated in the opposite polarity direction to the terminal voltage vector, and here, in consideration of this, polarity inversion and subtraction are combined into a simplified addition operation.

What the band pass filter 13 which suppresses the high frequency band and low frequency band from the output of the current difference component calculation unit 12, by multiplying the correction gain K _p to the filter output by the correction gain multiplication unit 14, a frequency weighting function multiplier 15 is multiplied by the angular frequency weighting function K _ω set in place of the division of the angular frequency command ω ^* to calculate the corrected angular frequency Δω _p ^* , and this is added to the angular frequency command ω ^*. , A new angular frequency command ω _p ^* according to the equation (14) is calculated. These 13-15 comprise the correction | amendment angular frequency command calculating part.

  In the above, the correlation between the current control timing and the compensation calculation timing by current differentiation is not clearly described, but this stabilization method compensates only a low frequency component as compared with current control. Therefore, even if there is a delay time of about the PWM period in the feedback of the stabilization compensation, there is no significant influence. Therefore, there is no need to limit the relationship with current control.

(4) Embodiment 2
This embodiment has the apparatus configuration shown in FIG. This figure differs from the configuration of FIG. 1 in that the current difference component calculation unit 12 is omitted. That is, in consideration of the fact that the amount of change in the (R · i _γ −ω ^* · L · i _δ ) component of the equation (14) is small, this is suppressed by the subsequent band-pass filter 13. The current difference component calculation unit 12 is omitted.

(5) Embodiment 3
This embodiment has the apparatus configuration shown in FIG. This figure is that the calculation function of the current difference component calculation unit 12 is changed from the configuration of FIG.

In the configuration of FIG. 1, the current components i _γ and i _δ used for the calculation of (R · i _γ −ω ^* · L · i _δ ) in the equation (14) are detected values of the synchronous current that is the apex of the triangular wave carrier. However, if the detection times t _2n and t _{(2n + 1)} are set to equal time differences before and after the time t _m , the two current components i _γ and i detected for the current difference are used. _{Even if} the average of _δ is used, a value equivalent to the current detection at time t _m can be obtained.

Therefore, the current difference component average value calculator 12A calculates the average value {i _γ (t _{2n + 1} ) + i _γ (t _2n )} / 2 of the current components i _γ and i _δ detected for the current difference and {i _δ (t _{2n + 1} ) + i _δ (t _2n )} / 2 is replaced with a current detection value equivalent to the current components i _γ and i _δ at time t _m , and this average value {i _γ (t _{2n + 1} ) + i _γ (t _2n )} / 2, {i _δ (t _{2n + 1} ) + i _δ (t _2n )} / 2, and the voltage drop due to the motor resistance R at the time when the current difference between the upper arm and the lower arm is detected Then, the speed electromotive force is calculated from the inductance L and the angular frequency command ω ^*, and converted into a current difference component generated by the voltage component using the time difference ΔTp between the inductance L and the current difference. The output of the average calculation unit 12A for the current difference component is added to the output of the average value of the current difference by the addition unit 12C.

(6) Embodiment 4
This embodiment has the apparatus configuration shown in FIG. The calculation in FIG. 3 always uses the latest values for the components (R · i _γ −ω ^* · L · i _δ ) and {i _γ (t _{2n + 1} ) −i _γ (t _2n )}. Actually, different conditions of V0 and V7 are alternately generated in the zero voltage vector period. Therefore, there is a possibility that the voltage drop of the main circuit of the inverter is different, or the current value of the fundamental wave component is fluctuated due to the influence of dead time or the like.

  Therefore, in the moving average value calculation unit 12B of the current difference component of this embodiment, the moving average value calculation unit 12A shown in FIG. 3 takes a moving average of two points of the latest detected current differential value and the previous current differential value. A function is added, and disturbance components are suppressed by statistical processing.

  4 shows a case where the arithmetic unit 12A of the third embodiment shown in FIG. 3 is changed to 12B, but the configuration of the first embodiment shown in FIG. 1 or the configuration of the second embodiment shown in FIG. It is possible to apply similar improvements to this.

  In addition, the above-described apparatus configuration has shown an example of the calculation unit configuration based on the above-mentioned equation (14). However, in principle, if it conforms to the equation (11), the position of the sampler can be converted into coordinates. It is clear that the same effect can be obtained in different forms other than the first to fourth embodiments, such as moving forward, changing the order of calculations, and changing the calculation format by including an inductance component in the coefficient. .

  According to the first to fourth embodiments, vibration suppression is applied when the current pulling method of starting the magnetic pole phase θ of a PM motor having no magnetic saliency following the current phase is applied. Suppression of speed vibration (shaft vibration) and step-out can be prevented.

(7) Simulation A simulation was performed by using the apparatus configuration of the fourth embodiment and a conventional current drawing method as an example, and the effect of the present invention was confirmed.

  FIG. 10 is a simulation waveform diagram of the conventional basic current drawing method. In this case, the variable speed frequency command establishes a direct current by current control so that the current value is commanded up to the time of 0.05S, and 0% to 10% in the period of 0.05S to 0.25S. The frequency was continuously increased to the speed of. Then, between time 0.4S and 0.65S, the frequency is continuously reduced from 10% to 0% speed, and then from 0% to -10% speed from time 0.65S to 0.85S. The frequency is increased in the reverse direction.

  The first stage of the chart of FIG. 10 shows the biaxial current component. Here, the current command of the control axis γ-δ axis and the actual current components of the actual magnetic pole axis d-axis and q-axis are shown. The actual current converted to the control reference γ-δ coordinate component is omitted because a current substantially equal to the command value flows. Even if the current of the control axis is stable, there is a phase difference ψ between the actual magnetic pole axis d-axis and the γ-axis, so the actual d-axis current and the q-axis current vary according to this phase difference ψ. From the pulsation state of the d-axis and q-axis currents, it can be estimated that the axis phase is oscillatory.

  The second stage is a three-phase alternating current waveform, and it can be confirmed that a current flows with a constant amplitude according to the current command and according to the frequency command.

  The third stage shows the torque generated in the rotor of the motor. This torque includes not only the output shaft component of the motor but also a torque component for accelerating / decelerating the inertia moment of the rotor of the motor. From the torque waveform, it can be seen that it is vibrational.

  The fourth stage compares the frequency command and the actual speed of the motor rotor. In contrast to the trapezoidal waveform frequency command, a positive and negative deviation occurs in the rotor speed.

  The fifth level is a current difference component, a component obtained by applying a band pass filter to the current difference component, and a frequency weight multiplied by this. For the current difference component, the direct current component is suppressed, the compensation amount is forcibly suppressed to zero when the speed is near zero, and the polarity of the compensation amount becomes positive or negative when the polarity of the frequency is reversed. It can be confirmed that it is reversed.

  However, since the compensation calculation amount has not yet been added to the frequency, the generated vibration is not attenuated and tends to be sustained.

  FIG. 11 is an example in which the vibration suppression of the fourth embodiment of the present invention is applied, and the above frequency correction amount is added and corrected to the command frequency.

  It can be clearly seen that the vibration component is suppressed by looking at the speed of the fourth stage. As a result, the fluctuation of the phase angle ψ between the γ-axis and the d-axis becomes gradual, and as a result, even if it starts to oscillate like the torque waveform in the second stage, a characteristic that gently attenuates can be realized.

  This vibration suppression effect is also effective at the reverse frequency. Further, although the compensation is stopped in the zero speed region, when the frequency command passes through the zero speed region in a short time, it can pass through the zero speed region before the vibration becomes large.

[0005]
An accurate magnetic pole phase cannot be estimated only by the magnetic pole phase estimation method using velocity electromotive force. Further, since the method (b) uses magnetic saliency, it cannot be applied as it is to a non-salient pole machine having no magnetic saliency. Further, when the current drawing method is applied, there remains a problem that causes shaft torsional vibration. Further, the method (c) has a problem that a large capacity power converter is required.
[0022]
It is an object of the present invention to accurately estimate the magnetic pole phase of a PM motor having no magnetic saliency up to a low speed such as 10% or less of the rated speed, and to perform velocity oscillation by a current drawing method based on the estimation of the magnetic pole phase. An object of the present invention is to provide a position sensorless control device for a permanent magnet synchronous motor that can suppress (shaft vibration) and prevent step-out.
[0023]
Since the current differential information of the zero voltage vector period used in the present invention is not affected by the dead time, the speed is reduced from a lower speed than the magnetic pole phase estimation method using a voltage command or the like input to the conventional PWM control. Focusing on the possibility that electromotive force information can be obtained, current differential information during zero voltage vector period is detected to obtain speed electromotive force information, and the magnetic pole phase is estimated by the magnetic pole phase estimation method using this speed electromotive force. Based on this estimated magnetic pole phase, velocity vibration (shaft vibration) is suppressed when the current drawing method is applied to a PM motor with no magnetic saliency. And
[0024]
(1) A synchronous motor using a permanent magnet that is a non-salient pole machine and has no damper winding as a field source is started by following the current phase of the magnetic pole phase θ, and the vibration information of the magnetic pole phase θ is estimated. A control device for a permanent magnet synchronous motor having means for correcting an angular frequency command,
The means for correcting the angular frequency command is:
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detection unit (8) for detecting current at times t _2n and t _{(2n + 1)} at least two points during each of the zero voltage vector periods;
A current for calculating a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points.

[0006]
A difference detector (11);
The speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the value obtained by dividing the time difference ΔTp by the inductance L A current difference component calculation unit (12) that outputs a current difference component that is a multiplication value of the speed electromotive force;
Since the output of the current difference detection unit and the output of the current difference component calculation unit are a positive value and a negative value, the difference between the two types of current difference is obtained by addition, and the direct current component is removed by a filter. to the correction gain K _p multiplies, further the correction gain K _p corrected angular frequency Δω by frequency output obtained by multiplying the in multiplying the angular frequency weighting function K _omega to suppress the zero magnitude of the compensation amount in the vicinity of zero calculating a _p ^*, which was added to the angular frequency command omega ^* calculates a new angular frequency command omega _p ^* corrected angular frequency command calculation section (13 to 15),
It is provided with.
[0025]
(2) The magnetic pole phase θ of a synchronous motor that uses a permanent magnet that is a non-salient pole machine and has no damper winding as a field source is started by following the current phase, and vibration information of the magnetic pole phase θ is estimated. A control device for a permanent magnet synchronous motor having means for correcting an angular frequency command,
The means for correcting the angular frequency command is:
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental current detection unit (8) that detects current at times t _2n and t _{(2n + 1)} at least two points during the zero voltage vector period;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
The current differences by removing the DC component is multiplied by the correction gain by the filter, further the correction gain K _p in the vicinity of the frequency is zero in the output obtained by multiplying the compensation amount of the angular frequency weighting function to suppress the magnitude to zero K _omega A corrected angular frequency command calculation unit (13 to 15) that calculates a corrected angular frequency Δω _p ^* by multiplying by and adds this to the angular frequency command ω ^* to calculate a new angular frequency command ω _p ^* ;
It is provided with.

[0007]
[0026]
(3) A synchronous motor using a permanent magnet which is a non-salient pole machine and has no damper winding as a field source is started by following the current phase of the magnetic pole phase θ, and the vibration information of the magnetic pole phase θ is estimated. A control device for a permanent magnet synchronous motor having means for correcting each frequency command,
The means for correcting the angular frequency command is:
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detection unit (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points symmetrical with respect to the upper and lower vertices of the triangular wave carrier signal during the zero voltage vector period;
An average value of current components of current detection values at two or more times t _2n and t _{(2n + 1)} symmetrical with respect to the vertices of the zero voltage vector period (V7, V0) of the upper arm and the lower arm is obtained. A speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the value obtained by dividing the time difference ΔTp by the inductance L An average value calculation unit (12A) that averages and outputs components of two types of detection periods of V7 and V0 for a current difference component that is a product of the speed electromotive force,
An average value output of current differences of current detection values at two or more times t _2n and t _{(2n + 1)} symmetrical with respect to the vertex of a zero voltage vector period output from the fundamental wave current detection unit (8) and the current difference Since the output of the component average value calculation unit (12A) is a positive value and a negative value, respectively, an error amount of these two kinds of current differences is obtained by addition (12C), and the direct current component is removed by using this added value with a filter. Then, the correction gain K _p is multiplied, and when the frequency multiplied by the correction gain K _{p is} multiplied by an angular frequency weighting function K _ω that suppresses the amount of compensation to zero when the frequency is near zero, the correction angular frequency Δω _p ^* the calculated corrected angular frequency command calculating unit for calculating the said angular frequency command omega ^* new angular frequency command omega is added to the _p ^* At the (13-15),
It is provided with.
[0027]
(4) Start a synchronous motor having a non-salient pole machine having a permanent magnet without a damper winding as a field source by making the magnetic pole phase θ follow the current phase, and estimate the vibration information of the magnetic pole phase θ. A control device for a permanent magnet synchronous motor having means for correcting each frequency command.

[0008]
There,
The means for correcting the angular frequency command is:
During the zero voltage vector period in the output voltage period by PWM control, the zero voltage vector period in which the switching element of the upper arm is conducted in all three phases or the zero voltage vector period in which the switching element of the lower arm is conducted in all three phases A fundamental wave current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points at
A moving average of current components at times t _2n and t _{(2n + 1)} at two or more points in two types of detection periods of the upper arm zero voltage vector period (V7) and the lower arm zero voltage vector period (V0), A speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the value obtained by dividing the time difference ΔTp by the inductance L A moving average value calculation unit (12B) that outputs a current difference component that is a multiplication value of the speed electromotive force;
The average value output and moving average value of the current difference of the current detection values at two or more times t _2n , t _{(2n + 1)} symmetrical with respect to the vertex of the zero voltage vector period output by the fundamental wave current detection unit (8). Since each output of the calculation unit (12B) is a positive value and a negative value, an error amount between these two types of current difference is obtained by addition (12C), and the DC value is removed from the addition value by a filter to obtain a correction gain. A corrected angular frequency Δω _p ^* is calculated by multiplying the output multiplied by K _p and further multiplied by the correction gain Kp by an angular frequency weighting function K _ω that suppresses the amount of compensation to zero when the frequency is near zero, A corrected angular frequency command calculation unit (13 to 15) for calculating a new angular frequency command ω _p ^* by adding this to the angular frequency command ω ^* ,
It is provided with.
[0028]
As described above, according to the present invention, the speed electromotive force information is obtained using the current differential information in the zero voltage vector period, and the magnetic pole phase is estimated by the magnetic pole phase estimation method using the speed electromotive force. By adding vibration suppression to the control method in the case of applying the current pulling method that starts the magnetic pole phase θ of a PM motor with no magnetic saliency following the current phase based on the magnetic pole phase Based on the estimation of magnetic pole phase of PM motor without magnetic saliency

Claims (4)

A permanent magnet having a phase estimation means for starting the magnetic pole phase θ of a synchronous motor having a field source as a permanent magnet which is a non-salient pole machine and does not have a damper winding, follows the current phase and estimates the magnetic pole phase θ. A position sensorless control device for a synchronous motor,
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points during each of the zero voltage vector periods;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the time difference ΔTp is divided by the inductance L. A current difference component calculation unit (12) that converts a current difference component that is a product of the value and the speed electromotive force;
An output obtained by adding the output of the current difference detection unit and the output of the current difference component calculation unit, removing the DC component by a filter, multiplying the sum by a correction gain K _p , and further multiplying the correction gain K _p angular frequency weight multiplied by the function K _omega calculates the correction angle frequency [Delta] [omega _p ^*, the correction angular frequency command calculator for calculating the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this to the ( 13-15)
A position sensorless control device for a permanent magnet synchronous motor, comprising: A permanent magnet having a phase estimation means for starting the magnetic pole phase θ of a synchronous motor having a field source as a permanent magnet which is a non-salient pole machine and does not have a damper winding, follows the current phase and estimates the magnetic pole phase θ. A position sensorless control device for a synchronous motor,
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points during the zero voltage vector period;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A correction angular frequency Δω _p ^* is calculated by multiplying an output obtained by removing the DC component from the current difference by a filter and multiplying the correction gain by the filter, and multiplying the output multiplied by the correction gain K _p by an angular frequency weighting function K _ω. the was added to the angular frequency command omega ^* calculates a new angular frequency command omega _p ^* corrected angular frequency command calculation section (13 to 15),
A position sensorless control device for a permanent magnet synchronous motor, comprising: A permanent magnet having a phase estimation means for starting the magnetic pole phase θ of a synchronous motor having a field source as a permanent magnet which is a non-salient pole machine and does not have a damper winding, follows the current phase and estimates the magnetic pole phase θ. A position sensorless control device for a synchronous motor,
The phase estimation means includes
In the output voltage period by PWM control, in the zero voltage vector period (V7) in which the switching element of the upper arm is conducted in all three phases, or in the zero voltage vector period (V0) in which the switching element of the lower arm is conducted in all three phases, A fundamental wave current detector (8) for detecting currents at times t _2n and t _{(2n + 1)} at least two points symmetrical with respect to the upper and lower vertices of the triangular wave carrier signal during the zero voltage vector period;
The average value of the current difference of the current detection values at time t _2n and t _{(2n + 1)} at two or more points symmetrical with respect to the vertices of the zero voltage vector period (V7, V0) of the upper arm and the lower arm. The speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L, and the angular frequency command ω ^* , and the time difference ΔTp is calculated by the inductance L. An average value calculation unit (12A) of a current difference component that is converted into a current difference component that is a multiplication value of the divided value divided by the speed electromotive force;
The average value output of the current difference of the current detection unit and the output of the current difference component calculation unit are added (12C), the DC value is removed by a filter, and the correction gain _Kp is multiplied. K _p and calculates the correction angle frequency [Delta] [omega _p ^* by multiplying the angular frequency weighting function K _omega output multiplication, it calculates the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this correction angle A frequency command calculation unit (13 to 15);
A position sensorless control device for a permanent magnet synchronous motor, comprising: A permanent magnet having a phase estimation means for starting the magnetic pole phase θ of a synchronous motor having a field source as a permanent magnet which is a non-salient pole machine and does not have a damper winding, follows the current phase and estimates the magnetic pole phase θ. A position sensorless control device for a synchronous motor,
The phase estimation means includes
During the zero voltage vector period in the output voltage period by PWM control, the zero voltage vector period in which the switching element of the upper arm is conducted in all three phases or the zero voltage vector period in which the switching element of the lower arm is conducted in all three phases A fundamental wave current detector (8) that detects currents at times t _2n and t _{(2n + 1)} at least at two points;
A current difference detector (11) that calculates a current difference from current detection values at times t _2n and t _{(2n + 1)} at two or more points;
A moving average of current differences at times t _2n and t _{(2n + 1)} at two or more points in two types of detection periods of the upper arm zero voltage vector period (V7) and the lower arm zero voltage vector period (V0). Then, the speed electromotive force is calculated from the time difference ΔTp between the times t _2n and t _{2n + 1} , the voltage drop due to the resistance R of the synchronous motor, the inductance L and the angular frequency command ω ^* , and the time difference ΔTp is calculated by the inductance L. An average value calculation unit (12B) of current difference components to be converted into a current difference component that is a multiplication value of the divided value divided by the speed electromotive force;
Wherein adding the output of the current difference and the current difference component calculation unit of the current detecting portion (12C), the added value by removing the DC component in the filter multiplied by the correction gain K _p, multiplied further the correction gain Kp Output angular frequency weight multiplied by the function K _omega calculates the correction angle frequency [Delta] [omega _p ^*, the correction angular frequency command calculator for calculating the angular frequency command omega ^* to a new angular frequency command omega _p ^* by adding this to the ( 13-15)
A position sensorless control device for a permanent magnet synchronous motor, comprising:

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