JP6565484B2 - Power converter - Google Patents

Description

  The present invention relates to a control device for a permanent magnet type synchronous motor that does not have a magnetic pole position detector, and more particularly to a technique for reducing torque shock when switching a control method according to speed.

In order to reduce the cost of a control device for a permanent magnet type synchronous motor (hereinafter also referred to as PMSM), so-called sensorless vector control that operates without using a magnetic pole position detector of a rotor has been put into practical use. As is well known, sensorless vector control realizes torque control and speed control by estimating the magnetic pole position and speed of the rotor from information on the terminal voltage and current of the motor, and performing current control based on the estimated position. .
However, since sensorless vector control has a problem in stability at low speed, in the low speed range, the current amplitude is constant and the angular velocity of the current is controlled to the speed command value, thereby pulling the PMSM rotor into the current. Driving techniques may be applied. Such a control method is hereinafter referred to as current drawing control.

A conventional technique for switching the PMSM control method from current pull-in control to sensorless vector control without shock is described in, for example, Patent Document 1.
In the technique described in Patent Document 1, when switching from current pull-in control to sensorless vector control, the expansion induced voltage is calculated from the angular velocity of the current of PMSM, the terminal voltage, and the current command value. The magnetic pole position estimation error is calculated from the angle, and the torque is calculated from the angular velocity of the PMSM current, the expansion induced voltage calculated value, and the current command value, and the magnetic pole position estimated value, speed estimated value, terminal voltage command value, current command value In addition, by initializing the torque command value, torque shock at the time of switching the control method is reduced.

Japanese Patent No. 5277724 (Claim 1, paragraphs [0036] to [0046], FIG. 9 etc.)

In the technique described in Patent Document 1, the current command value changes suddenly when switching from current pull-in control to sensorless vector control. If there is no error in the torque calculation value or electrical constant, the current command value can be calculated so that the torque is the same before and after switching the control method. Will not occur.
However, when there is an error in the torque calculation value or the electrical constant, there is a problem that a shock occurs when the control method is switched, and smooth variable speed operation cannot be performed.

  Therefore, the problem to be solved by the present invention is that a control device for a permanent magnet synchronous motor that reduces torque shock when switching the control method from current pull-in control to sensorless vector control, and enables smooth variable speed operation. Is to provide.

In order to solve the above-mentioned problem, the invention according to claim 1 regards the current, terminal voltage, and magnetic flux of a permanent magnet type synchronous motor driven by a power converter and not having a magnetic pole position detector as a vector, In the control device for a permanent magnet type synchronous motor, wherein the current and terminal voltage of the motor are decomposed into the vector components to control the power converter,
As a control mode of the electric motor,
The first operation mode for controlling the angular velocity of the current command value of the motor to the speed command value, and the rotor magnetic pole position estimated value and the speed estimated value calculated from the current and terminal voltage of the motor, A second operation mode for controlling the angular velocity to a speed command value,
When switching from the first operation mode to the second operation mode,
Means for calculating a magnetic pole position estimation error from the electric current of the motor, the terminal voltage, and the angular velocity of the current command value;
Means for calculating an initial value of the second current command value in the second operation mode using the first current command value in the first operation mode and the magnetic pole position estimation error;
Means for calculating a first current command value in the second operation mode from a torque command value of the motor;
Means for initializing a current compensation value using a deviation between the initial value of the second current command value in the second operation mode and the first current command value in the second operation mode.
Here, the first current command value in the first operation mode described above corresponds to _Iapple ^* , 0 in Equation 13 described later, and the magnetic pole position estimation error is δ _comp (= δ _Eex = −δ _Ψex ). The second current command value in the second operation mode corresponds to i _d ^* (i _γ ^* ), i _q ^* (i _δ ^* ), and the second current command value in the second operation mode. Is equivalent to i _dpull ^* and i _qpull ^* in Expressions 13 and 14, and the first current command value in the second operation mode is equivalent to i _dVector ^* and i _qVector ^* in Expression 14. The current compensation values correspond to i _dcomppul and i _qcomppull .
In the present invention, in the second operation mode,
Means for reducing the current compensation value to zero over time;
Means for adding the first current command value of the second operation mode and the current compensation value to calculate a second current command value;
And a means for controlling the electric current of the electric motor to the second current command value.
Here, as described in claim 5, for example, the first operation mode is a current pull-in control mode, and the second operation mode is a sensorless vector control.
According to the present invention, the current command value at the time of switching from the current pull-in control mode to the sensorless vector control can be made continuous, and the torque shock can be reduced as compared with the conventional case.

According to a second aspect of the present invention, there is provided the control device for the permanent magnet type synchronous motor according to the first aspect, further comprising means for limiting an upper limit value of the second current command value in the second operation mode. .
According to the present invention, it is possible to prevent the expansion induced voltage from becoming small when switching from the current drawing control mode to the sensorless vector control, and to stabilize the magnetic pole position / speed estimation using the expansion induced voltage.

According to a third aspect of the present invention, there is provided the permanent magnet type synchronous motor control apparatus according to the first or second aspect, wherein the upper limit value of the current compensation value is limited in the second operation mode. The upper limit value of the current compensation value is calculated from the deviation between the torque calculation value and the torque command value when switching from the first operation mode to the second operation mode, and the initial value of the current compensation value. And.
According to the present invention, it is possible to reduce the torque control error caused by the current compensation value when the load fluctuates.

The invention according to claim 4 is the control apparatus for the permanent magnet type synchronous motor according to any one of claims 1 to 3, wherein the upper limit value of the current compensation value is limited in the second operation mode. And means for calculating an upper limit value of the current compensation value from the speed estimated value and the initial value of the current compensation value.
ADVANTAGE OF THE INVENTION According to this invention, it can prevent that a terminal voltage is saturated with an electric current compensation value at the time of middle-high speed driving | operation of an electric motor.

  According to the present invention, it is possible to reduce the shock of torque when switching the control method as compared with the prior art, and it is possible to realize a smooth variable speed operation of the PMSM from a low speed region to a medium high speed region.

It is a control block diagram which shows embodiment of this invention. It is a figure which shows the definition of a coordinate axis. It is a vector diagram explaining the principle of the switching process from current drawing control to sensorless vector control. It is a block diagram which shows the structure of the electric current command calculating part in FIG. It is a figure for demonstrating the reduction coefficient of a current compensation value limiting value. It is a figure for demonstrating the reduction coefficient of a current compensation value limiting value.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
FIG. 1 is a control block diagram showing an embodiment of the present invention. This embodiment is an improvement over the prior art described as the second embodiment (FIG. 2) of Patent Document 1, and switches the PMSM control method from current pull-in control to sensorless vector control using extended induced voltage. The case relates to the control technology.

PMSM can realize high-performance torque control and speed control by controlling current and voltage on the d and q axes which are orthogonal rotation coordinate axes synchronized with the rotor. Here, for the d and q axes, the N pole direction of the magnetic poles of the rotor is defined as the d axis, and the direction advanced by 90 ° from the d axis is defined as the q axis. However, in the sensorless control, the positions of the d and q axes cannot be directly detected. Therefore, the control calculation is performed by defining the γ and δ axes, which are the estimated axes of the d and q axes, in the control device.
Figure 2 is a diagram showing the definition of these axes, angular difference d, the angular velocity of the q-axis and omega _r, gamma, an angular velocity of [delta] axis as omega _1, d, and q-axis and gamma, [delta] axis _Let (position estimation error) be θ _err .

First, based on FIG. 1, the operation | movement at the time of the current drawing control in 1st Embodiment is demonstrated with the structure of a control apparatus.
In FIG. 1, the input of the changeover switch 23 is set to the “S1” side, and the angular velocity ω ₁ of the γ and δ axes is controlled to the velocity command value ω ^* . Electrical angle calculator 24, gamma integrates the angular velocity omega _1, calculates the angle theta ₁ of the δ-axis.
In addition, the current command calculation unit 41 calculates γ and δ-axis current command values i _γ ^* and i _δ ^* using Equation 1 in a state where the inputs of the changeover switches 21 a and 21 b are set to the “S1” side. In Formula 1, i _pull ^* is a current command value given during current pull-in control.

Low pass filter 22a, 22b is, gamma, [delta] -axis current value i _gamma ^*, extracts i _[delta] ^* in the low-frequency components gamma, [delta] -axis current value i _.gamma.f ^*, and outputs the i _{delta] f} ^*.
On the other hand, the coordinate converter 14 converts the phase current detection values i _u and i _w detected by the u-phase current detector 11u and the w-phase current detector 11w, respectively, to γ and δ based on the angle θ ₁ of the γ and δ axes. Coordinates are converted to detected shaft current values i _γ and i _δ .

The subtractors 19a and 19b calculate deviations between the γ and δ-axis current command values i _γf ^* and i _δf ^* and the γ and δ-axis current detection values i _γ and i _δ , respectively. The current regulator 20 calculates γ and δ-axis voltage command values v _γ ^* and v _δ ^* so as to eliminate these deviations.
The γ and δ-axis voltage command values v _γ ^* and v _δ ^* are converted into phase voltage command values v _u ^* , v _v ^* and v _w ^* by the coordinate converter 15 based on the angle θ ₁ of the γ and δ axes. Is done.

The rectifier circuit 60 converts the three-phase AC voltage of the three-phase AC power source 50 into a DC voltage and supplies the DC voltage to a power converter 70 such as an inverter. The PWM circuit 13 uses the DC input voltage E _dc of the power converter 70 detected by the input voltage detection circuit 12 and the phase voltage command values v _u ^* , v _v ^* , v _w ^* to the power converter 70. Generate a gate signal. The power converter 70 controls the internal semiconductor switching element based on these gate signals, thereby controlling the terminal voltage of the PMSM 80 to the phase voltage command values v _u ^* , v _v ^* , v _w ^* .
By the above-described control, a current vector having an amplitude equal to I _pull ^* (see Equation 1) and an angular velocity equal to the speed command value ω ^* is generated, and the rotor of the PMSM 80 is drawn into this current vector, whereby the rotor so that the speed ω _r is controlled to the speed command value ω ^*.

Next, the operation at the time of sensorless vector control will be described. Here, the description will focus on parts that are different from the above-described current pull-in control, and description of overlapping parts will be omitted.
In FIG. 1, at the time of sensorless vector control, all inputs of the changeover switches 21a, 21b, and 23 are set to the “S2” side.

A deviation between the speed command value omega ^* and the speed estimated value omega ₁ is calculated by the subtracter 16, the speed regulator 17 calculates the torque command value tau ^* to eliminate the above deviation.
The current command calculation unit 18 calculates d and q-axis current command values i _d ^* and i _q ^* on the condition that the torque / current is maximized based on the torque command value τ ^* and a torque calculation value τ _pull described later. . The configuration and operation of the current command calculation unit 18 will be described later.

The d and q-axis current command values i _d ^* and i _q ^* calculated by the current command calculation unit 18 are low-pass filters as γ and δ-axis current command values i _γ ^* and i _δ ^* via the changeover switches 21a and 21b. It is input to 22a and 22b.
The operations of the subtractors 19a and 19b, the current adjuster 20, and the coordinate converters 14 and 15 after the low-pass filters 22a and 22b are the same as those in the case of the current pull-in control, and γ and δ-axis currents i _γ and i are obtained by these operations. _The phase voltage command values v _u ^* , v _v ^* , and v _w ^* are calculated so that _δ matches the γ and δ-axis current command values i _γ ^* and i _δ ^* .

なお、拡張誘起電圧の演算は、数式２におけるγ軸電圧指令値ｖ_γ ^＊，δ軸電圧指令値ｖ_δ ^＊の代わりに、図示されていない電圧検出回路を用いてＰＭＳＭの相電圧または線間電圧を測定し、これらの測定値と位置推定値θ_１とを用いて演算したγ軸電圧ｖ_γ，δ軸電圧ｖ_δを使用しても良い。 Next, the expansion induced voltage calculator 31 in FIG. 1 uses γ, δ-axis voltage command values v _γ ^* , v _δ ^*, γ, δ-axis currents i _γ , i _δ , speed estimated value ω _1, etc. The γ and δ-axis expansion induced voltage calculation values are obtained from Equation 2.

Note that the expansion induced voltage is calculated by using a voltage detection circuit (not shown) instead of the γ-axis voltage command value v _γ ^* and the δ-axis voltage command value v _δ ^* in Equation 2, and the PMSM phase voltage or line-to-line Voltages may be measured, and γ-axis voltage v _γ and δ-axis voltage v _δ calculated using these measured values and position estimated value θ ₁ may be used.

この拡張誘起電圧の角度δ_Ｅｅｘは、位置推定誤差θ_ｅｒｒに等しくなるため、位置推定誤差演算値θ_{ｅｒｒｅｓｔ}を数式４とする。

Angle calculator 32, gamma, [delta] axis extended electromotive force calculation value _Ψ exγ, from _Ψ exδ, the angle [delta] _Eex expansion induced voltage is calculated by Equation 3.

Since the angle δ _Eex of the extended induced voltage is equal to the position estimation error θ _err , the position estimation error calculation value θ _errest is _expressed by Equation 4.

The speed estimator 33 is configured by a PI controller that receives the position estimation error calculation value θ _errest , and the speed estimator 33 is equal to the speed estimated value ω ₁ (γ and δ-axis angular velocities according to Equation 5). ) Is calculated. In addition, the electrical angle calculator 24 calculates the magnetic pole position estimated value θ ₁ (equal to the angles of the γ and δ axes) by integrating the speed estimated value ω ₁ .

The angle difference θ _err between the γ and δ axes and the d and q axes is converged to zero by the action of the above-described extended induced voltage calculator 31, angle calculator 32, speed estimator 33, and electrical angle calculator 24. Can do. In other words, the magnetic pole position and speed of the PMSM 80 can be accurately calculated, whereby the torque and speed can be accurately controlled.

Next, the principle of switching processing from current pull-in control to sensorless vector control will be described with reference to the vector diagram of FIG.
In the current drawing control shown in FIG. 3A, the angle difference θ _err (equal to δ _Eex ) between the d and q axes and the γ and δ axes is an increasing function of the load torque. On the other hand, in the case of the sensorless vector control shown in FIG. 3B, the angle difference θ _err is controlled to be substantially zero.

Therefore, when switching from the current pull-in control to the sensorless vector control, the magnetic pole position estimation value theta ₁ is initialized to the angle of the d-axis, so that at this time the voltage vector v ^*, the current command vector i ^* continuous before and after the switching Initialize to. Further, the torque command value τ ^* is initialized with the torque before switching.
As shown in FIG. 3B, the current command vector i ^* is calculated by adding the current command vector i _Vector ^* during the sensorless vector control and the current compensation value vector i _compul . The initial value of the current compensation value vector i _compl is initialized by the deviation between the current command vector i ^* at the time of current pull-in control and the current command vector i _Vector ^* at the time of sensorless vector control, and the current compensation value vector i _compl Decrease to zero.

As a result, the current command vector i ^* becomes equal before and after switching from the current pull-in control to the sensorless vector control, and shifts to the current command vector i _Vector ^* during the sensorless vector control with time. As a result, the current command vector i ^* at the time of switching is continuous, and torque shock can be reduced.

Next, calculation processing when switching from current pull-in control to sensorless vector control in this embodiment will be described.
The expansion induced voltage calculator 31 calculates the γ and δ axis expansion induced voltages E _exγ and E _exδ in the state where the current drawing control is performed.
The expanded magnetic flux calculator 34 calculates γ and δ-axis expanded magnetic fluxes Ψ _exγ and Ψ _exδ from the γ and δ-axis expanded induced voltage calculated values E _exγ and E _{exδ according} to Equation 6.

なお、数式８の代わりに、前述した特許文献１に示されているごとく、数式３により求めたγ，δ軸拡張誘起電圧Ｅ_ｅｘγ，Ｅ_ｅｘδの角度δ_Ｅｅｘから電気角補償値δ_ｃｏｍｐを演算してもよい。 In addition, the angle calculator 35 calculates the angle δ _ψex of the expanded magnetic flux from the γ, δ-axis expanded magnetic flux ψ _exγ , ψ _exδ by Equation 7. Then, an electrical angle compensation value δ _comp is _obtained from the expanded magnetic flux angle δ _Ψex by Equation 8.

Instead of Equation 8, as shown in Patent Document 1 described above, the electrical angle compensation value δ _comp is calculated from the angle δ _{Eex of} the γ and δ-axis expansion induced voltages E _exγ and E _exδ obtained by Equation 3. May be.

なお、数式９の代わりに、特許文献１に示されているように、数式２により求めたγ，δ軸拡張誘起電圧演算値Ｅ_ｅｘγ，Ｅ_ＥＸδ、γ，δ軸電流ｉ_γ，ｉ_δ、及びγ，δ軸の角速度ω_１からトルク演算値τ_ｐｕｌｌを求めても良い。 On the other hand, the torque calculator 36 in FIG. 1 _obtains a torque calculation value τ _pull from Equation 9 from γ, δ extended magnetic flux calculation values ψ _exγ , ψ _exδ and γ, δ-axis currents i _γ , i _δ , and a speed regulator. The torque command value τ ^* output from 17 is initialized with the torque calculation value τ _pull .

Instead of Equation 9, as shown in Patent Document 1, γ, δ-axis expansion induced voltage calculation values E _exγ , E _EXδ , γ, δ-axis currents i _γ , i _δ , obtained by Equation 2, The torque calculation value τ _pull may be obtained from the angular velocity ω ₁ of the γ and δ axes.

Further, the output of the integrator constituting the speed estimator 33 is set to the angular speed ω ₁ of the γ and δ axes, and the output θ ₁ of the electrical angle calculator 24 is initialized by Expression 10.

数式１１，１２により、制御方法の切替前後で電流と端子電圧とが急変しないように、γ，δ軸電流指令値ｉ_γｆ ^＊，ｉ_δｆ ^＊及びγ，δ軸電圧指令値ｖ_γ ^＊，ｖ_δ ^＊を初期化することができる。 As the angle θ ₁ of the γ and δ axes is corrected by the electrical angle compensation value δ _comp , the γ and δ axis current command values i _γf ^* and i _δf ^* , which are the outputs of the low-pass filters 22a and 22b, and the current adjustment The γ and δ-axis voltage command values v _γ ^* and v _δ ^* , which are outputs of the device 20, are initialized by Equations 11 and 12, respectively.

According to Equations 11 and 12, the γ and δ-axis current command values i _γf ^* and i _δf ^* and the γ and δ-axis voltage command values v _γ ^* and v so that the current and the terminal voltage do not change suddenly before and after the control method is switched. _δ ^* can be initialized.

Next, details of the configuration and operation of the current command calculation unit 18, which is a feature of this embodiment, will be described based on the block diagram of FIG.
In FIG. 4, the current command calculator 121 outputs a desired torque from the torque command value τ ^* and the speed ω _r (sets the estimated speed value ω ₁ ) and outputs the terminal voltage to the maximum output of the power converter 70. The d and q-axis current command values i _dVector ^* and i _qVector ^* for controlling to below the voltage are calculated. Here, the d and q-axis current command values i _dVector ^* and i _qVector ^* are calculated so that the torque / current becomes maximum at a low speed when the terminal voltage is equal to or lower than the maximum output voltage of the power converter 70.

Further, the coordinate converter 122 calculates the current command values on the γ and δ axes in the current pull-in control from the current command value I _pull ^{* in the} current pull-in control and the electrical angle compensation value δ _comp in the sensorless vector control. The current command values I _dpull ^* and I _qpull ^* coordinate-converted to the q axis are calculated by Equation 13.

なお、ローパスフィルタ１２４ａ，１２４ｂの入力は零とし、ｄ，ｑ軸電流補償値ｉ_{ｄｃｏｍｐｐｕｌｌ}，ｉ_{ｑｃｏｍｐｐｕｌｌ}を時間と共に零まで減少させる。 At the time of switching from current pull-in control to sensorless vector control, a deviation between the d-axis current command value I _dpull ^* and the d-axis current command value i _dVector ^* is calculated by the subtractor 123a, and this deviation is the output of the low-pass filter 124a. The d-axis current compensation value i _dcomppull is initialized. Similarly, the deviation between the q-axis current command value I _qpull ^* and the q-axis current command value i _qVector ^* is calculated by the subtractor 123b, and the q-axis current compensation value i _qcomppull that is the output of the low-pass filter 124b is initialized by this deviation. Turn into. When these operations are expressed by equations, Equation 14 is obtained.

The inputs of the low-pass filters 124a and 124b are set to zero, and the d and q-axis current compensation values i _dcomppull and i _qcomppull are reduced to zero with time.

Limiter 125a is the upper limit value of d-axis current compensation value _{i Dcomppull,} limits using the d-axis current compensation value limit _{i dcomppulllim.} The upper limit value of the limiter 125a is i _dcomppullim and the lower limit value is (−i _dcomppullim ). Further, limiter 125b is an upper limit value of the q-axis current compensation value _{i Qcomppull,} limits using q-axis current compensation value limit _{i qcomppulllim.} The upper limit value of the limiter 125b is i _qcomppullim and the lower limit value is (−i _qcomppllim ).
Here, the d and q-axis current compensation value limit values i _dcomppullim and i _qcomppullim are calculated by a current compensation value limit value calculator 127 described later.

The adder 126a calculates the d-axis current command value i _d ^* by adding the d-axis current command value i _dVector ^* and the d-axis current compensation value i _dcomppull . The adder 126b calculates the q-axis current command value i _q ^* by adding the q-axis current command value i _qVector ^* and the q-axis current compensation value i _qcomppull .
By these arithmetic processes, d and q-axis current command values i _d ^* and i _q ^* at the start of sensorless vector control are the initial values of d and q-axis current command values i _dpull ^* and i _qpull ^* at the time of current pull-in control. As shown in FIG. 4, the _{process proceeds to} the d and q axis current command values i _dVector ^* and i _qVector ^* in the sensorless vector control with time.

By the way, when the d-axis current is large, the amplitude of the expansion induced voltage decreases, and the stability of the magnetic pole position / velocity estimation using the expansion induced voltage decreases. Therefore, to prevent the amplitude of the extension induction voltage decreases, when the d-axis current command value _{I dpull} ^* is larger than the upper limit value _{I DlimEEx} is, d, q-axis current compensation value _{i Dcomppull,} the initial value of _{i Qcomppull} , The calculation is performed by the equation 15 instead of the equation 14.

Next, the current compensation value limit value calculator 127 in FIG. 4 will be described.
When the load fluctuates, the d and q axis current compensation values i _dcomppull and i _qcomppul on the other hand cause torque control errors. Therefore, the d and q-axis current compensation value limit values I _dcomppullim and I _qcomppullim are decreased in accordance with the absolute value of the deviation between the torque command value τ ^* during sensorless vector control and the torque τ _pull during current pull-in control. Specifically, as shown in FIG. 5, the current compensation value limit value reduction coefficient K _idqcomp1 is calculated according to the absolute value of the deviation between the torque command value τ ^* and the torque τ _pull . In FIG. 5, Δτ ^* _thL indicates | τ ^* −τ _pull | when the reduction coefficient K _idqcomp1 starts to decrease from 1.0, and Δτ ^* _thH indicates that when the reduction coefficient K _idqcomp1 becomes zero. | Τ ^* −τ _pull |.

Further, in order to prevent the voltage from being saturated during the medium-high speed operation of the PMSM 80, the d and q-axis current compensation value limit values I _dcomppullim , I are set according to the absolute value of the speed ω _r (set the speed estimated value ω ₁ ). _{Reduce qcomppulllim} . Specifically, as shown in FIG. 6 calculates the reduction coefficient _{K Idqcomplim2} current compensation value limiting value according to the absolute value of the velocity omega _r. In FIG. 6, ω ^* _thL represents | ω _r | when the reduction coefficient K _idqcomplim2 starts to decrease from 1.0, and ω ^* _thH represents | ω when the reduction coefficient K _idqcomplim2 becomes zero. _r |
Then, d and q-axis current compensation value limit values I _dcomppullim , I _qcomppullim are calculated by Expression 16.

11u: u-phase current detector 11w: w-phase current detector 12: input voltage detection circuit 13: PWM circuit 14, 15: coordinate converters 16, 19a, 19b: subtractor 17: speed regulators 18, 41: current command Calculation unit 20: Current regulators 21a, 21b, 23: Changeover switches 22a, 22b: Low-pass filter 24: Electrical angle calculator 31: Extended induced voltage calculator 32, 35: Angle calculator 33: Speed estimator 34: Expanded magnetic flux Calculator 36: Torque calculator 50: Three-phase AC power supply 60: Rectifier circuit 70: Power converter (inverter)
80: Permanent magnet synchronous motor (PMSM)
121: current command calculator 122: coordinate converters 123a, 123b: subtractors 124a, 124b: low-pass filters 125a, 125b: limiters 126a, 126b: adders 127: current compensation value limit value calculators

Claims (5)

The current, terminal voltage, and magnetic flux of a permanent magnet type synchronous motor that is driven by a power converter and does not have a magnetic pole position detector are taken as vectors, and the current and terminal voltage of the motor are decomposed into the components of the vector. In the control device for the permanent magnet type synchronous motor that controls the power converter,
As a control mode of the electric motor,
The first operation mode for controlling the angular velocity of the current command value of the motor to the speed command value, and the rotor magnetic pole position estimated value and the speed estimated value calculated from the current and terminal voltage of the motor, A second operation mode for controlling the angular speed to a speed command value,
When switching from the first operation mode to the second operation mode,
Means for calculating a magnetic pole position estimation error from the electric current of the motor, the terminal voltage, and the angular velocity of the current command value;
Means for calculating an initial value of the second current command value in the second operation mode using the first current command value in the first operation mode and the magnetic pole position estimation error;
Means for calculating a first current command value in the second operation mode from a torque command value of the motor;
Means for initializing a current compensation value using a deviation between an initial value of a second current command value in the second operation mode and a first current command value in the second operation mode;
In the second operation mode,
Means for reducing the current compensation value to zero over time;
Means for adding the first current command value of the second operation mode and the current compensation value to calculate a second current command value;
Means for controlling the current of the motor to the second current command value;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to claim 1,
A control device for a permanent magnet type synchronous motor, comprising means for limiting an upper limit value of a second current command value in the second operation mode. A control device for a permanent magnet type synchronous motor according to claim 1 or 2,
In the second operation mode,
Means for limiting the upper limit of the current compensation value;
Means for calculating the upper limit value of the current compensation value from the deviation between the torque calculation value and the torque command value when switching from the first operation mode to the second operation mode, and the initial value of the current compensation value; ,
A control device for a permanent magnet type synchronous motor. It is a control device of a permanent magnet type synchronous motor given in any 1 paragraph of Claims 1-3,
In the second operation mode,
Means for limiting the upper limit of the current compensation value;
Means for calculating an upper limit value of the current compensation value from the speed estimation value and an initial value of the current compensation value;
A control device for a permanent magnet type synchronous motor. In the control device for the permanent magnet type synchronous motor according to any one of claims 1 to 4,
The control apparatus for a permanent magnet type synchronous motor, wherein the first operation mode is a current drawing control mode, and the second operation mode is a sensorless vector control mode.

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