JPS6176091A - Digitally controlling method of induction motor - Google Patents

Abstract

PURPOSE:To enhance the responsiveness for controlling by digitally controlling an exciting current as a reference, and correcting a counterelectromotive force, thereby smoothing the responsiveness at a transient time. CONSTITUTION:A speed command VCMD and a feedback speed voltage output from a frequency/voltage converter 4 are compared to obtain a torque command TC. An exciting current generator 9 obtains an exciting current I0 from a torque command TC. The secondary current generator 10 obtains the secondary current I2. The primary current amplitude and phase generator 12 obtains the primary current amplitude ¦I1¦ and a phase thetac. Phase current command calculators 15 obtain phase current commands, obtain voltage commands from a current deviation from a feedback current from a current loop, and superpose electromotive forces for correcting the counterelectromotive force calculated on the basis of the current I0 by an electromotive force generator 29 on the voltage command.

Description

Detailed Description of the Invention (Industrial Application Field) The present invention relates to a method for digitally controlling an induction motor, and in particular, a method for digitally controlling an induction motor based on an excitation current. Regarding the method.

(Prior art) Conventionally, there are various speed control methods for induction motors.The representative method is voltage/frequency (V/F).
There are two control methods: a slip frequency control method and a vector X1 method, and the vector control method is broadly classified into a magnetic flux detection type vector control method and a slip frequency vector control method. Recently, there has been a demand for fast-responsive control that takes into account transient phenomena using microprocessors, and vector DI
m has been noted [1]. FIG. 6 is a schematic block diagram of the vector control method, which controls the induction motor 1 by treating pressure and current as vectors.

Fig. 6(a) shows the magnetic flux detection type vector control system: rS6
A block diagram of the slip frequency controlled vector control system is shown in FIGS. These methods are
Conventionally, voltage and current were treated as scalar waves with amplitude and frequency, but rji4ff of three-phase voltage and current
Vector j− with two components formed by ¥ values
In this method, the instantaneous i+ti of each phase is controlled as 1. In contrast, the two-vector control method eliminates the instability and response limitations caused by scalar handling, and enables control equivalent to that of a first-class motor.

(Problems with the prior art) However, the vector control method performs control based on the secondary magnetic flux (the vector control method is based on the secondary magnetic flux as the d-axis and the q-axis orthogonal to it). In such a control method, as is clear from Figs. This was the current situation.

(Objective of the Invention) In order to solve the above-mentioned problems, the present invention performs vector control based on the excitation current, thereby increasing the responsiveness of the control and achieving control equivalent to that of a DC motor. It is an object of the present invention to provide a digital control method for an induction motor, which can achieve this goal and reduce costs.

(Summary of the Invention) The present invention comprises the following steps: obtaining an excitation current from the actual speed and torque command of an induction motor; obtaining a primary flow width and current phase based on the excitation current; obtaining a current command for each phase based on the current amplitude and current phase; obtaining a voltage command based on a current deviation of the feedback current from the current loop for each phase current command; The voltage command is provided with a step of gaining an electromotive voltage for back electromotive voltage correction calculated based on the excitation current.

(Example) An example of the present invention will be described in detail below with reference to the drawings.

The FzZ diagram is a simplified -phase equivalent circuit of an induction motor. In Fig. 2, vl is the terminal voltage, E is the electromotive voltage, I1 is the primary current, Io is the exciting current, and I2 is the load current. 1m indicates inductance, I2 indicates secondary resistance, and S indicates subtrim. Moreover, this can be expressed as a vector diagram as shown in FIG.

From the monovalent circuit for this phase, 14 Il is calculated, so that the - next current of this circuit is 1. Electromotive voltage E1. ) Each of the torques T can be expressed based on the excitation current io. That is, 1+=I・teni2 1+=(I in j M) La
...(1n and 2 ε+ = −jmlrn to −j 27Cf1m
Lo - (2) T1 engineering x-f-x X+vt
2L o21a)o I2 = “1 rn” S w L a”yjx 2pc
5J-1'rn'Lo"
River (3nz) However, P here represents the number of opposite poles.

Also, 1 phase 0 is θ=co-1! (4) Therefore, the relationship with the speed of the induction motor is expressed as shown in FIG.

(1) First, as is clear from Figure 4 (IL), 1#
, 1j! Current I. is constant below the base speed Nb, and the excitation TL is set so that the electromotive force E1 is constant below the base speed Nb.
Flow ■. is decreased as the speed N increases. Expressing this in the form of an equation, N≦Nbl nee j < 1. t[. = (constant) ... (5) N≧Nbl: J+'iτ
tjr. = K-Nb/N
---<6.

(2) Next, looking at the electromotive force E, as is clear from FIG. 4(b), the electromotive force E.

is increased up to the base speed Nb, and once the base speed Nb is reached, the value is controlled to be constant. For this purpose, the excitation current I0 is controlled as shown in FIG. 4(b).

(3) Next, let's look at the superfluid number.

As is clear from FIG. 4(C), the shear frequency fs is constant up to the base speed IIi, and changes in proportion to the output above the base speed Nb.

(4) Next, let's look at torque.

It becomes.

These characteristics are determined by the excitation current I. It can be considered based on. That is, as is clear from FIG. 4(b), the electromotive force EI is E+ = k + fIo
・=(7) Here, kl is a zero-order first-order TF which is an electromotive force constant, and t&amplitude II+1 is the above-mentioned (1
) As is clear from Eq.

lI+ l = 1Iol xl + +(""'
)'z=HDlxF]7Tenka...(9).

Next, the -order current phase θ is as is clear from the above equation (4).

θ Bow 4/L-' ``Giant = 54-'' 1v3 above Tako Eni... (9) Next, the torque T is as is clear from the above equation (3).

T = "Lx 2X S fern" I n""
k+5fIo”
It can be obtained as =·(lo).

Therefore, these characteristics can be shown as shown in FIG.

Therefore, the digital 17 control force V of the induction motor according to the present invention is shown as the eleventh threshold. In the diagram, l is a three-phase induction motor, 2 is a pulse generator, a nelator, 3 is a quadruple circuit, and 4
1 is a frequency/voltage converter, 5 is an analog/digital converter, 6 is an arithmetic circuit, 7 is a proportional-integral circuit, 8 is a clamp circuit, 9 is an exciting current generator, 10 is a secondary current generator, 11
12 is a primary current amplitude and phase generation section; 13 is a counter; 14 is an operation X ta+ path; 15 is a C calculation section for each phase current command;
9 to 21 are blocks having effects on gain, 22 to 2
4 is an arithmetic circuit, 25 is a pulse width modulation (PWM) circuit, 2
6.27 is an analog/digital converter, 28 is a converter
II! The circuit 29 is an electromotive voltage generating section.

Next, the operation of this digital control system for the induction motor will be explained.

The speed command VCMD and the feedback speed voltage output from the frequency/voltage converter 4 are compared in the ri4rL circuit 6, and 1
It is output as error ER. The error ER is determined by the proportional integral circuit 7. Is the torque command TC transmitted through the clamp circuit 8? ')
This torque command TC is sent to the exciting current generator 9. The signal is inputted to a secondary current generating section 10 and a sub-frequency dissipation generating section 11, respectively. In the excitation current generating section 9, the feedback actual speed TSA and torque command TC output from the analog/digital converter 5 are input, and the excitation current I is determined from the relationship shown in FIG. 4 (&). seek.

Next, in the secondary current generating section 10, the torque command TC
The secondary current I2 is determined from the relationship proportional to the secondary current.The heli frequency generator 11 calculates the heli frequency from the relationship shown in FIG. 4(c) from the feedback actual speed TSA and the torque command TC. Obtain fs Next, (1 turned excitation current I
. , find the primary current amplitude @lII1 from the secondary current I2. However, as shown in equation (1) above, the exciting current I
0 as a basis, one phase θC that can obtain the -order electric bran width 1■II I can be obtained from equation (4).

On the other hand, the subtraction frequency ωS output from the subtraction frequency generation section 11 is input to the counter 13.

Further, the speed frequency ωr outputted from the quadrupling circuit 3 is also manually input to the counter 13. And the counter 13 is 11
One person is added and the excitation phase ω of the rotating magnetic field.

Output. Its excitation phase ω. ii in the arithmetic circuit 14
It is added to the i-th phase Oc and outputs two phases θ.

The phase O is manually input to each phase current command lyl calculation unit 15 together with the negative current amplitude l111.

Therefore, in each phase current command computation section 15, J
, Ii, each phase current command is determined. In other words.

seek. In addition, here! E string! +1n table memory is increased to 11.

Next, each phase current command given 1'I is converted into a current deviation f' by the feedback current from the current feedback loop and the calculation circuits 16 to 18.

Next, the electric bias XY is multiplied by the gain Kl to obtain a voltage command.

Next, an electromotive voltage for back electromotive force correction is superimposed on this voltage command. To explain this point in detail, this electromotive force is:

The electromotive voltage EI is clear from the above-mentioned FIG. 4(b), and is expressed by the above-mentioned formula (2), that is, E+=-j2JffrnI.

More sought after. For this purpose, the excitation main iI is output from the excitation current generating section 9. and excitation phase ω. is input to the electromotive voltage generation section 29. It should be noted that it goes without saying that a table memory for sine sin should be used here as well. Therefore, the above equation (12
), that is, when the electromotive force for back electromotive force correction iE is superimposed,
What is the voltage value manually input to the PWM circuit 25 of each phase?

Vut=-EHuL0ktx(IuLc-[aa)Vv
a=ε1v + IQ × (1νc−TvF)Vwt−
= K+w+ KK Z (Ice - rWF).

At 00, ・IAI=・VFI■- is the actual current.

Although the uninvented invention has been explained by one example, the present invention is not limited to this example, and various modifications can be made in accordance with the gist of the invention, and these are excluded from the scope of uninvented IJ+. It's not something you do.

(Effects of the Invention) According to the present invention, since the excitation is based on the current and the induction motor is digitally controlled in a vector manner, there is no need to detect the secondary magnetic flux as in the past. In addition, in order to generate an electromotive force that corrects the back electromotive force that appears as a transient phenomenon, it is possible to smooth the response in the event of a transient phenomenon. Therefore, it is possible to improve the responsiveness of the control 1, to enable control equivalent to that of a DC motor, and to reduce costs.

Figure 4 is a block diagram of a digital control system for an induction motor to realize the present invention, and Figure 2 is a block diagram of a digital control system for an induction motor to realize the present invention.
An equivalent circuit diagram of the phase component, Figure 3 is a hector diagram based on the second factor, Figure 4 is a characteristic diagram showing the relationship between actual speed and various quantities,
FIG. 5 is a characteristic diagram showing the relationship between the overburden and various quantities, and FIG. 6 is a block diagram of a conventional vector control system.

9... Excitation current generation section, 11... Slip frequency generation section, 12... -order current amplitude and phase generation section, 15...
-Each phase current command calculation section, 29... electromotive force generation section.

Patent applicant Fanuc Co., Ltd. Agent Patent attorney Minoru Tsuji (1 other person) IO Figure 4 ((1) Figure 4 (C) Figure 4 (b) Figure 4 c) Figure 5 (0) ( C) (b) 手 wa ■ Ne city official calligraphy (spontaneous) 1. Display of the case 1982 Patent Application No. 179267 2 Name of the invention Digital control method for induction motor 3, person making the amendment Relationship to the case Patent applicant name FANUC Co., Ltd. Representative Seiemon Inaba 4, agent address 3-146 Kanda Ogawacho, Chiyoda-ku, Tokyo 101,
The object of correction is No. 1. NbL's 11j1 is 1" to N. Name of the invention, -A41, motive, Notary 1ljl U4 Method 2
, the scope of the claims includes a step of obtaining an IL current from the actual speed of the induction motor and a direct torque command, and a step of obtaining an IL current from the torque command of )1°:'6-.゛1
Step 2: Obtaining the primary current amplitude and current phase based on the excitation current, and obtaining each phase current command based on the current amplitude and current phase from IY torque. step, obtaining a voltage command based on the current deviation between the current command for each phase and the feedback current from the current loop; There is a growing demand for fast control of hector control Nbi: - II'' is 1'''.
11. A step of superimposing an electromotive voltage for back electromotive force correction +F is provided: , s4
Digital control method for electric motors.

3. Detailed description of the invention (industrial application?) Part 1 of Section 7 relates to a digital control method for a conductive fJ+ machine, in particular, the induction N and motor digital control method based on the excitation current. This invention relates to a digital control method for a J iFJ electric motor.

Conventionally, there are various speed control methods for induction 'IILI machines.The most typical method is voltage/frequency (
There are V/F) control force type, vertical frequency control type, and vector control force type, and the vector control type is broadly classified into magnetic flux detection type vector control l type and vertical frequency vector control type force type. Responsiveness that also takes into account excessive development using the f&N, Yaikuroprosensor has been attracting attention. FIG. 3 is a schematic block diagram of a vector control system, which controls the voltage and current of an induction motor by treating them as vectors. FIG. 6 (IL) shows a magnetic flux detection type vector control system, and FIG. 6 (,b) shows a block diagram of the Veri frequency control type vector f, II i11100. These methods conventionally convert voltage and current to amplitude and Lua wavenumber? ,? In this method, the instantaneous value of each phase is treated as a vector ψ having two components formed by the three-phase voltage and the instantaneous value of TLfQ, whereas the instantaneous value of TLfQ is treated as a scalar quantity. According to the vector system 01 method, there are no limits to instability and response due to scalar treatment, and control such as that of a DC motor becomes possible.

(Problem to be solved by the invention) However, the vector control method performs control based on the secondary magnetic flux (vector control i1i method based on the secondary magnetic flux as the d-axis and the q-axis perpendicular to it). Therefore, in such a control system, Il
At present, the secondary Ia end must be detected with certainty, and the system has a problem with responsiveness for control and has no choice but to increase cost.

The present invention has the following features: 1. In order to solve the problems described above, by performing pectonodonic control based on the excitation current, the responsiveness for the control i1σ can be improved, and the IIr function can be improved from the DC motor and the equivalent a The purpose of the present invention is to provide a digital control method for an induction motor that reduces costs.

(Means and effects for solving the problem -!) The present invention provides a method for converting π between the actual speed and command speed of a single induction motor into a force 1i.
+l: Force the actual speed to follow the finger.
In a digital control method for an electric motor, the actual speed N is calculated from the actual speed of the induction motor and a torque command with a difference of 11η.
(For 111F below the base velocity Nb, the excitation is -. When the base velocity is above Nb, the excitation is reduced in proportion to Nb/N.
j! a step of obtaining a primary current, which is a current flowing through the load '1', from the torque command of the difference, and determining a primary current amplitude and 'it' based on the excitation current and the
a step of obtaining a slip frequency from the actual speed and torque command; a step of obtaining an excitation phase from the slip frequency and the actual speed; A step of obtaining an Ichikawa command based on the current deviation between the current command upper type of each phase and the return current from the loop, and determining that the actual speed N is equal to or less than the base speed Nb based on the excitation III 1 (back electromotive force correction which is constant when it is equal to or higher than Nb) 1-, and a step of calculating the electromotive voltage for back electromotive force correction which is proportional to N at the time of It consists of a step of superimposing an electromotive force.

(Implementation M) Hereinafter, with reference to the drawings regarding an uninvented embodiment, f
F Explain in detail.

What is the second figure? This is a simplified equivalent circuit of the A-phase induction motor. In the fJS2 diagram, v+ is the terminal type I″F
BI is the back electromotive voltage, 11 is the primary current amplitude o is the exciting current,
I2 is the load current, gLrQ is the inductance, rz is the secondary II (resistance), and S is the subtraction.

Furthermore, this can be expressed as a vector diagram as shown in FIG.

As is clear from the equivalent circuit of this - phase component, the primary current amplitude of this circuit, the back electromotive force E1. The torque T is the excitation '1 and the second flow I, respectively. ), (can be expressed as the foundation, i.e.

r + sat. +t2 L + == (1+ jj1v61)i 6
Mountain (1nrλ white 1 = -j*frn IO = -j27cfirn
ta,” <2-) ding=’-X-S-g
x, n2 L. zQσ rz =-'F-1m'5tado" and λ = i-x2, 5ftrn'r6" r□...(3) However, here P represents the opposite.

Also, phase 0 is.

(4) Therefore, the relationship between the induction motor and the melon is expressed as shown in FIG.

(1) First, as is clear from Section 41; 4(a),
The exciting radio wave I ++ is (constant below the bottom speed N b,
At the bottoming out speed Nb or higher, the excitation la current is set so that the back emf 7L pressure E is constant! . is decreased as the speed N increases. Expressing this in a formula, for N16M, (1.-k (constant)
...(tanN≧Nb(=%vgo':l
Ie=K x Nb/N”・(1;
It becomes J.

(2) Next, let's look at the back electromotive force E1.

As is clear from FIG. 4(b), the back electromotive force E.

increases up to the base speed Nb, and once the base speed Nb is reached, the value is controlled to be constant. For that purpose, the excitation current■. is controlled as shown in FIG. 4 (IL).

(3) Next, let's look at the shear frequency.

As is clear from FIG. 4(C), the stagnation frequency fs is constant up to the bottom speed, and changes in proportion to the output below the base speed Nb.

(4) Next, let's look at torque.

As is clear from FIG. 4(d), the velocity is constant up to the base velocity Nb, and decreases above the base velocity Nb.

And these qualities are excitation current■. (Based on J5), as is clear from FIG. 4(b), the back electromotive force E is greater than or equal to the base velocity Nb.

E+=l++j ding□
...(7) (where 1 is the centripetal force constant) The zero-order primary current amplitude II+I is as clear from the above equation (1).

1 Koji-II・+Xl possible fan kaiseki [y =11.1x Month I Lower F...(It's an illusion.

Next, the -order current phase θ is as is clear from the above equation (4).

θ−−−1−Ryoichi = −1′ 2χfr, l− Shigeru t thread 1 ph 1 Midori 3 engineering
...(9) Next, the torque T is, as is clear from equation (3) in +tij, T= 4 x 2ycSftm2L♂ = k4s)Io"
・=<+ (k2.ks,
, + is 1 number).

Therefore, the quality properties shown by these equations (8), (9), and (10) can be expressed as shown in FIGS. 5(a), (b), and (c).

Therefore, a digital control method for an induction motor according to the present invention will be explained based on a block diagram of a digital control method NO for an induction motor shown in FIG. In the figure, l is a three-phase induction motor, 2 is a pulse generator, 3 is a quadruple circuit, and 4
is a frequency/voltage converter, 5 is an analog/digital converter, 6 is an arithmetic circuit, 7 is a proportional-integral circuit, 8 is a clamp circuit, 91', - excitation current generation section, IO is a secondary current generation section, 11 is Slip frequency ′l! 12 is a primary current amplitude and phase generator, 13 is a counter, and 14 is an arithmetic circuit.

15 is a current command calculation unit for each phase, 16 to 18 are calculation circuits,
19 to 21 are blocks having a gain; 22 to 24 are arithmetic circuits; 25 are pulse width modulation (PWM) circuits;
6.27 is an analog/digital converter, 28 is riI
'fi, circuit; 29 is a correction electromotive voltage generating section;

Next, the digital control operation of this induction motor will be explained.

The speed command VCMD and the feedback speed voltage output from the frequency/voltage converter 4 are compared in the arithmetic circuit 6, and: l I
j, output as E R. The error ER is determined by the proportional integral circuit 7. It is supplied to the clamp circuit 8 and a torque command TC is obtained. This torque command TC is sent to the exciting radio wave generator 9. The signal is input to the secondary current generating section 10 and the edge frequency generating section 11, respectively. In the excitation current generator 9, the actual return speed TSA output from the analog/digital converter 5,
) Manually input the torque command TC and obtain the excitation current (■) using the functional relationship shown in FIG. 4(a).

Next, in the secondary current generating section 10, the torque command Tc
From the relationship that C is proportional to the secondary current, the secondary electric current L2 is calculated. Based on the function relationship that is reduced, the number of cycles fs is obtained.

Next, the obtained excitation current ■. , the -order current amplitude 11 and phase θC are determined from the secondary current I2. Here, as shown in equation (1) above, the excitation current ■. On the basis of tt, negative order current oscillation II+1 can be generated by 7jl. Further, the phase θC can be determined from equation (4).

On the other hand, the slip frequency ωS output from the slip frequency generator 11 is input to the counter 13.

Further, the speed frequency ωr output from the quadrupling circuit 3 is also input to the counter 13. Then, in the counter 13, both are added and the excitation phase ω of the rotating magnetic field is obtained.

Output. Its excitation phase ω. is fi in the arithmetic circuit 14
It is added to the phase OC of ii and outputs one phase θ.

The phase 0 is manually inputted to each phase current command calculating section 15 together with the -order current +tM@JII+l.

Therefore, in each phase current command calculation section 15, the following formula is used.
, tl, each phase current command is determined. In other words, find . Note that here we will be able to use the sine wave (SIα) table memory.

Next, current deviations are obtained from the current commands for each phase that have been deviated by the feedback current from the feedback loop and the calculation circuits 16 to 18.

Next, the voltage command is obtained by multiplying the current deviation by the gain.

Next, an electromotive force for back electromotive force correction is added to this k pressure command. I will explain this point in detail below.
The back electromotive force E is 6N as shown in Figure 4 (b).
．． m I.

Yo℃ is found. For this purpose, the excitation mt current I is output from the excitation ra current generating section 9. and of the excitation phase. is input to the corrected pressure/U pressure generation section 29. It goes without saying that the table memory for the sine wave (sin) is set to one village and five in this case as well.

Therefore, when the Equation (12) in 1, that is, the electromotive voltage for back electromotive voltage correction is superimposed, the voltage value input to the PWM circuit 25 of each phase is as follows.

V (1, ε curvature ε+u 10 K K K (IIJ+c
-Lap) Vve = E w f K engineering < (Iv
c-Lvt')+1/Wl: -Elw +K X people (
Iw−−τWF).

Here, IU, LVF, and Iwp are actual currents.

Although the present invention has been described with reference to one embodiment, the present invention is not limited to this embodiment, and may be modified in various ways according to the spirit of the present invention, and these are excluded from the scope of the invention. It's not something you do.

(Effects of the Invention) According to the present invention, since the induction motor is digitally controlled in a vector I-le manner based on the excitation current, there is no need to perform secondary & i-east detection as in the past. ,
In addition, in order to generate an electromotive voltage that compensates for the back electromotive force that appears as a transient phenomenon, it is possible to smoothen the response during overheating. Therefore: fjl ilj
It is possible to increase the response of 15 times and make the control equivalent to that of the +I(/Q1!!lJ machine 11 functions) and reduce costs.

4. Brief explanation of the drawings 1: The tSi diagram is a professional diagram of the digital aJ1 iJ one-type induction motor for realizing the present invention, and Figure 2 is the equivalent circuit for one phase of the induction motor It4. Fig. 3 is a hector diagram based on Fig. 2, Fig. 41y4 is a characteristic diagram showing the relationship between actual speed and various quantities, and Fig. 5 is a special P1 diagram showing the IA relationship between the swell and various quantities. FIG. 6 shows a conventional vector control system.

9... Excitation 'i17 current generation generation section... Naheli wave number generation section, 12...-order current amplitude and phase generation section, 1
5...Each phase current command section 1'X section, 29...Elevating II
'Generation part.

Claims (1)

[Claims] Obtaining an excitation current from the actual speed and torque command of the induction motor; Obtaining a primary current amplitude and current phase based on the excitation current; Obtaining each phase current command based on the primary current amplitude and current phase. a step of obtaining a voltage command based on a current deviation between the current command of each phase and a feedback current from the current loop; 1. A digital control method for an induction motor, the method comprising the step of increasing the electromotive force of the motor.