JP3209853B2 - Control device for synchronous motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an AC servomotor, particularly a synchronous motor known as a reluctance motor.

[0002]

2. Description of the Related Art As a conventional improved reluctance motor control device, there is a control device for detecting a rotor position and a rotor speed and performing precise speed control as shown in FIG.

A conventional example shown in FIG. 11 will be described.

The synchronous motor has a salient pole type rotor 6, to which a position detector 5 is mechanically coupled, and outputs a position signal DS.

The rotor rotation position AR is the position of the magnetic pole of the rotor 6 with respect to the U-phase winding position as shown in FIG.

The speed command SI and the speed signal SD detected by the speed detecting means 3 are compared by the adder 1 to obtain a speed deviation ES, and the torque control means 2 performs a compensation operation such as proportional, integral or differential. Obtain a torque command T.

The field current command circuit 24 receives the rotor rotational position AR and the speed signal SD detected by the rotor position detecting means 4 as inputs, and outputs the following field current commands SFIU and SFI.
V, SFIW are created.

SFIU = SF · SIN (AR) SFIV = SF · SIN (AR + 120 °) SFIW = SF · SIN (AR + 240 °) where SF is the amplitude of the field current and depends on the speed signal SD. Below the rotation speed NB, at a certain constant value,
Above the base rotation speed NB, SF increases as the speed signal SD increases so that the value obtained by multiplying NB and SF becomes a constant value.
Is decreasing.

The armature current command circuit 21 outputs a torque command T
The following armature current commands SAIU, SAIV, and SAIW are output with the input and the rotor rotation position AR.

SAIU = T ・ SIN (AR + 90 ゜) SAIV = T ・ SIN (AR + 90 ゜ +120 ゜) SAIW = T ・ SIN (AR + 90 ゜ +240 ゜) Each current command SIU, SIV, SIW is expressed by the following equation.
Each armature current command and each field current command are added to each adder 25,
26 and 27.

SIU = SAIU + SFIU SIV = SAIV + SFIV SIW = SAIW + SFIW The angle representing the direction of the combined vector of these phase currents is angle A with respect to the direction of the rotor magnetic pole in FIG.

The current control circuit 7 includes current commands SIU, SI
V, SIW are amplified as inputs, and three-phase currents IU, IV, IW are supplied to the windings U, V, W of the three-phase motor.

As a result of such control, the rotor 6
The control of the field magnetic flux and the armature current component that generates the torque are appropriately controlled according to the direction of the magnetic poles of the rotor 6.
Can generate any clockwise or counterclockwise torque,
Speed control of the motor is performed.

[0014]

The control device shown in FIG. 11 has the following problems since the motor characteristic has an electromagnetic non-linear element or the like in spite of its simple logic. Was.

Since the motor magnetic circuit is non-linear, a field magnetic flux proportional to the field current cannot be obtained and an expected output torque cannot be obtained.

When a large armature current component is applied to obtain a large output torque, the field magnetic flux changes due to the armature current component, and the expected torque cannot be obtained.

There is a voltage drop due to the leakage inductance of the motor winding, and the expected torque cannot be obtained at high speed rotation.

To control more strictly, each compensation control or the like is required, and as a result, the content of the control becomes complicated and the cost increases.

[0019]

According to the present invention, in order to solve the above-mentioned problems, the relative phase angle between the rotor and the motor current is adjusted so that the magnetic flux in the motor becomes appropriate, that is, the excitation current component of the motor current and the relative phase angle. Relative phase angle creating means for creating a relative phase angle so that an armature current component becomes appropriate, and current amplitude command means for creating an appropriate motor current amplitude by compensating for non-linearity between the motor current amplitude and generated torque. And, was prepared.

[0020]

The rotation speed N of the electric motor is determined by the relative phase angle creation means.
And the torque command, the relative phase angle A can be obtained by a relatively simple algorithm that reduces the relative phase angle A according to the rotational speed in a range where a certain function AS (T) and the torque command value T are large.

The current amplitude command means sets the current amplitude command to a substantially constant value in a range where the torque command is small, and increases the torque command value T according to a certain function IO (T) as the torque command value T increases. An optimum current amplitude is obtained by a relatively simple algorithm that further increases the current command as the value becomes higher.

[0022]

FIG. 1 shows an embodiment of the present invention. The description of the same components as in FIG. 11 showing the conventional example is omitted.

In FIG. 1, in order to determine the relative phase angle A between the magnetic pole of the rotor 6 and the vector direction of the motor current to an optimum value, a relative phase angle creating means 9 is provided. The signal SD is input. The obtained relative angle A is added to the rotor rotational position AR in the adder 15 and sent to the current command means 10.

The torque command T and the speed signal SD
Is further sent to the current amplitude command means 8 and its output IOS
Is sent to the current command means 10. The current commands SIU, SIV, SIW for each phase of the current command means 10 are converted by the current control circuit 7 into three-phase currents IU, IV, IW to the windings U, V, W, respectively.

FIG. 2 shows the characteristics of the relative phase angle creating means 9. The characteristics depend on the torque command T and the rotation speed N of the electric motor.

The characteristic shown in FIG. 2 is a characteristic according to the function AS (T) when the rotational speed is equal to or lower than the base rotational speed NB, and is also a characteristic according to the function AS (T) when the torque command T satisfies −T1 ≦ T ≦ + T1. It is.

When the rotation speed is equal to or higher than the base rotation speed NB and the torque command value T exceeds a predetermined value T1, the torque command T
Together with the function AS (T), and converges to the minimum value AMN of the relative phase angle A. When the torque command value T decreases below the predetermined value -T1, the relative phase angle A increases with the torque command T from the function AS (T), and the relative phase angle A
Converges to the maximum value AMX. An example of a specific algorithm for realizing this characteristic is shown in the flowchart of FIG.

FIG. 4 shows a representative characteristic ADC (T) of a portion having a rotation speed dependency and a torque dependency in the characteristics of FIG.
It is.

In the flowchart of FIG. 6, first, a characteristic AX excluding the characteristic AS (T) of FIG. 2 is obtained.

AX = ADC (| T |)-(N3- | N |) .KA (1) However, if AX is negative according to this formula, it is replaced with zero.

N3 is the maximum number of revolutions.

KA is a coefficient representing the rotation speed dependency.

Next, the direction of the torque command value T is determined and A
The characteristic of the relative phase angle A in FIG. 2 is obtained by adding or subtracting the variable AX to S (T).

However, if the obtained relative phase angle A does not fall within the range of the maximum value AMX and the minimum value AMN of the relative phase angle, it is replaced with the maximum value AMX or the minimum value ANM.

As described above, the characteristics shown in FIG. 2 can be obtained by a relatively easy algorithm.

Although the variable AX is defined in the above equation (1), this is described as an example of the method, and another similar function or a more strict function can be used.

FIG. 3 shows an example of the characteristics of the current amplitude command means 8. The characteristics depend on the torque command T and the rotation speed N of the electric motor.

The characteristic shown in FIG. 3 is a characteristic according to the function IO (T) when the rotation speed is equal to or lower than the base rotation speed NB, and is also a characteristic according to the function I0 (T) when the torque command T is -T1 ≦ T ≦ + T1. It is.

When the rotation speed is equal to or higher than the base rotation speed NB and the torque command value T exceeds a predetermined value T1, the torque command T
Together with the function IO (T).

An example of a specific algorithm for realizing this characteristic is shown in the flowchart of FIG.

FIG. 5 shows a representative characteristic IDC (T) of a portion having a rotational speed dependence and a torque dependence in the characteristic of FIG.
It is.

In the flowchart of FIG. 7, first, a characteristic IOX is obtained except for the characteristic IO (T) of FIG.

IOX = IDC (| T |)-(N3- | N |) .KB (2) However, if IOX is negative according to this formula, it is replaced with zero.

KB is a coefficient representing the rotation speed dependency.

Next, the characteristic of the current amplitude command A shown in FIG. 3 can be obtained by adding the IOX to the function IO (T).

As described above, the characteristics shown in FIG. 3 can be obtained by a relatively easy algorithm.

Although the variable IOX is defined in the above equation (2), this is described as an example of the method, and another similar function or a more strict function can be used.

The current command means 10 outputs a current command SI for each phase.
U, SIV, SIW are obtained according to the following equation.

SIU = IOS · SIN (AR-A + 90 °) SIV = IOS · SIN (AR−A + 90 ° −120 ° ) SIW = IOS · SIN (AR−A + 90 ° −240 ° ) The magnetomotive force component acting on the field magnetic circuit including the value is proportional to IOS · SIN (A), and the current component flowing through the winding facing the rotor, that is, the armature current component is a value proportional to IOS · COS (A). Becomes

Since the output torque of the motor is proportional to the product of the field magnetic flux and the armature current orthogonal to the field magnetic flux, if the current amplitude value IOS and the relative phase angle A can be arbitrarily controlled, the output torque can be arbitrarily determined. As a result, high-precision speed control and position control can be performed.

However, not all types of synchronous motors can be operated efficiently by the above control.

According to the present invention, a more efficient operation can be realized in a synchronous motor capable of stably generating a field magnetic flux by the magnetic poles of the rotor and flowing a current interlinking the field magnetic flux.

An example of a synchronous motor particularly suitable for the present invention will be described.

FIG. 8 is an example of a sectional view of a rotor of a synchronous motor. Reference numeral 30 denotes a shaft of the rotor, 31 denotes an electromagnetic steel plate which is laminated, and 32 denotes a magnetic insulating portion formed by stamping out the electromagnetic steel plate.

Each magnetic path is connected between the outer peripheral portion of the rotor and the intermediate portion of each magnetic pole. The main purpose is to facilitate the handling and assembly of the magnetic steel sheet 31 and to firmly fix the magnetic steel sheet 31 to the shaft 30 as a rotor. It is electromagnetically inevitable.

In the rotor having such a structure, the magnetic reluctance to the adjacent magnetic poles is small, and the field magnetic flux can be easily excited. Since the magnetic reluctance viewed from both ends in the rotation direction of each magnetic pole is large, the rotor faces each magnetic pole. The structure is such that the disturbance of the field magnetic flux of each magnetic pole due to the current flowing through the stator is small. As a result, the synchronous motor can be operated more efficiently.

When the outer peripheral portion of the rotor is connected, there is an effect of reducing the torque ripple caused by the slot and an effect of reducing the magnetic resistance of the field.

FIG. 9 is another example of a sectional view of a rotor of a synchronous motor. Numeral 30 denotes a shaft of the rotor, numeral 33 denotes an electromagnetic steel plate which is laminated, and numeral 34 denotes a magnetic insulating portion formed by stamping out the electromagnetic steel plate.

Each magnetic path is connected between the outer peripheral portion of the rotor and the intermediate portion of each magnetic pole, for the same purpose and effect as in the example of FIG.

The example of FIG. 9 is different from FIG. 8 in that the space between the magnetic poles is more effectively used as magnetic poles, and that the change in the rotation direction of the rotor magnetic resistance is relatively smooth.

The current control circuit 7 has, for example, a configuration as shown in FIG. 10, and includes an inverter 54 composed of a power transistor and current detectors 55, 56, 57 for detecting the currents IU, IV, IW of each phase of the motor. Current command value SI for each phase
Current control circuits 51, 52, and 53 that feed back the current detection values of the respective phases to U, SIV, and SIW and supply drive signals to the respective power transistors of the inverter 54 from the respective difference signals.

By performing such control, specifically, the exciting current component IF of the motor becomes substantially constant below the base rotation speed NB, and above the base rotation speed NB, the motor back-induced voltage reduces the power supply voltage of the control device. Can not be exceeded.

The increase in the current amplitude value at the base rotational speed NB or more shown in FIG. 3 compensates for the decrease in the output torque due to the decrease in the field magnetic flux at the high speed rotation.

A method for realizing such motor control by a microprocessor is a similar function as the method for realizing the AS (T) in FIG. 2, the IO (T) in FIG. 3, and the functions in FIGS. Is stored in a mathematical formula, and calculated each time,
There is a method of storing a function pattern on a memory and reading out the value of ADC (N) corresponding to the rotation speed N each time.

Of course, the implementation of the algorithms shown in FIGS. 6 and 7 can be easily realized by a microprocessor.

In the control shown in FIG. 1 which is an embodiment of the present invention, not only excellent speed control is basically realized, but also various compensation controls are controlled because functions and algorithms having high flexibility are realized. The algorithm can be incorporated with little change.

For example, as a compensation example, the compensation control that the field current component and the actual field magnetic flux are non-linear and have a saturation characteristic is performed by changing the slope of the function AS (T) shown in FIG. The compensation control for the so-called armature reaction, which is the influence of the armature current component on the field magnetic flux, can be made by changing the compensation so that the function AS (T) of the function AS (T) increases as the amplitude of the torque command increases. The inclination can be made gradual, and the maximum relative phase angle AMX is reduced and the minimum relative phase angle AM is set for the voltage shortage of the current control circuit due to the voltage drop due to the leakage inductance of the motor winding.
This can be dealt with by increasing N.

As described above, in the control device of the present invention, the AS (T) in FIG. 2 is changed, and the IO (T) in FIG. 4. Various compensation controls can be realized relatively easily by changing the functions in FIGS. 5 and 5 and slightly changing the algorithm for calculating the control variables in FIGS.

Therefore, according to the present invention, even if various compensation controls are performed, an increase in the load on the control CPU due to the addition of a new algorithm and an increase in cost due to the addition of hardware do not occur.

In the embodiment of the present invention, the relative phase angle A
Has been introduced from zero to 180 °, the characteristics of the motor to which the present invention is applied are that the relative phase angle A is zero to 180 °.
The characteristics of ゜ and 180 ° to 360 ° are the same,
The same control characteristics can be obtained even if the control is performed by shifting the control phase angle by 180 °.

Although the present invention has been described with respect to an example of a method in which the control of the voltage and current of the motor is directly handled by the three-phase AC amount, the so-called three-phase two-phase conversion is performed, and the notation and control are performed using dq coordinates. Are equivalent and are included in the present invention.

The present invention can be applied to a polyphase alternating current other than a three-phase alternating current by modifying the present invention.

The speed controller, the position detector, and the like can be variously modified, and are included in the present invention.

[0074]

According to the control device of the present invention, highly accurate speed control from low-speed rotation to high-speed rotation and from low-torque output to high-output torque can be controlled by a relatively simple control algorithm.

More specifically, the motor magnetic circuit has a non-linearity,
As a method of compensating for the influence of the change in the field magnetic flux due to the armature current component, the voltage drop due to the leakage inductance of the motor winding, etc., the characteristics shown in FIGS. 2, 3, 4 and 5 are corrected. 6. It can be realized without changing the basic control mode by slightly changing the control variable calculation algorithm in FIG.

Accordingly, high-precision speed control can be realized, and at the same time, the configuration of the control device can be simplified, so that the cost can be reduced.

[Brief description of the drawings]

FIG. 1 is a diagram showing an embodiment of a reluctance motor control device according to the present invention.

FIG. 2 is a characteristic diagram of a torque command T and a relative phase angle A in the embodiment.

FIG. 3 shows a torque command T and a current amplitude command I in the embodiment.
It is a characteristic view with OS.

FIG. 4 shows a torque command T and a variable ADC in the embodiment.
It is a characteristic diagram with (T).

FIG. 5 shows a torque command T and a variable IO (T) in the embodiment.
FIG.

FIG. 6 is a flowchart illustrating an algorithm for obtaining a relative phase angle A in the embodiment.

FIG. 7 is a flowchart illustrating an algorithm for obtaining a current amplitude command IOS in the embodiment.

FIG. 8 is a sectional view of a rotor of a synchronous motor suitable for the present invention.

FIG. 9 is a rotor sectional view of another synchronous motor suitable for the present invention.

FIG. 10 is a diagram showing a specific example of a current control circuit 7;

FIG. 11 is a diagram showing an example of the related art.

[Explanation of symbols]

 2 Torque control means 3 Speed detection means 4 Rotor position detection means 6 Rotor 7 Current control circuit 8 Current amplitude command means 9 Relative phase angle creation means 10 Current command means

Claims (3)

(57) [Claims] 1. A control device for a synchronous reluctance motor that obtains a rotational force by utilizing a fact that a magnetic resistance varies depending on a rotational position of a rotor of an electric motor, wherein: a position detecting means for detecting a rotational position and a speed of the rotor; A torque control means that obtains a speed deviation from the speed command of the motor and the detected speed of the rotor, compensates the speed deviation for proportionality, integration, and differentiation, and outputs a torque command, and a direction in which one magnetic pole of the rotor faces. Rotor position
Three congruent but the motor current flowing to the three-phase alternating current winding of the stator
When the torque command value T is zero, the value of the relative phase angle A, which is an angle formed in the rotation direction with respect to the direction of the resultant vector , is 9 electrical degrees.
0 °, and gradually decreases as the torque command value increases from zero. When the torque command value approaches the maximum value, the relative phase angle A also gradually approaches zero degree, and similarly, the torque command value changes from zero to a negative value. The relative phase angle A gradually becomes 180 ° as it decreases.
A relative phase angle creating means for assuming a value of a predetermined function AS (T) asymptotically as follows: a current amplitude command value IOS of the electric motor is changed to a substantially constant value or a torque command T when the torque command value T is from zero to a predetermined value T1. Current amplitude command means which becomes a predetermined function IO (T) gradually increasing with the torque command value when the torque command value is T1 or more, and a relative phase angle A and a current amplitude command. Current command means for inputting a value IOS to create a current command value for each phase of the motor; and a current control circuit for supplying a current to each phase winding of the motor in accordance with the current command value for each phase of the motor. Control device for synchronous reluctance motor. 2. The method according to claim 1, wherein the relative phase angle A is set so that the motor terminal voltage becomes equal to or less than a controllable voltage value when the rotation speed is equal to or higher than a predetermined rotation speed and equal to or higher than a predetermined torque. Accordingly, when the torque command value is positive, the torque command value decreases from the value of the predetermined function AS (T) as the absolute value of the motor speed increases and the torque command value increases. Is the motor speed
2. The control device for a synchronous reluctance motor according to claim 1, wherein the value of the predetermined function AS (T) is increased with an increase in the absolute value and a decrease in the torque command value. 3. The current amplitude command value IOS is set to a value equal to the motor speed so that the effective magnetic flux of the motor decreases at a speed equal to or higher than a predetermined speed and equal to or higher than a predetermined torque. The control device for a synchronous motor according to claim 1, wherein the value is increased from a value of the predetermined function IO (T) according to a torque command value.