JPH07308088A - Control device for synchronous motor - Google Patents

Abstract

PURPOSE:To accomplish control applicable to a wide range o rotation, from low speed to high speed, and effective operation of a synchronous motor using relatively simple components at low cost with respect to a control device for synchronous motors that obtains rotational force using the phenomenon that magnetic resistance varies depending on the rotational position of the rotor of a motor. CONSTITUTION:The control device for synchronous motors includes a relative phase angle creating means 9 that creates such a relative phase angle A as to make proper the exciting current component of motor current; and a current amplitude commanding means 8 that compensates the non-linearity of motor current amplitude and generated torque and thereby creates a proper motor current amplitude.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an AC servomotor, particularly a synchronous motor known as a reluctance motor.

[0002]

2. Description of the Related Art As a conventional improved reluctance motor controller, there is a controller for detecting a rotor position and a rotor speed as shown in FIG.

The conventional example of FIG. 11 will be described.

The synchronous motor has a salient-pole rotor 6, and a position detector 5 is mechanically coupled to the rotor 6 to output a position signal DS.

The rotor rotation position AR is the position of the magnetic pole of the rotor 6 with respect to the U-phase winding position as shown in FIG.

The speed command SI and the speed signal SD detected by the speed detecting means 3 are matched by the adder 1 to obtain the speed deviation ES, and the torque control means 2 performs the compensation calculation such as proportionality / integration / derivative. Obtain the torque command T.

The field current command circuit 24 receives the rotor rotational position AR detected by the rotor position detecting means 4 and the speed signal SD as input, and the following field current commands SFIU, SFI.
V, SFIW is created.

SFIU = SF.SIN (AR) SFIV = SF.SIN (AR + 120.degree.) SFIW = SF.SIN (AR + 240.degree.) Where SF is the amplitude of the field current and depends on the speed signal SD, and the base of the motor. Below a rotation speed NB, at a certain constant value,
Above the base speed NB, SF is increased as the speed signal SD increases so that the value obtained by multiplying NB and SF becomes a constant value.
Has a characteristic of decreasing.

The armature current command circuit 21 uses the torque command T
And the rotor rotation position AR are input, and the following armature current commands SAIU, SAIV, SAIW are output.

SAIU = T.SIN (AR + 90 °) SAIV = T.SIN (AR + 90 ° + 120 °) SAIW = T.SIN (AR + 90 ° + 240 °) Each current command SIU, SIV, SIW is expressed by the following equation.
Each armature current command and each field current command are added to each adder 25,
It is obtained by adding 26 and 27.

SIU = SAIU + SFIU SIV = SAIV + SFIV SIW = SAIW + SFIW The angle representing the direction of the combined vector of these phase currents is angle A with respect to the direction of the rotor magnetic pole in FIG.

The current control circuit 7 controls the current commands SIU, SI
Amplification is performed by using V and SIW as inputs, and three-phase currents IU, IV and IW are supplied to the respective windings U, V and W of the three-phase motor.

As a result of such control, the rotor 6
Since the control of the field magnetic flux and the armature current component that generates torque are appropriately controlled according to the direction of the magnetic poles of the rotor 6,
The torque of right rotation or left rotation can be arbitrarily generated in
The speed of the electric motor is controlled.

[0014]

The control device shown in FIG. 11 has the following problems because the electric motor characteristics actually have electromagnetic non-linear elements etc. even though it is good in terms of simple logic. It was

Since the magnetic circuit of the electric motor is non-linear, the field magnetic flux proportional to the field current cannot be obtained and the expected output torque cannot be obtained.

When a large armature current component is supplied to obtain a large output torque, the field magnetic flux changes due to this armature current component, and the expected torque cannot be obtained.

There is a voltage drop due to the leakage inductance of the motor winding, and the expected torque cannot be obtained at high speed rotation.

If stricter control is required, each compensation control or the like is required, and as a result, the control content becomes complicated and the cost also increases.

[0019]

In order to solve the above problems, the present invention determines the relative phase angle between the rotor and the motor current so that the magnetic flux in the motor becomes appropriate, that is, the exciting current component of the motor current. Relative phase angle creating means for creating a relative phase angle so that the armature current component becomes appropriate, and current amplitude command means for creating a suitable motor current amplitude by compensating for the non-linearity between the motor current amplitude and the generated torque. And so on.

[0020]

Operation: The relative phase angle creating means allows the rotation speed N of the electric motor to be changed.
The relative phase angle A determined by the torque command can be obtained by a relatively simple algorithm that reduces the relative phase angle A according to the number of revolutions in a range in which a certain function AS (T) and the torque command value T are large.

The current amplitude command means sets the current amplitude command to a substantially constant value in a range where the torque command is small, and when the torque command value T becomes large, it becomes a large value in accordance with a certain function IO (T), and the rotation speed is higher than the base rotation speed NB. The optimum current amplitude is obtained by a relatively simple algorithm that further increases the current command when it becomes higher.

[0022]

EXAMPLE FIG. 1 shows an example of the present invention. Description of the same components as those of the conventional example shown in FIG. 11 is omitted.

In FIG. 1, in order to determine the relative phase angle A between the magnetic poles of the rotor 6 and the vector direction of the electric motor current to an optimum value, a relative phase angle creating means 9 is provided, the input of which is a torque command T and a speed. The signal SD and are input. The calculated relative angle A is added to the rotor rotational position AR by the adder 15 and sent to the current command means 10.

Further, the torque command T and the speed signal SD
Is further sent to the current amplitude command means 8 and its output IOS
Is sent to the current command means 10. The current commands SIU, SIV, SIW for each phase of the current command means 10 are converted into three-phase currents IU, IV, IW to each winding U, V, W in the current control circuit 7, respectively.

The characteristics of the relative phase angle generating means 9 are shown in FIG. The characteristic depends on the torque command T and the rotation speed N of the electric motor.

The characteristic shown in FIG. 2 is a characteristic that follows the function AS (T) when the rotational speed is less than the basic rotational speed NB, and also a characteristic that follows the function AS (T) when the torque command T is -T1≤T≤ + T1. Is.

When the rotation speed is equal to or higher than the base rotation speed NB and the torque command value T exceeds the predetermined value T1, the torque command T
Along with this, it decreases from the function AS (T) and converges to the minimum value AMN of the relative phase angle A. When the torque command value T decreases from the predetermined value −T1, the relative phase angle A increases from the function AS (T) along with the torque command T, and the relative phase angle A
Of the maximum value AMX. An example of a specific algorithm for realizing this characteristic is shown in the flowchart of FIG.

FIG. 4 shows a representative characteristic ADC (T) of a portion of the characteristic shown in FIG. 2 which has a rotational speed dependency and a torque dependency.
Is.

In the flowchart of FIG. 6, first, the characteristic AX excluding the characteristic AS (T) of FIG. 2 is obtained.

AX = ADC (| T |)-(N3- | N |) · KA (1) However, when AX becomes negative according to this calculation formula, it is replaced with zero.

N3 is the maximum number of revolutions.

KA is a coefficient representing the rotation speed dependency.

Next, the direction of the torque command value T is discriminated and A
The characteristic of the relative phase angle A in FIG. 2 is obtained by adding or subtracting the variable AX to S (T).

However, when the calculated relative phase angle A is not within the range of the maximum value AMX and the minimum value AMN of the relative phase angle, the maximum value AMX or the minimum value AMN is replaced.

Thus, the characteristics shown in FIG. 2 can be obtained by a relatively easy algorithm.

The variable AX is defined in the equation (1), but this is explained as an example of the method, and other similar function or more strict function can be used.

An example of the characteristics of the current amplitude command means 8 is shown in FIG. The characteristic depends on the torque command T and the rotation speed N of the electric motor.

The characteristic shown in FIG. 3 is a characteristic that follows the function IO (T) when the rotational speed is less than the basic rotational speed NB, and also a characteristic that follows the function I0 (T) when the torque command T is -T1≤T≤ + T1. Is.

When the rotation speed is equal to or higher than the base rotation speed NB and the torque command value T exceeds the predetermined value T1, the torque command T
With the function IO (T).

An example of a concrete algorithm for realizing this characteristic is shown in the flowchart of FIG.

FIG. 5 shows a representative characteristic IDC (T) of a portion of the characteristic shown in FIG. 3 which has a rotational speed dependency and a torque dependency.
Is.

In the flowchart of FIG. 7, first, the characteristic IOX excluding the characteristic IO (T) of FIG. 3 is obtained.

IOX = IDC (| T |)-(N3- | N |) KB (2) However, when IOX becomes negative according to this calculation formula, it is replaced with zero.

KB is a coefficient representing the rotation speed dependency.

Next, the characteristic of the current amplitude command A in FIG. 3 can be obtained by adding the IOX to the function IO (T).

Thus, the characteristics shown in FIG. 3 can be obtained by a relatively easy algorithm.

The variable IOX is defined in the equation (2), but this is explained as an example of the method, and other similar function or more strict function can be used.

The current command means 10 is a current command SI for each phase.
U, SIV, SIW are obtained according to the formula below.

SIU = IOS.SIN (AR-A + 90 °) SIV = IOS.SIN (AR-A + 90 ° + 120)
SIW = IOS / SIN (AR-A + 90 ° + 240)
As a result of such control, the magnetomotive force component acting on the field magnetic circuit including the rotor is a value proportional to IOS.SIN (A), and the current component flowing in the winding facing the rotor, that is, the armature current component is The value is proportional to IOS · COS (A).

Since the output torque of the electric motor is proportional to the product of the field magnetic flux and the armature current orthogonal to the field magnetic flux, if the current amplitude value IOS and the relative phase angle A can be controlled arbitrarily, the output torque will also be arbitrary. Since it can be controlled, highly accurate speed control and position control are possible.

However, any type of synchronous motor cannot be efficiently operated by the above control.

The present invention can realize more efficient operation in a synchronous motor in which a magnetic field flux can be stably generated by the magnetic poles of the rotor and a current interlinking with the magnetic field flux can flow.

An example of a synchronous motor particularly suitable for the present invention will be shown.

FIG. 8 is an example of a rotor sectional view of a synchronous motor. Reference numeral 30 is a rotor shaft, 31 is an electromagnetic steel plate and is laminated, and 32 is a magnetic insulating portion made by punching out the electromagnetic steel plates.

Each magnetic path is connected to the outer peripheral portion of the rotor and the middle portion of each magnetic pole. The main purpose of this is to make the electromagnetic steel plate 31 easy to handle and assemble, and to firmly fix it to the shaft 30 as a rotor. And electromagnetically, the necessity is low.

In the rotor having such a structure, the magnetic resistance to the adjacent magnetic poles is small and the field magnetic flux can be easily excited, and the magnetic resistance seen from both ends in the rotational direction of the magnetic poles is large, so that the magnetic poles face each other. The disturbance of the field magnetic flux of each magnetic pole due to the current flowing through the stator is small. As a result, the synchronous motor can be operated more efficiently.

When the outer peripheral portion of the rotor is connected, there is an effect of reducing the torque ripple caused by the slot and an effect of reducing the magnetic resistance of the field.

FIG. 9 is another example of a sectional view of the rotor of the synchronous motor. Reference numeral 30 is a shaft of the rotor, 33 is a laminated electromagnetic steel plate, and 34 is a magnetic insulating portion formed by punching electromagnetic steel plates.

The magnetic paths are connected to the outer peripheral portion of the rotor and the middle portions of the magnetic poles for the same purpose and effect as in the example of FIG.

The example of FIG. 9 is different from that of FIG. 8 in that the space between the magnetic poles is more effectively utilized as magnetic poles, and the change in the rotational direction of the rotor magnetic resistance is relatively smooth.

The current control circuit 7 has a structure as shown in FIG. 10, for example, and an inverter 54 composed of power transistors and current detectors 55, 56 and 57 for detecting the currents IU, IV and IW of the motor phases. Current command value SI for each phase
It is composed of current control circuits 51, 52 and 53 which feed back the detected current value of each phase to U, SIV and SIW and supply a drive signal from each difference signal to each power transistor of the inverter 54.

By carrying out such control, specifically, the exciting current component IF of the electric motor is kept substantially constant below the base speed NB, and the back electromotive force of the electric motor corresponds to the power supply voltage of the control device above the base speed NB. You can prevent it from being exceeded.

Further, the increase of the current amplitude value at the base speed NB or more shown in FIG. 3 compensates for the decrease of the output torque due to the decrease of the field magnetic flux at the high speed rotation.

A method for realizing such electric motor control by a microprocessor is as follows: AS (T) in FIG. 2, IO (T) in FIG. 3, and functions similar to those in FIGS. 4 and 5. Is stored as a mathematical formula, and is calculated and calculated each time,
There is a method in which the function pattern is stored in the memory and the value of ADC (N) corresponding to the rotation speed N is read out each time.

Of course, the algorithms shown in FIGS. 6 and 7 can be easily realized by a microprocessor.

In the control of FIG. 1 which is an embodiment of the present invention, not only basically excellent speed control is realized, but also it is realized by a function and an algorithm having high flexibility, so various compensation control is controlled. It can be incorporated with almost no change in the algorithm.

For example, as an example of compensation, the compensation control of the fact that the field current component and the actual field magnetic flux are non-linear and have a saturation characteristic is performed by setting the slope of the function AS (T) shown in FIG. The compensation control of the so-called armature reaction, which is an influence of the armature current component on the field magnetic flux, can be dealt with by changing so as to perform inverse compensation, and as the amplitude value of the torque command increases, the function AS (T) This can be dealt with by making the slope gentle, and regarding the voltage shortage of the current control circuit due to the voltage drop due to the leakage inductance of the electric motor winding, the maximum relative phase angle AMX is reduced and the minimum relative phase angle AM is decreased.
It can be dealt with by increasing N.

As described above, in the control device of the present invention, AS (T) in FIG. 2 is changed, IO (T) in FIG. 4. By changing the functions of FIGS. 4 and 5 and by slightly changing the control variable calculation algorithms of FIGS. 6 and 7, various compensation controls can be realized relatively easily.

Therefore, in the present invention, even if various compensation controls are performed, the load on the control CPU due to the addition of a new algorithm and the cost increase due to the addition of hardware do not occur.

In the embodiment of the present invention, the relative phase angle A
Was introduced from 0 to 180 °, the characteristic of the electric motor of the present invention is that the relative phase angle A is from 0 to 180 °.
The characteristic of ° and the characteristic of 180 to 360 ° are the same,
Even if the control phase angle is shifted by 180 ° for control, the same control characteristic can be obtained.

In the example of the present invention, the method of directly controlling the voltage and current of the electric motor with the three-phase alternating current amount is described. However, so-called three-phase / two-phase conversion is performed and written and controlled using dq coordinates. However, they are equivalent and are included in the present invention.

The present invention can be applied to a multi-phase alternating current other than the three-phase alternating current by modifying the present invention.

The speed control unit, the position detection unit, and the like can be modified in various ways and are included in the present invention.

[0074]

With the control device of the present invention, it is possible to control the speed from the low speed rotation to the high speed rotation and from the low torque output to the high output torque with high accuracy using a relatively simple control algorithm.

Specifically, the magnetic circuit of the electric motor is non-linear,
As a method of compensating for the influence of the change in the field magnetic flux due to the armature current component and the voltage drop due to the leakage inductance of the motor winding, the characteristics of FIGS. 2, 3, 4, and 5 should be modified. 6 and FIG. 7 can be realized without changing the basic control form by only slightly changing the control variable calculation algorithm.

Therefore, highly precise speed control can be realized, and at the same time, the structure of the control device can be simplified, so that the cost can be reduced.

[Brief description of drawings]

FIG. 1 is a diagram showing an embodiment of a reluctance motor control device according to the present invention.

FIG. 2 is a characteristic diagram of a torque command T and a relative phase angle A in the embodiment.

FIG. 3 is a torque command T and a current amplitude command I in the embodiment.
It is a characteristic view with OS.

FIG. 4 is a torque command T and a variable ADC in the embodiment.
It is a characteristic view with (T).

FIG. 5 is a torque command T and a variable IO (T) in the embodiment.
FIG.

FIG. 6 is a flowchart showing an algorithm for obtaining a relative phase angle A in the embodiment.

FIG. 7 is a flowchart showing an algorithm for obtaining a current amplitude command IOS in the embodiment.

FIG. 8 is a rotor sectional view of a synchronous motor suitable for the present invention.

FIG. 9 is a sectional view of a rotor of another synchronous motor suitable for the present invention.

10 is a diagram showing a specific example of a current control circuit 7. FIG.

FIG. 11 is a diagram showing an example of a conventional technique.

[Explanation of symbols]

 2 torque control means 3 speed detection means 4 rotor position detection means 6 rotor 7 current control circuit 8 current amplitude command means 9 relative phase angle creation means 10 current command means

Claims (3)

[Claims] 1. A control device for a synchronous motor that obtains a rotational force by utilizing the fact that magnetic resistance varies depending on the rotational position of a rotor of an electric motor, and a position detecting means for detecting the rotational position and speed of the rotor; This is the direction in which one magnetic pole of the rotor is oriented, and the torque control means that obtains a speed deviation from the speed command and the detected speed of the rotor, performs proportional, integral, and differential compensation on this speed deviation to output the torque command. When the torque command value T is zero, the value of the relative phase angle A between the rotor position and the vector direction of the motor current flowing through the three-phase AC winding of the stator is 90 ° in electrical angle, and the torque command value is zero. The relative phase angle A gradually decreases as the torque command value approaches the maximum value as the torque command value approaches the maximum value. Similarly, the relative phase angle A gradually decreases as the torque command value decreases from zero to a negative value. 1 Relative phase angle creating means for taking a value of a predetermined function AS (T) that asymptotically approaches 80 °, and a current amplitude command value IOS of the electric motor, which is a substantially constant value or torque command from a torque command value T of zero to a predetermined value T1. A value that slightly increases from a constant value according to T, and a current amplitude command means that becomes a predetermined function IO (T) that gradually increases with the torque command value when the torque command value is T1 or more, the relative phase angle A and the current. A current command means for inputting the amplitude command value IOS to create a current command value for each phase of the motor, and a current control circuit for supplying a current to each phase winding of the motor according to the current command value for each phase of the motor. Characteristic synchronous motor control device. 2. The relative phase angle A is set to the motor rotation speed and the torque command value so that the motor terminal voltage becomes equal to or lower than a controllable voltage value when the rotation speed is equal to or higher than a predetermined torque and is equal to or higher than a predetermined torque. Accordingly, when the torque command value is positive, the predetermined function AS increases as the torque command value increases.
It is designed to increase from the value of (T), and when the torque command value is negative, it decreases from the value of the predetermined function AS (T) as the torque command value decreases. The control device for a synchronous motor according to claim 1. 3. The current amplitude command value IOS sets the current amplitude command value IOS to the motor rotation speed so as to compensate for the reduction ratio in which the effective magnetic flux of the motor decreases and the torque decreases when the current amplitude command means exceeds the predetermined rotation speed and exceeds the predetermined torque. The control device for a synchronous motor according to claim 1, wherein the control function is increased from the value of the predetermined function IO (T) according to the torque command value.