JP2010119263A - Motor and controller for the same - Google Patents

Abstract

<P>PROBLEM TO BE SOLVED: To provide a motor capable of attaining miniaturization, improvement in efficiency, cost reduction and enhancement in quality by the motor that does not use permanent magnets or does not use fewer permanent magnets. <P>SOLUTION: In the motor, teeth for salient poles and a rotor including the salient poles consisting of a soft magnetic material are configured by three phases, full-pitch winding and concentrated winding so that each tooth 117, 118 and 119 for a stator 110 and the salient poles 161 and 162 constituted of the soft magnetic material of the rotor can generate an attraction force discretely in a high flux density. In addition, windings 111 to 116 are formed in a motor configuration, capable of being driven by a one-way current, and the motor can be driven at a low cost with high efficiency; and further, a magnetic saturation reducing measure that applies bias flux and approximately doubling the flux fluctuation range of the soft magnetic material is carried out, and a maximum torque can be increased. <P>COPYRIGHT: (C)2010,JPO&INPIT

Description

  The present invention relates to a motor mounted on an automobile, a truck, and the like and a control device thereof.

Three-phase AC motors have been widely used conventionally. FIG. 66 is an example of a longitudinal sectional view showing the schematic configuration. 811 is a motor output shaft, 812 is a rotor core, 817 and 818 are N-pole permanent magnets and S-pole permanent magnets mounted on the rotor surface, 813 is a bearing, 814 is a stator core, 815 is a coil end of a winding, and 816 is a motor case. It is.
67 is a transverse sectional view showing a section AA-AA in FIG. This motor has three-phase alternating current, two poles and twelve slots, and the windings are full-pitch winding and distributed winding.
The teeth of the stator are 821, 822, 823, 824, 825, 826, 827, 828, 829, 82A, 82B, and 82C in the circumferential direction. Three-phase windings are wound in each slot sandwiched between the teeth, the U-phase windings are 82D, 82J, 82K, and 82Q, and the V-phase windings are 82H, 82N, 82P, and 82G, W-phase windings are 82M, 82E, 82F, and 82L.

FIG. 68 is a winding diagram, and the horizontal axis indicates each position in the rotational direction of the stator as an electrical angle. Since this motor is an example of a two-pole motor, the mechanical angle and the electrical angle are the same.
U is the terminal of the U-phase winding, the U-phase current Iu is applied, V is the terminal of the V-phase winding, the V-phase current Iv is applied, and W is the W-phase winding, the terminal is supplied with the W-phase current Iw. , N is the neutral point of the three-phase Y connection. 831 is a connecting wire for the U-phase winding, 832 is a connecting wire for the V-phase winding, and 833 is a connecting wire for the W-phase winding.
FIG. 69 is a connection diagram of a three-phase winding, and is a Y connection. 834 is a U-phase winding, 835 is a V-phase winding, and 836 is a W-phase winding.
The surface magnet type brushless motor shown in FIGS. 66 to 69 is widely used as an excellent motor.
Japanese Patent Laying-Open No. 2005-110431 (FIGS. 1 to 6)

However, from the viewpoints of higher performance, smaller size, and lower cost, there are the following problems depending on the application.
As basic characteristics of the motor, the force F is expressed by Fleming's law, and the torque T is expressed by the following equation.
F = BIL ……………………………………………………………………………… (1)
T = FR …………………………………………………………………………………… (2)
Here, I is the acting current, L is the length of the winding that works effectively, and R is the radius of the rotor. In the case of the motor shown in FIG. 67, the maximum magnetic flux density Bmax = 2 [T] of the magnetic steel plate at the tooth portion, the circumferential length Ws of the slot near the air gap portion between the stator and the rotor, and the circumferential direction of the tooth Assuming that the lengths Wt are the same, the average magnetic flux density acting as the motor torque is 1 [T]. Therefore, as a problem of the motor shown in FIG. 67, there is a problem that the maximum magnetic flux density Bmax = 2 [T] obtained as a magnetic material is not sufficiently utilized.
The peak torque is an important characteristic from the viewpoint of miniaturization of the motor, but the peak torque is also insufficient from the same viewpoint.

As inferred from FIG. 67, the problem of the stator is that the opening of the slot is narrow, and the winding factor is low due to the difficulty of winding of the winding, so the torque decreases, Since the length of the coil end in the axial direction of the rotor is increased, there is a problem that the motor is increased in size, and the productivity of the winding is also decreased, resulting in a problem of production cost.
As a problem of the rotor shown in FIG. 67, there is a problem of the rotor strength and a problem of the maximum allowable rotational speed due to the centrifugal force strength. In the case of a configuration capable of withstanding the centrifugal force acting on the permanent magnets 817 and 818, there is a problem of the reinforcement cost. Since the output Pout of the motor is expressed as a product of torque and rotational angular velocity ω, there is a problem that the rotational speed restriction becomes an output restriction.
In the case of a surface magnet type brushless motor, field weakening control is difficult, and constant output control by field control is difficult.

A so-called rare earth magnet composed of neodymium Nd, iron Fe, and boron B is generally used for a high-performance permanent magnet type brushless motor, but it is an expensive material and has a problem of cost. In particular, in recent years, it has attracted attention and concern as a problem of resource depletion.
In addition, motors are used under various load conditions. Electric vehicles, hybrid vehicles, etc. require a large maximum torque, but they are relatively light loads during normal operation, and the motor efficiency at light loads is high. It works strongly as a fuel economy. From the viewpoint of motor efficiency, there are Joule loss due to current flowing in the winding, iron loss due to rotation of field magnetic flux, mechanical loss caused by bearings, etc., and iron loss due to permanent magnet field at light loads. There are not a few operating areas where this is a problem. In particular, in an operation region in which the motor rotates and the torque does not need to be generated, the iron loss component torque is also called “drag torque”, and the presence of the magnetic flux of the permanent magnet itself may be a problem.

68 is a winding diagram of the motor of FIGS. 66 and 67, and the horizontal axis indicates the rotation angle of the rotor in electrical angle. Three-phase currents Iu, Iv, and Iw are supplied from the three-phase input terminals U, V, and W to the three-phase windings. 831 is a connecting wire of the U-phase winding, 832 is a connecting wire of the V-phase winding, 833 is a connecting wire of the W-phase winding, and N is a neutral point of the star connection. This winding diagram is full-pitch winding and distributed winding, each winding is complicated and difficult to manufacture, the winding space factor is reduced, and the coil ends are axial due to winding crossing There is a problem that becomes longer.
FIG. 69 shows a configuration when the motor is driven by a three-phase inverter. 84D is a DC power source such as a battery, 841, 842, 843, 844, 845, and 846 are power transistors, and 847, 848, 849, 84A, 84B, and 84C are diodes arranged in parallel to the respective power transistors. 834 is a U-phase winding, 835 is a V-phase winding, and 836 is a W-phase winding. Many power transistors and the like are necessary, and there is a problem of the cost.
In addition, since two transistors are connected in series when current is supplied to the winding, there is a problem of inverter efficiency due to on-loss of the transistors. As a matter of course, a motor and a control device including an inverter are a set, and competitiveness is required in terms of performance, size, and cost.

FIG. 70 shows an example of a three-phase AC motor having a two-pole multi-flux barrier type rotor.
The stator is the same as in FIG. A rotor 851 is a thin magnetic path made of a soft magnetic material, and 852 is also called a flux barrier because a slit-like air gap blocks magnetic flux. In this figure, the upper side and the lower side of the rotor are directions in which the magnetic flux of the rotor passes, that is, the magnetic poles of the rotor. Since the magnetic flux density of these magnetic poles and the slits are alternately arranged, there is a problem that the maximum magnetic flux density cannot be used because it is about half the saturation magnetic flux density of the soft magnetic material. The power factor of this type of motor is about 0.6 to 0.8, not high.
In addition, the rotor has many thin portions of soft magnetic material, and there is a problem of securing strength against centrifugal force at high speed rotation. The inverter to drive is an inverter as shown in FIG.

A cross-sectional view of another conventional motor is shown in FIG. A motor called a switched reluctance motor, which has six magnetic poles on the stator and four magnetic poles on the rotor. Although much research has been done, there are still few examples of practical application.
JP-A-2002-272071 (FIG. 1) 861 is an A-phase stator pole, 867 and 868 are A-phase windings, 864 is a negative A-phase tooth, 86E and 86D are negative A-phase windings, 865 is a B-phase stator pole, 86F and 86G are B-phase windings, 862 is a negative B-phase tooth, 86A and 869 are negative B-phase windings, 863 is a C-phase stator pole, 86B and 86C Is a C-phase winding, 866 is a negative C-phase tooth, and 86J and 86H are negative C-phase windings. 86L is a salient pole of the rotor.

The switched reluctance motor shown in FIG. 71 does not use a permanent magnet and is low-cost. The winding of the stator has a simple configuration with concentrated winding, and the magnetic flux acting on the salient pole of the stator and the salient pole of the rotor is Since it operates at the saturation magnetic flux density of the electromagnetic steel sheet, it is possible to use an electromagnetic phenomenon with a high magnetic flux density, and because the rotor is robust, high-speed rotation is possible.
The problem with the switched reluctance motor shown in FIG. 71 is that the radial force acting between the stator and the rotor changes with rotation to a position that differs by 90 ° in the circumferential direction, and the drive current acts like a switch. Therefore, there is a problem that the radial deformation of the stator is particularly large, and vibration and noise are large. With respect to the use efficiency of the windings, the current that is energized to generate torque is energized to four of the twelve windings shown in FIG. 71, and 4/12 = 1/3 windings. There is a problem that the use efficiency is low, and as a result, the Joule loss, which is the heat generation of the windings, becomes large.

  72 shows an example of an inverter for driving the motor of FIG. 871, 872, 873, 874, 875, and 876 are power transistors, and 877, 878, 879, 87A, 87B, and 87C are diodes arranged in antiparallel to the power transistors and the windings 87D, 87E, and 87F. Energy can be returned to the DC power supply 84D, or a flywheel current can be passed through each transistor and winding. Here, the winding 87D is a winding in which A-phase windings 867, 868 and 86E, 86D are wound in series. The winding 87E is a winding in which B-phase windings 86F and 86G and 86A and 869 are wound in series. The winding 87F is a winding in which C-phase windings 86B and 86C and 86J and 86H are wound in series.

  The motor of the present invention has a slot for winding a three-phase full-pitch winding, a stator core having six teeth at an electrical angle of 360 °, and full-pitch turns arranged in each slot of the stator. The rotor has a concentrated three-phase winding and a main salient pole formed of a soft magnetic material. Each winding is supplied with a current in one direction instead of alternating current, and can be driven by a simpler inverter. Since the stator salient poles and the rotor salient poles act so as to attract each other, if a normal silicon steel plate is used, operation with a magnetic flux density of about 2 Tesla is possible, and a large torque can be obtained. it can. Also, the motor does not require an expensive permanent magnet and is inexpensive.

In addition, the motor of the present invention is provided with two main salient poles and two auxiliary salient poles between the rotor electrical angle of 360 °, so that it is possible to drive with low torque ripple, especially at low speed rotation. It is.
The torque can be effectively obtained by setting the circumferential width Hm of the main salient pole magnetic pole, the circumferential width Hh of the auxiliary salient pole magnetic pole, and the width Hs of the opening of the slot in a predetermined relationship.
In the rotor of the present invention, the main salient pole magnetic pole and the auxiliary salient pole magnetic pole are alternately arranged in the circumferential direction. You can also. Moreover, the motor of this invention can reduce each magnetic saturation and can increase a torque by adding a soft magnetic body to the rotor axial direction end of a stator tooth.

Further, the motor of the present invention can reduce the respective magnetic saturation and increase the torque by adding a soft magnetic material and a permanent magnet to the stator tooth or the rotor axial end of the rotor magnetic pole.
In addition, the motor of the present invention can reduce the magnetic saturation and increase the torque by adding a soft magnetic body and a coil for exciting magnetic flux to the stator tooth or the rotor axial end of the rotor magnetic pole. it can.
Moreover, the motor of this invention can reduce each magnetic saturation and can increase a torque by arrange | positioning a permanent magnet in the rotor side end vicinity of the slot of a stator.

Further, in the motor of the present invention, the shape of the stator teeth or the rotor main salient pole magnetic pole or the rotor auxiliary salient pole magnetic pole is larger in the circumferential width on the back yoke side than the circumferential width of the tip. By adopting the configuration, each magnetic saturation can be reduced and the torque can be increased.
Moreover, the motor of this invention can reduce each magnetic saturation and can increase a torque by adding a soft magnetic body to the rotor axial direction end of the auxiliary salient pole magnetic pole of the rotor.
The motor of the present invention can improve the distribution of magnetic flux and increase the torque by providing a slit in the main salient pole of the rotor from one magnetic pole to the other.

Moreover, the motor of this invention can improve a power factor and improve a torque by improving a magnetic flux distribution by arrange | positioning a permanent magnet to the said slit.
The motor of the present invention can increase torque more effectively by combining the various methods described above.
In the motor of the present invention, the cross-sectional area SBT of the winding arranged on the opening side of the slot is made smaller than the cross-sectional area SBB of the winding arranged on the back yoke side, thereby increasing the current density on the slot opening side. As a result, the leakage magnetic flux near the slot opening can be reduced, the magnetic saturation of the soft iron portion can be reduced, and the maximum torque of the motor can be increased.

In the motor of the present invention, the soft magnetic material SMA on the tooth slot opening side has a saturation magnetic flux density relatively larger than the saturation magnetic flux density of the soft magnetic material SMB on the tooth slot back side, so that the soft iron portion The magnetic saturation of the motor can be reduced and the maximum torque of the motor can be increased.
Moreover, the motor of this invention can reduce the loss of a rotor effectively by using an amorphous metal for the soft magnetic body of a rotor.
Further, in the N-pole motor of the present invention, the number of winding bundles of the stator is 3N / 2, and the number of windings arranged on the stator from the motor center to the outer periphery is one at the same circumferential angle. Or it is comprised so that it may be two pieces, the interference and crossing of each coil | winding can be reduced, the space factor of a coil | winding can be improved, and it can be set as a highly productive coil | winding.

In the motor of the present invention, the shape of some slots is expanded toward the back yoke side in the vicinity of the axial end of the stator core, and the shape of other slots is axial in the vicinity of the axial end of the stator core. Since the slot shape has been improved, each winding can be smoothly and smoothly wound in a circular arc shape at the corner of the stator core, and simultaneously wound in each slot. Winding physical interference can be greatly reduced. As a result, the amount of protrusion of the coil end portion in the rotor axial direction can be greatly reduced.
Furthermore, the problem that the magnetic path of the back yoke is reduced due to the deformation of the slot shape can be reduced by arranging the stator core on the outer peripheral side of the coil end portion.
In addition, by expanding the slot near the rotor axial end, it is possible to secure a space for crossing the insulating paper between the winding and the stator core, and to easily connect the insulating paper in the slot and the insulating paper near the stator core end. Can do.

In addition, since the motor of the present invention can drive the current of each winding even with a unidirectional current, one end KD of each winding of the motor is connected to the stable potential of the power source of the inverter, and the winding close to one end KD is connected. A motor system that is wound to a position closer to the stator core in the slot, reduces the current flowing through the stray capacitance between the winding and the stator core, and is resistant to electromagnetic noise, including the inverter.
Further, the motor of the present invention is composed of a combination of split cores in which the circumferential width of the soft magnetic material of the stator is divided into a width in the vicinity of 360 ° in electrical angle, and the winding of each winding is within the split core. It is a complete structure. In this configuration, three winding bundles are arranged in the circumferential width of 360 ° of the stator. Therefore, a motor can be manufactured by combining a plurality of divided cores wound with windings.

In addition, the motor of the present invention can be configured as a composite motor in which the first stator and the rotor are arranged on the outer diameter side of the motor and the second stator and the rotor are arranged on the inner diameter side of the motor.
In the motor of the present invention, the first rotor is disposed on the outermost diameter side of the motor, the second rotor is disposed on the outermost inner diameter side of the motor, and the first rotor and the second rotor are disposed between the first rotor and the second rotor. A first stator and a second stator are arranged, and a first set of windings of both motors are wound between a slot of the first stator and a slot of the second stator. The second set of windings are wound between the slots. The problem that the area of the inner slot differs from the area of the outer slot can be solved by winding the second set of windings.

In the motor of the present invention, the interior of the motor core is joined by welding or the like at an electrical multiple pitch of 360 °. This is because when an electromagnetic steel sheet is welded by welding or the like, an electrically closed circuit may be formed in the motor because the welded portion is electrically conductive. When there is a magnetic flux that changes in linkage with the closed electric circuit, a current that loops through the closed electric circuit flows, causing a problem of Joule loss.
The motor of the present invention operates with a three-phase direct current or alternating current, but the idea can be expanded to a four-phase motor for higher performance. Furthermore, the motor can be expanded to a five-phase motor. In particular, in the case of the five-phase motor, the harmonic component of the motor vibration can be greatly reduced compared to the three-phase and four-phase motors. Can be realized.

Moreover, the motor of this invention can reduce the magnetic flux leakage and magnetic saturation of a rotor by arrange | positioning a permanent magnet or a conductor around a rotor magnetic pole.
In the motor of the present invention, a plurality of windings of the same phase are wound in parallel in each slot of a three-phase stator, and a diode D1 is connected in series to one side of the parallel winding to supply power. Similarly, the diodes D2 and D3 are connected in series to the power supply for the other two-phase windings, and the other winding is connected to the transistors TR1, TR2 and TR3 for controlling the motor current. That is, each winding is a winding such as a bi-directional winding and separated into a powering winding and a regenerative winding, thereby realizing simplification, miniaturization, high efficiency and low cost of the inverter.

In the motor according to the present invention, one terminal of each three-phase winding of the motor is connected to a power source, and the other end of each of the three-phase windings is connected to each driving transistor TR4, TR5, TR6. Each diode D4, D5, D6 is connected to each connection point between each other end of the phase winding and each driving transistor TR4, TR5, TR6. In the case of driving with a unidirectional current, the inverter can be simplified, reduced in size, increased in efficiency, and reduced in cost. At this time, the regenerative power is regenerated to the DC power source by the DC-DC converter. Further, when driving a plurality of motors, the DC-DC converter can be shared, so when the motor of the present invention is driven with a unidirectional current, it can be driven with three transistors.
In the motor of the present invention, one terminal of each three-phase winding of the motor is connected to the emitter of the transistor TR7, and each other end of the winding is connected to the collectors of the transistors TR8, TR9, and TR10. When the motor of the present invention is driven with a unidirectional current, it can be driven with four transistors.

In the motor of the present invention, one terminal of each three-phase winding of the motor is connected to each emitter of the transistors TR11, TR12, and TR13, and each other end of the winding is connected to the collectors of the transistors TR14, TR15, and TR16. Connecting. When the motor of the present invention is driven with a one-way current, a three-phase current can be freely driven, and torque and efficiency can be improved as compared with the driving method using the four transistors.
In the motor of the present invention, one terminal of the winding WU of the motor is connected to the emitter of the transistor TR17 and the collector of the transistor TR18, and one terminal of the winding WV is connected to the emitter of the transistor TR19 and the collector of the transistor TR20. And one terminal of the winding WW is connected to the emitter of the transistor TR21 and the collector of the transistor TR22, and the other ends of the windings WU, WV, and WW are connected to each other. It can be driven by a three-phase alternating current.

  In the motor of the present invention, the transistor TR23 and the transistor TR24 are connected in series between the positive voltage and the negative voltage of the power source, one end of the winding WU is connected to the middle point, and the other end of the winding WU is connected. Is connected to an intermediate point between the transistor TR25 and the transistor TR26 arranged in series between the positive voltage and the negative voltage of the power supply, and the transistor TR27 and the transistor TR28 are connected in series between the positive voltage and the negative voltage of the power supply. Then, one end of the winding WV is connected to the intermediate point, and the other end of the winding WV is connected to the intermediate point between the transistor TR29 and the transistor TR30 arranged in series between the positive voltage and the negative voltage of the power source, The transistor TR31 and the transistor TR32 are connected in series between the positive voltage and the negative voltage of the power supply, one end of the winding WW is connected to the middle point, and the other end of the winding WW is connected to the positive voltage of the power supply. Connecting to an intermediate point of the transistor TR33 and the transistor TR34 disposed in series between the voltage. Twelve transistors can control three-phase positive and negative currents that are completely free, and the performance of the three-phase motor can be maximized.

Further, the motor and the control device of the present invention alternately drive the main salient pole magnetic pole and the auxiliary salient pole magnetic pole of the rotor at low speed rotation, and the drive time width is substantially proportional to the circumferential width of each magnetic pole. Current control for generating torque is performed, and motor control means is provided for reducing the current for driving with the auxiliary salient pole as the rotation speed increases. In low-speed rotation, there is little problem of noise that causes the stator core of the motor to vibrate near the drive frequency, and in high-speed rotation, control with the main salient pole is mainly controlled so that the stator core can be prevented from vibrating near the drive frequency.
In addition, the motor and the control device of the present invention include a data map in which current information of each winding according to the torque, rotational position, rotational speed, etc. of the motor is mapped, and is simple based on nonlinear and complicated control map information. Control with a control algorithm. Furthermore, magnetic flux information, inductance information, and voltage information can be added to the map information.

  The best mode for carrying out the present invention will be described in detail by the following examples.

Example 1
FIG. 1 shows a two-pole motor 110 according to an embodiment of the present invention.
Each slot of the stator 11F is wound with full-pitch and concentrated winding. 111 and 114 are A-phase windings, 113 and 116 are B-phase windings, and 115 and 112 are C-phase windings. Winding. The teeth 117, 118, 119, 11A, 11B, and 11C sandwiched between the slots constitute salient poles. The circumferential width represented by the electrical angle of the tooth tip is Ht, the circumferential width represented by the electrical angle of the opening of the slot is Hs, and the sum of both widths (Ht + Hs) is 60 in electrical angle. °. The rotational position of the rotor is indicated by θr.
The rotor 11E is a salient pole-shaped rotor made of a soft magnetic material. Most of the portion of 11D is a space, and it is possible to embed a nonmagnetic material for the purpose of reducing windage loss during rotation. The circumferential width represented by the electrical angle of the rotor salient pole is Hm as shown.

FIG. 1 shows an example of a two-pole motor whose operation is easy to explain. However, it is possible to increase the number of poles, and FIG. 2 shows a modified example of an eight-pole motor. The slots of the stator 12T are 121, 122, 123, 124, 125, 126, 127, 128, 129, 12A, 12B, 12C, 12D, 12E, 12F, 12J, 12K, 12L, 12M, 12N, 12P, 12Q, 12R. , 12S, and the rotor 12U has eight salient poles 12V.
Next, the operation of the motor shown in FIG. 1 will be described with reference to FIGS.
In this example, the slot opening width Hs is 20 °, and the rotor salient pole width Hm is 40 °.
FIG. 4 shows the rotor rotational position θr expressed in electrical angle on the horizontal axis, the current of each phase, and the torque of each phase. The A-phase current is Ia, the B-phase current is Ib, and the C-phase current is Ic.

  The torque of the motor shown in FIGS. 1, 2, and 3 is such that the winding is full-pitch winding and concentrated winding, and the stator teeth 117, 118, 119, 11A, 11B, and 11C are arranged almost all around. Therefore, in order to generate torque, current is passed through at least two windings to generate torque. Then, a reluctance torque is obtained by generating an attractive force between the salient pole-like teeth of the stator and the salient pole of the rotor. The torque generated between the stator teeth 11C and 119 and the rotor salient pole 11E is Ta, the torque generated between the starter teeth 118 and 11B and the rotor salient pole 11E is Tb, and the starter teeth 117 and 11A. Is a torque generated between the rotor salient poles 11E and Tc. It should be noted that at this time, each attractive force generates the same attractive force and torque regardless of whether the direction of the magnetic flux is positive or negative.

  When the rotor is in the vicinity of the rotational position of θr = 30 ° shown in FIG. 3A, a positive current Ia is supplied to the A-phase winding 131 and a negative current − is supplied to the A-phase winding 134 on the opposite side. Run Ia. At the same time, a positive current Ic is supplied to the C-phase winding 135 and a negative current -Ic is supplied to the opposite C-phase winding 132. No current flows through the B-phase windings 133 and 136. The currents Ia, Ib, and Ic of each phase are currents shown in (A), (C), and (E) of FIG. In this state, in accordance with Ampere's law, the magnetomotive forces of the A-phase current Ia and the C-phase current Ic act in the direction indicated by the thick arrows from the stator salient poles 11A to 117 in the direction indicated by the arrows. Magnetic flux is induced in the direction of 117. And the torque Tc shown to (F) of FIG. 4 generate | occur | produces in the counterclockwise rotation direction CCW at a rotor. Here, the permeability of the soft magnetic part of the stator and the rotor is sufficiently large, the permeability of the wide space between the stator and the rotor is sufficiently small, and the magnetic resistance of the narrow air gap between the stator and the rotor is sufficiently large In a simple model that is assumed to be small, the magnetic field strength [A / m] acting in the vicinity of the stator teeth 118, 119, 11B, and 11C is almost zero, and the magnetic flux that passes through these teeth in the radial direction is almost zero. The torque is almost zero.

  When the rotor rotates to CCW and rotates to the vicinity of the rotational position of θr = 50 ° shown in FIG. 3B, a positive current Ic flows through the C-phase winding 135, and a negative current flows through the C-phase winding 132. -Flow Ic. At the same time, a positive current Ib is supplied to the B-phase winding 133, and a negative current −Ib is supplied to the opposite B-phase winding 136. No current flows through the A-phase windings 131 and 134. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the bold arrows in the direction of the stator salient poles 118 to 11B, and the magnetic flux is induced in the direction indicated by the arrows. The Then, the rotor generates a torque Tb shown in FIG. However, the magnetic flux passes through the air portion in the vicinity of the slot openings of the windings 132 and 135, and since the magnetic resistance is large, the magnetic flux density is small and the torque Tb is not large. As the rotor salient pole 11E approaches the stator teeth 118 and 11B, the torque Tb increases rapidly.

  When the rotor rotates to CCW and rotates to near the rotation position of θr = 70 ° shown in FIG. 3C, the rotor further rotates with the torque Tb generated in CCW under the same current condition. Similarly, the torque Tb shown in FIG. 4D is generated from the rotor rotational position θr = 90 ° to θr = 110 ° in FIG.

When the rotor rotates to the vicinity of the rotational position of θr = 110 ° shown in FIG. 3E, a positive current Ib flows through the B-phase winding 133 and a negative current −Ib flows through the B-phase winding 136. At the same time, a positive current Ia is supplied to the A-phase winding 131, and a negative current −Ia is supplied to the opposite A-phase winding 134. No current flows through the C-phase windings 132 and 135. In this state, in accordance with Ampere's law, magnetomotive forces of the A-phase current Ia and the B-phase current Ib act in the direction indicated by the thick arrows in the direction from the stator salient poles 11C to 119, and magnetic flux is induced in the direction indicated by the arrows. The The rotor generates torque Ta as shown in FIG.
When θr is from 110 ° to around 120 °, the magnetic flux passes through the air portions near the windings 133 and 136, so that the magnetic resistance is large and the generated torque Ta is small. In the vicinity of the rotor rotational position θr = 130 ° in FIG. 3F, the torque Ta increases rapidly.

Here, in FIGS. 3A and 3D, the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed. In FIGS. 3B and 3E, the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed. In FIGS. 3C and 3F, the rotor angle is rotated by 60 °, but the operation is similar. However, the direction of the magnetic flux of the rotor is reversed.
As shown in FIGS. 3 and 4, the rotor can be rotated by changing the current that is sequentially supplied depending on the rotor rotational position θr. The motor torque Tm obtained by transferring the generated torques Ta, Tb, and Tc of each tooth is shown by a solid line in FIG. When the rotation direction end of the rotor salient pole 11E reaches the opening of the slot, the torque Tm decreases. There are many applications in which such a partial torque reduction is not a problem. In order to reduce this torque drop, there is a method of skewing the stator or rotor, and other methods will be described later.

  In these operations shown in FIG. 3, the direction of the magnetic flux of each salient pole of the stator is the same direction, the direction of the magnetic flux of the rotor is reversed depending on the rotational position, and A phase current Ia, B phase current Ib, C phase current Ic. These directions can be driven by a current in one direction. As for the magnitude of the current, the explanation has been made assuming that the magnitude of the current in each phase is the same, but it is also possible to change the current balance of each phase or to pass the current in all three phases. Different values can be taken for the width Hm of the rotor salient pole, the width Ht of the tip of the tooth, and the width Hs of the opening of the slot. As for the tip shape of the teeth of the stator, a simple salient pole shape has been illustrated and described, but there are various structures such as a structure that narrows the opening of the slot and a structure that widens the air gap with the rotor at the circumferential end of the tooth. Can be modified. Similarly, various modifications can be made to the shape of the rotor salient poles.

Next, the torque T generated by the motor will be considered qualitatively. In motors for automobiles, the torque required for motors during normal operation is ½ or less of the maximum torque, the frequency of using the maximum torque is low, and there are few applications where the motor efficiency at the maximum torque does not matter so much. There is. In order to reduce the size and cost of the motor in such applications, the maximum torque characteristic is important.
Now, the torque characteristics are divided into a linear operation region Aa in which the soft magnetic material of the portion where the stator teeth and the rotor salient poles face each other is not magnetically saturated, and a nonlinear operation region As in which the magnetic saturation is performed. In the motor model of FIG. 1, the rotor rotation position θr is considered near 30 °.
In the linear operation region Aa, when the magnetic flux density is Bx, the number of windings Nw of each phase, the current Ix, the proportional coefficient Kb of the magnetic flux density, the thickness Tc of the rotor axis of the motor, the rotational angular velocity ωr of the rotor, and the rotor radius R The magnetic flux density Bx and the magnetic flux φ are given by the following equations.
Bx = Kb × Ix × Nw ……………………………………………………………… (3)
φ = Tc × ωr × t × R × Bx = Tc × ωr × t × R × Kb × Ix × Nw (4)

At this time, the input power Pin is expressed by the following equation assuming that the winding resistance Ra is zero, the iron loss Pfe is zero, and the mechanical loss of the motor is zero.
Pin = 2 × V × Ix = 2 × Nw × dφ / dt × Ix (5)
The mechanical output Pout is expressed by the following equation.
Pout = T × ωr …………………………………………………………………… (6)
Therefore, from these equations, the torque T of the motor in the linear operation region Aa is given by the following equation.
T = 2 × Nw × dφ / dt × Ix / ωr
= 2 × Nw × d (Tc × ωr × t × R × Kb × Ix × Nw) / dt × Ix / ωr
= 2 × Nw × Tc × ωr × R × Kb × Ix × NwIx / ωr
= 2 × Nw × Tc × R × Kb × Nw × Ix ² …………………………………… (7)

The torque T in the linear operation region Aa is proportional to the proportional coefficient Kb of the magnetic flux density and is a value proportional to the square of the current Ix.
On the other hand, the saturation magnetic flux density is obtained as Bsat for the torque T in the non-linear operation region As where the soft magnetic material is magnetically saturated. The magnetic flux φ is given by the following equation.
φ = Tc × ωr × t × R × Bsat ……………………………………………… (8)
Torque is given by:
T = 2 × Nw × dφ / dt × Ix / ωr
= 2 × Nw × d (Tc × ωr × t × R × Bsat) / dt × Ix / ωr
= 2 × Nw × Tc × R × Bsat × Ix …………………………………… (9)

  The torque T in the non-linear operation region As of the soft magnetic material is a value proportional to the saturation magnetic flux density Bsat and the current Ix. Naturally, this maximum torque T is also proportional to the number of turns Nw of each phase, the thickness Tc of the motor in the rotor axial direction, and the rotor radius R. From this result, it was qualitatively verified that the maximum torque characteristic of the motor, which is an important characteristic for reducing the size and cost of the motor, greatly depends on the saturation magnetic flux density Bsat. However, as a condition for satisfying the formula (8), there is no magnetic saturation except in the portion where the stator and the rotor face each other, and the magnetic flux density in the circumferential space 11D of the rotor 11E is compared with the magnetic flux density of the soft magnetic body portion. It is necessary to be sufficiently small. These conditions are assumed to be equivalent for the motors being compared.

  The saturation torque density Bsat operating as a motor is important for the maximum torque of the motor, and it can be said that the maximum magnetic flux density is used at the operating point of FIG. A magnetic flux density of about 2 Tesla can be utilized if it is made of a normal electromagnetic steel sheet. The magnetic flux density acting on the torque generation of the conventional motor shown in FIG. 67 and FIG. 70 acts at the average magnetic flux density of the stator teeth and slots, so it acts roughly at half the saturation magnetic flux density Bsat. This is a low value. Therefore, in the motor model of FIG. 1, when the maximum torque of the motor is output near the rotor rotational position θr = 30 °, it can be said that the soft magnetic material is used up to the limit of the saturation magnetic flux density Bsat.

  Next, a method for improving the torque reduction portion shown in FIG. For example, in FIG. 3B, there is a problem in that the magnetic resistance near the opening of the slot increases and the torque decreases. As shown in FIG. 5, with the circumferential width of the rotor salient pole 11E, by adding soft magnetic projections 151, 152, 153, 154 closer to the inner diameter side than the rotor radius R, in the vicinity of this rotational position. Torque can be increased and improved. The distance between the stator teeth 118 and the protrusions 152 is shortened, and the magnetic resistance therebetween is reduced. The relationship between the stator teeth 11B and the protrusions 153 is the same.

(Example 2)
Next, another motor example of the present invention is shown in FIG.
For ease of explanation, a two-pole motor is shown. An auxiliary salient pole magnetic pole 162 is added to the rotor between the main salient pole magnetic pole 161 and the motor shown in FIG. The circumferential width represented by the electrical angle of the auxiliary salient pole magnetic pole 162 is Hh. In this example, the four salient poles of the rotor have an electrical angle difference of 90 ° from each other, and are arranged at equal intervals.
FIG. 7 shows a motor in which the motor shown in FIG. Compared to the motor of FIG. 2, an auxiliary salient pole magnetic pole 173 is added to the rotor 171. Reference numeral 172 denotes a main salient pole magnetic pole. Between the electrical angles of 360 °, the four salient poles of the rotor are arranged at equal intervals with an electrical angle difference of 90 ° between them.

The purpose of the motor configuration in FIG. 6 is to improve the torque reduction portion shown in FIG. The operation of the motor shown in FIG. 6 will be described with reference to FIGS. In this example, the circumferential width Hs of the opening of the slot is 20 °, the circumferential width Hm of the main salient pole magnetic pole 161 of the rotor is 40 °, and the circumferential width of the auxiliary salient pole magnetic pole is 20 °. .
FIG. 9 shows the rotor rotational position θr expressed in electrical angle on the horizontal axis, the current of each phase, and the torque of each phase. The A-phase current is Ia, the B-phase current is Ib, and the C-phase current is Ic.
The torque of the motor shown in FIGS. 6, 7, and 8 is such that the winding is full-pitch winding and concentrated winding, and the stator teeth 117, 118, 119, 11A, 11B, and 11C are arranged almost all around. Therefore, in order to generate torque, current is passed through at least two windings to generate torque. Then, a reluctance torque is obtained by generating an attractive force between the salient pole-like teeth of the stator and the salient pole of the rotor.

  The stator teeth 11C and 119 generate torque generated between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 of the rotor, and the stator teeth 118 and 11B connect the main salient pole magnetic pole 161 and the auxiliary salient pole of the rotor. The torque generated between the magnetic pole 162 and the stator teeth 117 and 11A between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 is Tc. It should be noted that at this time, each attractive force generates the same attractive force and torque regardless of whether the direction of the magnetic flux is positive or negative.

  When the rotor is in the vicinity of the rotational position of θr = 30 ° shown in FIG. 8A, a positive current Ia is supplied to the A-phase winding 111 and a negative current − is supplied to the A-phase winding 114 on the opposite side. Run Ia. At the same time, a positive current Ic is supplied to the C-phase winding 115, and a negative current −Ic is supplied to the opposite C-phase winding 112. No current flows through the B-phase windings 113 and 116. The currents Ia, Ib, and Ic in each phase are the currents shown in (A), (C), and (E) of FIG. In this state, in accordance with Ampere's law, the magnetomotive forces of the A-phase current Ia and the C-phase current Ic act in the direction indicated by the thick arrows from the stator salient poles 11A to 117 in the direction indicated by the arrows. Magnetic flux is induced in the direction of 117. Then, the torque Tc shown in FIG. 9F is generated in the counterclockwise direction CCW. Here, the permeability of the soft magnetic part of the stator and the rotor is sufficiently large, the permeability of the wide space between the stator and the rotor is sufficiently small, and the magnetic resistance of the narrow air gap between the stator and the rotor is sufficiently large In a simple model that is assumed to be small, the magnetic field strength [A / m] acting in the vicinity of the stator teeth 118, 119, 11B, and 11C is almost zero, and the magnetic flux that passes through these teeth in the radial direction is almost zero. The torque is almost zero.

  When the rotor rotates to CCW and rotates to the vicinity of the rotation position of θr = 50 ° shown in FIG. 8B, a positive current Ia flows through the A-phase winding 111, and a negative current flows through the A-phase winding 114. -Flow Ia. At the same time, a positive current Ib is supplied to the B-phase winding 113, and a negative current −Ib is supplied to the opposite B-phase winding 116. No current flows through the C-phase windings 115 and 112. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the thick arrows in the direction of the stator salient poles 118 to 11B. Magnetic flux is induced in the direction of 119. Then, the rotor generates torque Ta as shown in FIG. In this motor model, since the circumferential width of the auxiliary salient pole magnetic pole 162 of the rotor is as narrow as 20 °, the torque width near θr = 60 ° shown in FIG. 9B is narrow.

  When the rotor rotates to CCW and rotates to the vicinity of the rotational position of θr = 70 ° shown in FIG. 8C, a positive current Ic flows through the C-phase winding 115 and a negative current flows through the C-phase winding 112. -Flow Ic. At the same time, a positive current Ib is supplied to the B-phase winding 113, and a negative current −Ib is supplied to the opposite B-phase winding 116. No current flows through the A-phase windings 111 and 114. In this state, in accordance with Ampere's law, the magnetomotive forces of the B-phase current Ib and the C-phase current Ic act in the direction indicated by the thick arrows in the direction of the stator salient poles 118 to 11B. Magnetic flux is induced in the direction of 11B. Then, the rotor generates a torque Tb shown in FIG. And it rotates to CCW and rotates to the rotation position of (theta) r = 90 degrees shown to (d) of FIG.

The motor shown in FIG. 6 has a periodicity of 60 ° in electrical angle, and a similar drive can be performed at a cycle of 60 °. The relative relationship between the operation at θr = 30 ° in FIG. 8A and the operation at θr = 90 ° in FIG. 8D is opposite in the direction of current and the direction of magnetic flux. However, both torques T have the same magnitude in the CCW direction. Thus, when the rotational position θr is between 90 ° and 150 °, torque can be generated and rotated in a similar manner to the motor operation when the rotational position θr is between 30 ° and 90 °. Similarly, it can be rotated in a similar motion between 150 ° and 210 °, between 210 ° and 270 °, between 270 ° and 330 °, and between 330 ° and 30 °. Specifically, as shown in FIG. 9, the currents Ia, Ib, and Ic that are sequentially energized according to the rotor rotational position θr are changed to obtain torques Ta, Tb, and Tc, and the rotor is rotated.
A solid line in FIG. 9G shows the motor torque Tm obtained by transferring the generated torques Ta, Tb, and Tc of each tooth. The torque Tm in (G) of FIG. 9 has a characteristic of a figure in which the torque slightly decreases when the current of each phase winding is switched. This partial reduction in torque can be solved by setting the circumferential widths Hm and Hh of the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 of the rotor slightly wider.

Next, the relationship between the circumferential width Hs of the opening of the slot, the circumferential width Hm of the main salient pole magnetic pole 161 of the rotor, the circumferential width Hh of the auxiliary salient pole magnetic pole 162, and the output torque T of the motor will be described. . First, consider the conditions under which a continuous torque can be generated. Consider the conditions under which the motor torque Tm can be generated continuously in the CCW direction at the rotor rotational position of FIG. Consider the case where the main salient pole magnetic pole 161 of the rotor is rotating to CCW and reaches the tooth 117, and the rotation position θr is such that the left end of the main salient pole magnetic pole 161 coincides with the left end of the tooth 117.
The rotational position θr in FIG. 10 is just the rotational position at which the main salient pole magnetic pole 161 can no longer generate torque in the CCW direction. Consider the conditions under which the auxiliary salient pole 162 can generate torque in the CCW direction at this rotational position θr. The condition is that the sum Hg of the gap width between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 and the width of the auxiliary salient pole magnetic pole 162 is set to 60 ° in the circumferential width Hs of the opening of the slot. It is larger than the added width Hf.
Hg> Hf ……………………………………………………………………………… (10)
(360 ° − (Hm + Hh) × 2) / 4 + Hh> 60 ° + Hs (11)

Next, as shown in FIG. 11, a condition is considered in which the main salient pole magnetic pole 161 does not generate torque in the clockwise direction CW when the auxiliary salient pole magnetic pole 162 generates torque in the CCW direction. The condition is that the width Hb of the gap between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 is larger than the width Ht of the stator teeth.
Hb> Ht ……………………………………………………………………………… (12)
(360 ° − (Hm + Hh) × 2) / 4> 60 ° −Hs (13)
For example, when the circumferential width Hs of the slot opening and the circumferential width Hm of the main salient pole magnetic pole 161 are assumed, and the circumferential width Hh of the auxiliary salient pole magnetic pole 162 meeting the conditions is obtained, (11) The width Hh can be expressed by the following equation from the equation and the equation (13).
Hm + 2Hs−60 ° <Hh <−Hm + 2Hs + 60 ° …………………… (14)

In order to simplify the conditions, the circumferential width Hm of the main salient pole magnetic pole 161 is larger than the circumferential width Hh of the auxiliary salient pole magnetic pole 162.
Hm> Hh …………………………………………………………………………… (15)
Further, since the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 drive a rotation angle of at least 60 °, the following condition is satisfied.
Hm + Hh> 60 ° ………………………………………………………………… (16)
An example of specific widths that satisfy the conditions of the expressions (14), (15), and (16) is shown in the table of FIG. The horizontal axis is the circumferential width Hs of the opening of the slot, the vertical axis is the circumferential width Hm of the main salient pole magnetic pole 161, and the minimum value that satisfies the condition of the circumferential width Hh of the auxiliary salient pole magnetic pole 162 in the table It is indicated by Min and the maximum value Max. For example, when the circumferential width Hs = 15 ° of the opening of the slot and the circumferential width Hm = 50 ° of the main salient pole magnetic pole 161, the circumferential width Hh of the auxiliary salient pole magnetic pole 162 satisfying the condition is the minimum value. The maximum value is 20 ° from 20 °.

In FIG. 12, a thick double frame is shown, but the outer thick frame is a range in which the correlation between Hs, Hm, and Hh has a slight margin. The inner thick line frame is a range in which the degree of freedom of selection of each value is further large.
Note that the circumferential width Hh of the auxiliary salient pole magnetic pole 162 in FIG. 12 may be practically used even if the range between the minimum value and the maximum value is slightly different. For example, the auxiliary salient pole magnetic pole 162 and the teeth of the stator may generate attraction force and torque even if the auxiliary salient pole magnetic pole 162 and the teeth 119 are not opposed to each other in the radial direction in FIG. it can. Therefore, continuous torque can be generated even with a value of Hh that is slightly smaller than Hh represented by the equation (11) simply created as a model. Furthermore, since there are many applications that can be used even with a somewhat continuous motor torque, the present invention can be put into practical use even if the expressions (11) and (13) are slightly deviated.

(Example 3)
Next, modifications and improvements of the shape of each part of the motor and the like shown in FIGS. 1 and 6 will be described.
Although the slot shape of the stator has been described as a substantially rectangular shape, a shape 222 having a narrow slot opening may be used as shown in the slot shape of FIG. In particular, in a motor provided with the auxiliary salient magnetic pole 162, if the magnetic pole widths Hm and Hs have a margin, the rotational change rate dφ / dθ of the acting magnetic flux φ is used as the saturation magnetic flux density Bsat of the soft magnetic material (9 ) Formula torque can be obtained. However, when the slot opening is narrow, when a large current is applied to the winding of the slot, a large amount of leakage magnetic flux is generated in the slot opening, causing problems of leakage inductance and magnetic saturation of teeth.

The tips of the teeth of the stator and the tips of the magnetic poles of the rotor face each other at the air gap surfaces, and torque is generated by the mutual attractive force, but each can be deformed into various shapes according to the purpose. The torque ripple or the rotational change rate dFr / dθ of the radial attractive force Fr tends to cause vibration and noise. As a countermeasure, in FIG. 13, the tip of the stator tooth and the tip of the magnetic pole of the rotor are arcuate, but the circumferential end 222 of the tip of the tooth or the end 223 of the tip of the rotor magnetic pole is connected to the stator. By deforming so that the air gap between the rotor and the rotor increases, the rotational change rate dFr / dθ of the suction force Fr in the radial direction can be reduced, and vibration and noise can be reduced.
In order to reduce the magnetic saturation in the vicinity of the teeth and the rotor magnetic poles, the width of each tooth in FIG. 13 is widened from the tip of the teeth to the back yoke, and widened from the tip of the auxiliary salient pole 162 to the back yoke. It is effective to do. Further, as one of the methods for reducing the magnetic saturation of the teeth, a slit-shaped gap 221 may be provided in order to limit the magnetic flux passing through the main salient pole magnetic pole 161. Note that the shape of the slit-shaped gap 221 may be other shapes such as a triangle, a circle, and a square.

  Next, the case where the motor torque T is asymmetric between the CCW and the CW side will be described with reference to FIG. For example, good continuous torque is required in the CCW rotation direction, and discontinuous torque in the CW rotation direction may be used, or only CCW side unidirectional torque is required, or CCW side torque and This is the case when the magnitude of the CW side torque is different. In FIG. 14, the angles Hv and Hw between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 are different. The motor of FIG. 14 is an example of a structure in which torque output to the CCW side is easier. Further, not only the distance between the rotor magnetic poles but also the shape of the rotor magnetic poles can be asymmetrical. Thus, the one-way torque characteristic can be improved by making the shape of the rotor magnetic poles asymmetrical.

Example 4
Next, a method for improving the maximum torque of the motor will be described.
The torque characteristics of the motor of the present invention shown in FIG. 6 are characteristics as shown in FIG. 15, for example. In the region from 0 to A1 where the current I is small, the torque characteristic shown by the equation (7) is shown, and the characteristic like a square function of the current is shown. In the current region from A1 to A2 in FIG. 15 where the magnetic flux density of a part of the motor approaches the saturation magnetic flux density, the torque increases from T1 to T2 as the current increases, as shown by equation (9). The torque characteristic that is a linear function of current is shown. Further, when the current value is larger than A2, a portion where the motor magnetic circuit part is magnetically saturated is generated at a place other than the air gap portion between the stator magnetic pole and the rotor magnetic pole in the motor. As the rate of increase decreases, torque saturation characteristics such as torques T2 to T3 at currents A2 to A3 in FIG. 15 are shown.

The increase in the maximum torque shown here is to improve the torque at T3 to T4 as shown by a partial chain line in the torque characteristics shown in FIG. Technically, the magnetic saturation inside the motor is reduced and the air gap between the stator magnetic pole and the rotor magnetic pole is improved so that a larger magnetic energy can be given.
First, in which part of the motor magnetic saturation is likely to occur will be described with reference to FIGS. 16 and 17. FIG. 16 shows a state in which the main salient pole 161 faces the stator teeth 117 and 11A and the stator teeth are magnetically saturated. The magnetic flux passing through the tooth 117 passes from the tooth 114 through the main salient pole pole 161 of the rotor, the magnetic flux component 241 passing through the tooth 117, the current of the winding 111, and the leakage magnetic flux component 242 passing near the opening of the slot, and , A leakage magnetic flux component 243 passing near the opening of the slot due to the current of the winding 112. In the state of FIG. 16, leakage magnetic flux components 242 and 243 that do not contribute to torque generation are added, so that the magnetic flux passing through the teeth 117 is more likely to be saturated than the main salient pole magnetic pole 161 of the rotor.

The state of FIG. 17 is also a state in which magnetic saturation is caused inside the motor. The magnetic flux passing through the auxiliary salient pole magnetic pole 162 of the rotor passes from the tooth 11C through the auxiliary salient pole magnetic pole 162, the magnetic flux component 252 passing through the tooth 119, and the leakage flux component 253 passing through the vicinity of the opening of the slot due to the winding current. The magnetic flux component 251 leaks from the side surface of the auxiliary salient pole magnetic pole 162 to the teeth 119 due to the currents of the windings 114 and 113. In the state of FIG. 17, leakage flux components 251 and 253 that do not contribute to torque generation are added, and the auxiliary salient pole magnetic pole 162 is narrower than the teeth, so that the auxiliary salient pole magnetic pole 162 is likely to be magnetically saturated.
As described with reference to FIGS. 16 and 17, the places where the magnetic saturation is likely to occur are the teeth of the stator and the auxiliary salient poles 162 of the rotor.

Hereinafter, six methods for improving the maximum torque of the motor will be described. These methods can also be used in combination.
FIG. 18 shows a first method for improving the maximum torque. FIG. 18A is an example of a cross-section AC-AC in FIG. 262 is a tooth portion of the stator core, 263 is a coil end of the winding, and 261 is a rotor core. In FIG. 18B, a soft magnetic body 264 is added in the rotor axial direction in order to reduce the magnetic saturation of the teeth of the stator core. The tooth magnetic path becomes thicker and magnetic saturation is reduced. At this time, the flow of magnetic flux passes not only in the radial direction and the circumferential direction but also in the rotor axial direction. Therefore, when the soft magnetic bodies 262 and 264 are laminated bodies of electromagnetic steel plates, There is a problem that the loss increases. Therefore, in the case of the configuration of FIG. 18B, as shown by a broken line 246 in FIG. 16, a plurality of cuts that are substantially parallel to the tooth portion of the electromagnetic steel sheet are provided, or an electrical insulating film is applied to the soft magnetic powder. Measures to reduce eddy currents are effective, such as using a compressed powder core.

FIG. 19 shows a second method for improving the maximum torque. In FIG. 19A, a soft magnetic body 271 and a permanent magnet indicated by a broken line are added in place of the soft magnetic body 264 shown in FIG. This broken line portion is enlarged and shown in FIG. 272 is a permanent magnet, and 271 is a soft magnetic material. One of the motor control methods shown in FIG. 16 is that the direction of the current in each slot is fixed and the control is performed with direct current. In that case, the direction of the magnetic flux passing through each tooth is one direction, which is characteristic. And it forms in the direction of the permanent magnet 272 so that it may become the direction opposite to the direction of the magnetic flux 275 which passes along the tooth | gear 262, and the magnetic flux 274 is obtained.
FIG. 20 shows the magnetic characteristics of the teeth 262 at this time. The horizontal axis is the magnetic field strength H [A / m], and the vertical axis is the magnetic flux density B [T]. First, when a large current shown in FIG. 16 is applied to the motor without the soft magnetic body 273 and the permanent magnet 272, the magnetic flux density of the teeth 262 changes from zero to the Ba point, and the magnetic flux density becomes B1.

  Next, in the state where the soft magnetic body 273 and the permanent magnet 272 are added, the magnetic flux in the reverse direction passes through the teeth 262, so that it is the Bb point or the Bc point. When a large current shown in FIG. 16 is energized in this state, the magnetic flux density of the teeth 262 changes from zero to the Ba point, and the magnetic flux density is substantially the same value as B1. However, the change in the magnetic flux density is B3 or B4 shown in FIG. 20, which is significantly increased from B1. For example, if B1 and B2 are substantially the same value, twice the magnetic flux can be passed through the tooth 262. Expressed as an increase in torque, this corresponds to increasing the torque at T3 to T4 in FIG. In the case of the configuration of FIG. 19B, it can be said that the time change rate of the rotor axial component of the magnetic flux passing through the teeth 262 is small in principle, but cannot be ignored. Measures to reduce eddy currents are also effective.

  FIG. 21 shows a third method for improving the maximum torque. In FIG. 21A, a soft magnetic body 293 indicated by a broken line and an excitation winding are added in place of the soft magnetic body 273 and the permanent magnet 272 shown in FIG. This broken line portion is enlarged and shown in FIG. Reference numeral 292 denotes a soft magnetic material, and reference numeral 291 denotes an excitation winding. The electromagnetic action is the same as that of the soft magnetic body 273 and the permanent magnet 272 of FIG. 19, and a magnetic flux 294 in the direction opposite to the magnetic flux 275 passing through the rotor can be created. And the maximum torque of a motor can be improved. One feature of the method of FIG. 21 is that the magnetic operating point can be easily changed by changing the magnitude of the current. For example, in the case of a light load, the current can be made zero, unnecessary control is not performed, and a current necessary for the winding 291 can be supplied only when a large torque is required.

  As will be described later, the direction of the current can be reversed. In the method shown in FIG. 19 as well, the magnitude of the magnetic flux 274 can be controlled by inserting and removing the permanent magnet 272 with another actuator, and the direction of the magnet can be reversed. In addition, the methods of FIGS. 19 and 21 can be variously modified, such as a modification in which the positions of the soft magnetic material, the permanent magnet, and the winding are combined with teeth adjacent in the circumferential direction. In addition, regarding the inverter that drives the motor, various inverters can be used, including not only an inverter that allows only one-way current to flow to each winding, but also an inverter that allows a free current to flow.

FIG. 22 shows a fourth method for improving the maximum torque. In this method, permanent magnets 301, 302, 303, 304, 305, and 306 are arranged in the openings of the slots.
As indicated by the magnetic flux indicated by the N and S poles and the arrows 308 and 309, the direction of the permanent magnet is opposite to the magnetomotive force due to the current in each slot, and the leakage magnetic flux in the vicinity of the slot opening is offset and reduced. is doing. For example, as illustrated in FIG. 22, when a positive current flows through the windings 111 and 115 and a negative current flows through the windings 112 and 114, the rotor is not illustrated, but the arrow 307 Assume that the magnetic flux shown is passing. The direction of the magnetic flux 308, 309, 30 A, 30 B of the permanent magnet is opposite to the direction of the magnetic flux 307. As a result, similar to the effect of the structure shown in FIG. 19, the magnetic flux density of the teeth 117 and 11A is negatively biased, so that it cancels out the magnetic flux 307 when a current flows, so the magnetic flux of the teeth 117 and 11A. The density can be reduced and the maximum torque of the motor can be increased.

In particular, this method can be applied even when the length Wt of the motor in the rotor axial direction is increased. Therefore, the method of FIG. 22 is a technology that can be easily adopted even by a motor with a large output.
On the other hand, the methods of FIGS. 18, 19, and 21 tend to gradually decrease the effect ratio as the rotor axial length Wt of the motor increases, which is an effective method for a thin motor. is there.
As can be seen from the slot shape and the arrangement of the permanent magnets in FIG. 22, these permanent magnets can also be used as members for fixing the coils in the slots. Further, it is possible to change the shape such as the vertical and horizontal dimensions of the permanent magnet. Note that changes in characteristics due to the temperature rise of the permanent magnets are required. For example, a member having a large thermal resistance or a liquid cooling pipe for cooling can be disposed between the permanent magnet and the winding in each slot.

  A fifth method for improving the maximum torque is shown in FIGS. FIG. 23A shows a shape in which a soft magnetic body 311 for reducing magnetic saturation is added to the cross-sectional view taken along the cross-section AD-AD in FIG. Reference numeral 313 denotes a rotor core, and both ends of the rotor core 313 are auxiliary salient poles 162. FIG. 23B is a side view of FIG. Like the shape of the auxiliary salient pole magnetic pole 162 in FIG. 23B, the shape of the auxiliary salient pole magnetic pole 162 is trapezoidal from the tip to the rotor center, and the magnetic path is thickened. By thickening the base of the auxiliary salient pole 162, magnetic saturation is reduced and the maximum torque of the motor is improved.

  FIG. 23 shows a sixth method for improving the maximum torque. A soft magnetic material 311 for reducing magnetic saturation is added to the side surface of the auxiliary salient pole magnetic pole 162. The magnetic path of the auxiliary salient pole magnetic pole 162 becomes thick, magnetic saturation is reduced, and the maximum torque of the motor can be improved. The soft magnetic body 311 is disposed in the vicinity of the auxiliary salient pole magnetic pole 162. However, when rotating at high speed, a large centrifugal force is applied and the strength is required. As shown, a soft magnetic material connected in a ring shape in the circumferential direction may be used. It should be noted that the soft magnetic material in the vicinity of the auxiliary salient pole magnetic pole 162 is effective for measures for reducing eddy currents such as the breaks and the dust core. When a soft magnetic material is added, there is a degree of freedom in the direction of the magnetic flux, and it can be driven by a current control method in which the magnetic flux in the magnetic path portion alternates. Although various methods for improving the maximum torque of the motor have been described, by increasing the maximum torque, it is possible to reduce the size of the motor and to achieve a reduction in cost.

(Example 5)
Next, the motor structure and driving method for realizing constant output control of the motor of the present invention will be described. The motor of the present invention shown in FIGS. 1 and 6 etc. basically generates the magnetic flux of the motor by current, has high controllability of the magnetic flux, and can rotate at high speed. The fact that the rotor has a robust structure is also fundamentally important for realizing high-speed rotation. The motor of the present invention can realize a high output by increasing the rotation speed and can realize a motor having a high output density. However, in order to obtain high torque at high speed rotation, there is also a technical problem in reducing torque ripple at high speed rotation. In driving an automobile, a large torque is required for moving up and down a slope at a low speed, and a high-speed rotation is required even if the torque is slightly small on a highway, and so-called constant output characteristics are often required.

First, a basic high-speed driving method and problems of the motor of the present invention will be described. The motor of FIG. 6 is considered on the characteristics of the rotational speed ωr and torque T of FIG. Now, let Tc be the continuous rated torque when a current In that is capable of continuous energization is energized. Let the current voltage VM of the inverter that drives the motor be the rotation speed at which the motor voltage becomes VM when the current In is energized is the base rotation speed ωbr. In FIG. 26, the ideal constant output characteristic is a characteristic in which the torque is directed toward P4 in a shape in which the torque is inversely proportional to the rotational speed from the operating point of P3. In the case of torque Tb = 2Tc which is twice the continuous rated torque, the characteristic is from P1 to P2.
In the rotational speed region larger than the base rotational speed ωbr, the motor voltage V is proportional to the time change rate of the interlinkage magnetic flux φ and the number of turns Nw, and therefore V = Nw × dΦ / dt = Nw × dΦ / dθ × dθ / dt. If this voltage V exceeds the power supply voltage VM, control becomes difficult. Here, dΦ / dθ is the rotation change rate of the flux linkage φ, and dθ / dt = ωr is the rotational speed of the rotor.
In the constant output control often used, the flux linkage φ is controlled to be small, and constant output control is realized by so-called field weakening control or field weakening control. In the motor of the present invention, conversely, one method is to increase the flux linkage φ, narrow the change range of the magnetic flux density using magnetic saturation, and reduce the rotational change rate dΦ / dθ of the flux linkage φ. realizable.

  Specifically, the method of driving at a high rotational speed equal to or higher than the base rotational speed ωbr is performed by reducing the current value, but the output torque naturally decreases. In a current region that is large to some extent, even if the current is reduced, the magnetic flux φ does not decrease in proportion to the current, so that the amount of current reduction increases. Further, by changing the phase of the current, the current can be driven with less reduction while utilizing the leakage inductance of the motor. As a result, the characteristic indicated by the broken line from the operating point P3 to P5 in FIG. The realistic characteristic of the torque Tb = 2Tc that is twice the continuous rated torque is also a characteristic from the operating point P6 to P5. In the torque region indicated by these broken lines, the torque ripple becomes large, but since it is a high frequency torque ripple, there are many applications that can be used without problems. However, there is a problem that the torque indicated by P5 and the broken line is smaller than the torque value of the ideal constant output characteristic, and a problem that torque ripple increases.

  In contrast to the ideal characteristic indicated by the solid line, the cause of the decrease in torque as indicated by the broken line is that the motor is operated in a region where the reduction of the magnetic flux in the motor and the generation of torque cannot be achieved at the same time. For example, an ideal constant output characteristic can be obtained if the time change rate of the magnetic flux φ is the same at the operating point P6 at the operating point P3, the time rate of change of the magnetic flux φ is the same, and the torque T can be output to 1/2 of Tc. It will be. At this time, since the rotational speed is 2 × ωbr at the operating point P6, it means that the induced voltage of the motor becomes the power supply voltage VM when the magnetic flux φ is reduced to ½ of that at the operating point P3. . That is, if the rate of change in rotation of the magnetic flux can be varied, constant output characteristics can be obtained. As described above, the control corresponds to the field weakening control for controlling the field current component in the synchronous motor or the field weakening control for actively reducing the permanent magnet magnetic flux component by the current.

  Next, a method for reducing the rotational change rate of the motor magnetic flux in the motor of the present invention will be described. FIG. 24 has the same configuration as FIG. 21B, but the current direction of the excitation winding 291 is reversed and the direction of the magnetic flux 294 is reversed. The direction of the magnetic flux 294 is the same direction as the magnetic flux 275 that passes through the rotor. Now, the magnetic flux density BX of the tooth 262 is considered on the coordinates of the magnetic field strength H and the magnetic flux density B in FIG. First, when the motor current is zero and the current flows to the excitation winding 291 and the magnetic flux 294 is generated, the magnetic flux density BX of the teeth 262 is at the operating point Bb in FIG. In this state, it is assumed that the motor current is supplied and the magnetic flux 261 is induced to reach the operating point Ba in FIG. The magnetic flux density B1 at the operating point Ba is close to 2 Tesla, and is an operating point that is almost close to the saturation magnetic flux density.

  When the current of each phase is controlled in this state and the motor rotates, the magnetic flux density BX of the teeth 262 reciprocates between the points Bb and Ba, and the magnetic flux density BX reciprocates from B5 to B1. . Accordingly, the voltage induced in the winding of the motor is proportional to the time change rate of the interlinkage magnetic flux, and thus becomes a voltage proportional to the magnetic flux density B6 = B1-B5. As a result, the voltage induced in the motor winding is reduced to almost ½ compared to 4 when the value of the excitation winding 291 is zero. In the synchronous motor, field weakening control can be performed, and the same effect as when performing constant output control can be obtained. The point Bb is the current of the excitation winding 291 and can be controlled freely from 0 Tesla to nearly 2 Tesla, so that it can be controlled effectively. In addition, the magnetic resistance of the magnetic path through which the magnetic flux 294 passes is small because it does not include a large gap or the like, and can be controlled with a relatively small current. In addition, when the motor current is driven in one direction and operates at the point Bb by the current of the excitation winding 291, the hysteresis loop of the soft magnetic material of the tooth is small, eddy current loss and hysteresis loss are also small, and iron loss is reduced. You can also do it.

  The configuration in FIG. 24 is the same as the configuration in FIG. 21 except that the direction of the magnetic flux 261 caused by the motor current and the direction of the magnetic flux 292 caused by the excitation current of the winding 291 are different. Accordingly, by variably controlling the relative current direction of the motor current and the exciting current of the winding 291, the effect of increasing the maximum torque of the motor and the effect of improving the constant output characteristics can be obtained with the same configuration. .

  Next, another method for reducing the rotation change rate of the motor magnetic flux and improving the constant output characteristics will be described. 25, the current direction of the motor is reversed in FIG. 19, and the direction of the magnetic flux 681 is reversed. The magnetic flux 274 and the magnetic flux 681 excited by the permanent magnet 272 are oriented in the same direction at the tooth 262, and the tooth is likely to be magnetically saturated. In FIG. 27, the magnetic flux density BX of the tooth 262 is set to the operating point Bf, and when the motor current is applied, the magnetic flux density BX moves to the operating point Be, and the fluctuation of the magnetic flux density is B8 = B2-B7. As a result, the same effect as the excitation by the winding current in FIG. 24 can be obtained, and the characteristics of the constant output control can be improved.

Next, another method for reducing the rotation change rate of the motor magnetic flux and improving the constant output characteristics will be described. In FIG. 22, the magnetic flux direction of the permanent magnet and the magnetic flux excited by the drive current are set in the same direction, the same effect as the excitation by the winding current in FIG. 24 can be obtained, and the characteristics of constant output control can be improved. .
The configuration of FIG. 25 is the same as the configuration of FIG. 19 except that the direction of the magnetic flux 681 caused by the motor current and the magnetic flux 274 caused by the exciting current of the winding 291 are different. Therefore, by variably controlling the direction of the motor current, the effect of increasing the maximum torque of the motor and the effect of improving the constant output characteristics can be obtained with the same configuration. Normally, permanent magnets are often used while being fixed in the motor. However, the permanent magnets can be moved in and out of their magnetic paths by other actuators to change the effect of the permanent magnets, or even the direction of the magnets can be reversed. Is possible. In particular, when used for driving an electric vehicle or the like, it is not necessary to change the operation mode so quickly, and it is realistic to move the permanent magnet with another actuator.

  In addition, as one method of reversing the current direction when generating a large torque and when performing constant output characteristics, there is a method of switching the connection between each winding and the inverter in the reverse direction using an electromagnetic contactor or the like. In an electric vehicle or the like, a large torque mode may be set for low-speed rotation, and a constant output control mode may be set for high-speed rotation. The winding switching time in the magnetic contactor can be technically switched in a short time of about 0.1 seconds, and can be speeded up to such an extent that it does not interfere with operation. In addition, a method of switching the windings in conjunction with a mechanism that is changed by the number of revolutions, such as a shift lever of an automobile transmission, is also possible.

(Example 6)
Next, FIG. 28 shows a method for improving the motor of the present invention shown in FIG. The motor of FIG. 28 is an example of a 4-pole motor.
The slots in which the A-phase windings are wound are 321, 324, 327 and 32 A. The slots around which the B-phase windings are wound are 323, 326, 329, and 32C. Slots around which the C-phase winding is wound are 325, 328, 32B, and 322.
The winding of each slot is omitted and not shown. 32L is a rotor, 32D, 32E, 32F, and 32G are slits, which are gaps obtained by punching out a part of the electromagnetic steel sheet by press punching. It is also possible to fill a member having a large magnetic resistance. In order to maintain the centrifugal force of the outer periphery of the rotor, a connecting portion such as 32H may be added.

Moreover, in order to ensure the strength of the rotor, the outer periphery of each magnetic pole has a shape in which the outer periphery is connected as indicated by 32J. As shown in the portion of 32K, the slit shape is such that when the magnetic flux passes from the tip of the rotor magnetic pole to the inside of the rotor, the width of the magnetic path through which the magnetic flux from the tip of the rotor magnetic pole to the inside of the rotor does not decrease. The tip of the is gradually narrowed. The slit increases the magnetic resistance before and after the slit, acts to separate the magnetic flux, and prevents the magnetic flux from being biased near the tip of the rotor magnetic pole, thereby making the torque uniform during rotation and increasing the torque. ing. In part, it also has the effect of reducing magnetic saturation.
29 shows a four-pole motor. In FIG. 28, each rotor salient pole has two layers of slits, whereas slits 632, 633, 634, 635, 636, 637, 638, and 639 are provided on each rotor salient pole. There are 4 layers. Although it has an asymmetric structure, it can also be made of three layers. A connecting portion 631 corresponding to centrifugal force can also be added. The magnetic path of the rotor salient pole indicated by the arrow 63A has a shape in which both ends of the slit are sharp so that the magnetic flux can pass easily.

Next, FIG. 30 shows a method for improving the motor of the present invention shown in FIG. This is a 4-pole motor. The rotor 620 includes a main salient pole magnetic pole 621 and an auxiliary salient pole magnetic pole 622. Similarly to FIG. 28, slits 32D, 32E, 32F, and 32G are added.
Next, an example in which a permanent magnet is added to the motor of FIG. 30 is shown in FIG. Reference numerals 511, 512, 513, and 514 denote permanent magnets. Permanent magnets can be added to the motors of FIGS. The distribution of magnetic flux in the rotor can be changed by these permanent magnets, and the torque can be improved. Further, the methods shown in FIGS. 18 to 31 can be used in combination.

(Example 7)
Next, FIG. 32 shows a motor having the same function as the motor shown in FIG. 6 and having a different winding configuration. This motor is a two-pole motor, and the winding is short-pitch winding and concentrated winding. The motor shown in FIG. 32 has a structure of full-pitch winding and concentrated winding, and the main aim is to apply a magnetomotive force to a specific tooth as shown in FIG. A large suction force is generated between the rotor salient poles. The motor shown in FIG. 32 concentrates windings indicated by 612 and 613 around the teeth 117 in order to generate a suction force to the teeth 117. Similarly, windings 614 and 615 are wound around the tooth 118, windings 616 and 617 are wound around the tooth 119, windings 618 and 619 are wound around the tooth 11A, and windings 61A and 61B are wound around the tooth 11B. A wire is wound and the windings of 61C and 611 are wound around the tooth 11C.

  As the operation of the motor, it may be energized so as to be the same as the current shown in FIG. The windings 612 and 613 and the windings 618 and 619 are connected in anti-series, the windings 616 and 617 and the windings 61C and 611 are connected in anti-series, the windings 61A and 61B, the windings 614 and 615, Are connected in reverse series to form three sets of three-phase windings. Each winding of the motor shown in FIG. 32 has a winding cross-sectional area in the slot that is halved compared to each winding shown in FIG. 6, so that the winding resistance of that portion increases and the copper loss increases. It will be. Since two types of windings are inserted into the same slot, a wasteful space is likely to be generated, and there is also a problem of the winding space factor. From a different point of view, the winding pitch is a short-pitch winding with an electrical angle of 60 ° or less, and the winding coefficient is ½ or less, so that the efficiency is low.

  However, the winding length of the coil end portion of the motor shown in FIG. 32 is about 1/3 shorter than that of the motor shown in FIG. 6 because the electrical angle is 180 °. Therefore, in the case of a thin and flat motor shape with a high winding ratio in the coil end portion, the motor configuration of FIG. 32 is advantageous in terms of winding resistance. In addition, the motor of FIG. 32 has an advantage that the length of the coil end in the rotor axial direction can be reduced to about ½ compared to the motor of FIG. In the case of a motor for high-speed rotation, the number of poles of the motor becomes small due to the limit of the driving frequency and the coil end becomes long. Also in the case of such a high-speed rotary motor, the concentrated winding configuration as shown in FIG. 32 is advantageous. Note that the relationship between the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 of the rotor is as described above, and is the relationship of each angle as shown in FIG.

  Specifically, the configuration of the motor shown in FIG. 32 is advantageous in the configuration of the motor. Now, let Lp be the pitch between adjacent slots, t be the rotor axial thickness of the stator core, and let k be the resistance per unit length of the winding. The resistance of the full-pitch winding is k × (2t + 2Lp), and the resistance of the short-pitch winding and the concentrated winding is k × 2 × (2t + 2 × 1/3 × Lp). Here, it is considered that the winding thickness of the short-pitch winding and concentrated winding of FIG. 32 is halved. And, the condition that the resistance value of the short-pitch winding and the concentrated winding is smaller is k × 2 × (2t + 2 × 1/3 × Lp) <k × (2t + 2Lp), and t <1/3 × Lp. is there. Therefore, in the case of a thin motor in which the rotor axial thickness t of the stator core is 1/3 or less of the pitch Lp between slots, the motor having the configuration shown in FIG. 32 is advantageous in terms of simple calculation of the joule loss of the winding. . In general, short-pitch winding and concentrated winding are easier to manufacture, and the effect of shortening the coil end length is taken into consideration, so that t <2/3 × Lp. In this case, in the case of a thin motor in which the rotor axial thickness t of the stator core is 2/3 or less of the pitch Lp between the slots, the motor having the configuration shown in FIG. It can be said that it is advantageous in terms of simple calculation.

(Example 8)
Next, the relationship among the tooth shape, slot shape, current distribution in the slot, magnetic characteristics and magnetic flux distribution of the tooth, magnetic saturation, and maximum torque of the motor will be described.
FIG. 33 is a sectional view showing a part of the stator. 331 is the back yoke, 332 is the tooth, WTB is the circumferential width of the tooth tip, WTA is the circumferential width of the root of the tooth, WSB is the circumferential width of the slot opening, 333, 334, and 335 are each slot Winding.
Assuming that a positive current is passed through the winding 334 and a negative current is passed through the winding 335 and that the rotor magnetic pole exists near the teeth 332, the magnetic flux 336 passes from the rotor side to the back yoke side of the teeth 332. At the same time, a leakage flux 337 is generated around the winding 336, and a leakage flux 338 is generated around the winding 335. When both the currents are large, the magnetic flux density on the back yoke side of the teeth 332 increases, causing magnetic saturation and reducing the maximum torque of the motor. In the following, four methods for solving this problem are shown.

In the first method, as shown in FIG. 34, the width WTA of the tooth root is made larger than the width WTB of the tooth tip, and the tooth thickness is increased from the tip to the back yoke side. Magnetic saturation at the back of the teeth can be reduced. At this time, the center of the current distribution moves from the back yoke side of the slot to the opening side of the slot, as can be seen by comparing the trapezoidal slot of FIG. 33 and the rectangular slot of FIG. You can see that
As shown in FIG. 34, the second method uses windings of different thicknesses in the same slot, narrows the winding on the opening side of the slot, and increases the current density on the slot opening side. Leakage magnetic fluxes 347 and 348 are reduced.
As shown in FIG. 35, the third method is a method in which the circumferential width WSB of the slot opening is increased and the leakage magnetic fluxes 357 and 358 are reduced.

  The fourth method is to change all or part of the teeth 352 to a material having a high saturation magnetic flux density. For example, as shown in the figure, when the shape of the tooth tip 359 is narrowed and the magnetic resistance is increased by reducing the tooth tip to reduce the leakage magnetic fluxes 357 and 358, the magnetic flux 356 from the rotor side cannot be sufficiently passed. If the soft magnetic material 359 is changed to a material such as permendur, which has a higher saturation magnetic flux density than a normally used electromagnetic steel sheet, the magnetic flux 356 from the rotor side can be sufficiently passed. Such a material having a high saturation magnetic flux density is often expensive, and can be partially used where it is highly necessary. With respect to the shape, it is possible to reduce the magnetic saturation on the back yoke side of the tooth or to fix the tooth tip portion 35A to the stator core by changing the shape to 35A or the like.

Example 9
Next, the problem of winding will be verified. After that, a method for solving the winding problem will be described. 66, 67, and 68 as conventional examples, the problem is that the arrangement of each winding is complicated and difficult to manufacture, the problem that the space factor of the winding is reduced, and the coil end portion due to the intersection of windings The problem of the lengthening in the axial direction was shown.
FIG. 36 is a diagram modeling the shape of an example of a stator of the motor of the present invention having four poles as viewed from the rotor axial direction. The A-phase windings are 361, 362, 367, and 368. The B phase windings are 363, 364, 369, and 36A. The C-phase windings are 365, 366, 36B, and 36C. The windings in the same slot are divided in both directions in the circumferential direction so that the winding bias at the coil end portion is small. FIG. 37 is a longitudinal sectional view of the vicinity of the coil end portion of the motor of FIG. Reference numeral 381 denotes a stator core in which electromagnetic steel plates are laminated in the rotor axial direction. Reference numeral 384 denotes a winding in the slot, 385 denotes a coil end, and LCE1 denotes a coil end length which is a protruding amount from the stator core 381.

Reference numerals 386 and 387 denote insulating papers for maintaining insulation between the stator core 381 and the windings in the vicinity of the slots. The insulating paper 387 normally protrudes about 10 mm from the end of the stator core 381 in order to ensure insulation. Reference numeral 382 denotes a rotor, and reference numeral 383 denotes an air gap portion between the stator core 381 and the rotor 382. The alternate long and short dash line is the rotation center of the rotor.
With such a configuration, a 2-pole motor and a 6-pole motor can be wound in the same manner, and the windings at each part of the coil end are almost uniformly distributed over the entire circumference. However, a plurality of windings overlap the coil end portion, resulting in a complicated arrangement, and there is a problem of productivity. There is also a problem that the winding space factor in the slot is lowered due to difficulty in winding. The length of the coil end in the rotor axial direction can be shortened to some extent by molding with a mold or the like because the windings of the coil end portion are dispersed. However, there is a limit to molding, and there is a problem that the coil end is long.

  Next, a winding winding method and a winding arrangement method for reducing these problems will be described with reference to FIG. FIG. 38 is a diagram showing a model of the winding slot position and the winding arrangement of the coil end portion as seen from the rotor axial direction of the stator example of the motor of the present invention having 4 poles and 12 slots. A positive A-phase winding is wound around the slot 371 and a negative A-phase winding is wound around the slot 374 with the winding 37D. Then, a positive A-phase winding is wound around the slot 377 and a negative A-phase winding is wound around the slot 37A with the winding 37G. Similarly, a positive B-phase winding is wound around the slot 373 and a negative B-phase winding is wound around the slot 376 with the winding 37E. Then, a positive B-phase winding is wound around the slot 379 and a negative B-phase winding is wound around the slot 37C with the winding 37H. Similarly, a positive C-phase winding is wound around the slot 375 and a negative C-phase winding is wound around the slot 378 with the winding 37F. Then, a positive C-phase winding is wound around the slot 37B and a negative C-phase winding is wound around the slot 372 with the winding 37J.

In the winding order of the windings in FIG. 38, the windings 37D, 37F, and 37H arranged on the outer diameter side are wound before the windings 37E, 37G, and 37J. The reason is that, for example, the winding 37E covers the end surfaces of the slots 374 and 375 in the rotor axial direction, so that the windings 37D and 37F must be wound before the winding 37E. Therefore, for example, if the windings 37D and 37F are wound, the winding 37E can be wound before the winding 37H.
As shown in FIG. 38, the number of poles of the motor that can be wound with the concentrated winding wound on the outer diameter side and the inner diameter side is an integer multiple of four. In this case, since the number of windings is small and mutual physical interference is small, the windings can be wound efficiently and with high space.

  FIG. 39 shows an example of a stator of a 6-pole motor, in which the number of poles of the motor is not an integer multiple of 4. The A-phase windings are the winding 73K wound from the slots 731 to 734, the winding 73N wound from the slots 737 to 73A, and the winding 73R wound from the slots 73D to 73G. The B-phase windings are a winding 73L wound from slots 733 to 736, a winding 73P wound from slots 739 to 73C, and a winding 73S wound from slots 73F to 73J. The C-phase windings are a winding 73M wound from slots 735 to 738, a winding 73Q wound from slots 73B to 73E, and a winding 73T wound from slots 73H to 732. Among these windings, the winding 73T has an irregular shape and arrangement compared to other windings. Thus, if it is made partially irregular, similar windings can be obtained with 6 poles, 10 poles, 14 poles, 18 poles, and the like.

  Next, the winding shape and the core shape in the vicinity of the coil end portion of the motor shown in FIG. 38 will be described with reference to FIGS. 40, 41, 42, and 43. FIG. 40 is a cross-sectional view of the cross-section AF-AF of FIG. 38 on the slot 375 side. The coil end 385 of FIG. 37 is indicated by a broken line for comparison. 391 is a stator core, 396, 397, and 398 are insulating papers, 394 is a winding wound around a slot 375, and 395 is a coil end portion thereof. From the middle of the slot toward the rotor axial end surface of the stator core, it is gradually bent toward the outer diameter side and connected to the coil end 395. Reference numeral 399 denotes a winding 37E, which is located in the circumferential direction. As a result, it can be seen that the length LCE2 of the coil end in FIG. 40 is significantly shortened compared to the length LCE1 in the rotor axial direction of the conventional coil end.

  FIG. 41 is a sectional view of the section AE-AE in FIG. 38 on the slot 376 side. 40B is a stator core, 403, 404 and 405 are insulating papers, 406 is a winding wound around a slot 376, and 402 is a coil end portion thereof. The shape of the windings 406 and 402 is shown in FIG. 42 in which the arc shape is linearly expanded when viewed from the stator side. The horizontal axis in FIG. 42 is the circumferential direction, and the vertical direction is the rotor axial direction. Reference numerals 411, 412, 413, 414, and 415 denote inner peripheral side shapes of the respective teeth. The windings 416 and 417 are the winding 37E of FIG. 38 and the windings 406 and 402 of FIG. When the winding 416 reaches the portion 418 from the middle of the slot, it is bent in the circumferential direction and connected to the coil end portion. The same applies to the bent portion 419. Note that the winding 401 in FIG. 41 is the winding 37F in FIG. 38, and is located in the circumferential direction.

In the conventional motor, the shape of the facing surface of each tooth to the rotor is a quadrangular shape as indicated by a broken line 41A in FIG. 413, 414, 415. Considering the winding arrangement shown in FIG. 38, since the windings are not arranged at the corners of all teeth, the corners of all teeth are curved like the corners of each tooth in FIG. It is not necessary to have a shape, but the same tooth shape is used in order to maintain the electromagnetic balance of the entire motor.
As described above, since the corner shape of each tooth is bent into an arc shape without difficulty, the amount of protrusion of the coil end 402 shown in FIG. 41 in the rotor axial direction is short, and the size of the LCE 2 is suppressed. Thus, by devising the slot shape of the stator core in the vicinity of the coil end, the length of the coil end in the rotor axial direction can be shortened, and the space factor can also be improved.

  As described above, the shape 39B of the stator core where the winding 394 in FIG. 40 is bent is a smooth curve or a circular arc. This is a gentle curve shape so that the windings 394 and 395 do not have an unreasonable shape such as a sharp bend. Similarly, the shape of the insulating paper 397 is a gentle curved shape. However, since it is necessary to manufacture various types of electromagnetic steel sheets in order to manufacture the stator core 391 formed by laminating electromagnetic steel sheets, there is an equipment burden such as a mold for punching the electromagnetic steel sheets having these shapes. growing. In view of this, it is possible to consider a method in which the winding can be gently bent and the burden on the equipment such as the mold is not excessive. Specifically, these curvilinear shapes are replaced with a two-step, multi-step shape. For example, it is an example of a three-stage shape such as a slot shape 407 indicated by a broken line of the stator core 40B in FIG.

The stator core is often composed of laminated electromagnetic steel sheets, and the smooth shape change of the slots increases the cost burden of a die for punching the electromagnetic steel sheets. If it is a shape like the slot shape 407, since it can manufacture by laminating | stacking three types of electromagnetic steel plates, it is possible to reduce the cost of production equipment. However, in the case of the stepped shape, a space portion at the corner is formed, so that the space factor between the surrounding electromagnetic steel sheet and the winding is slightly lowered.
On the other hand, in the shape of the curved portion of the electromagnetic steel sheet shown in FIGS. 40 and 41, the punched shape of each laminated electromagnetic steel sheet is further formed into a tapered shape, and the finished shape obtained by laminating the electromagnetic steel sheets is an electromagnetic steel sheet. The punched portion of each electromagnetic steel sheet can be formed to have a curved shape or a linear shape shown in FIG.

Further, since the circumferential magnetic path of the back yoke portion in the vicinity indicated by the broken line 39A (see FIG. 40) is decreased, as shown in FIG. An annular soft magnetic body 408 is added. The soft magnetic body 408 can secure the circumferential magnetic path of the back yoke portion without increasing the length in the rotor axial direction of the motor.
In FIG. 40, the soft magnetic body 408 is omitted. As a specific configuration example of the soft magnetic body 408, as indicated by three broken lines 39D, an electromagnetic steel plate near the rotor axial end of the stator core 40B can be bent in the rotor axial direction. Even when the magnetic flux passes, more magnetic flux can be changed in the same steel plate, so that the magnetic flux change in the surface direction of the magnetic steel plate is small, and an excessive eddy current does not flow.

  From the viewpoint of eddy currents, 39A in FIG. 40, 39D and 408 in FIG. 41, 264 in FIG. 18B and nearby electrical steel sheets, 271, 273 in FIG. 19 and nearby electrical steel sheets, 292 in FIG. 23, 162 and 311 in FIG. 23, nearby electromagnetic steel sheets in FIG. 24, nearby electromagnetic steel sheets in FIG. 24, 271, 273 in FIG. 25 and nearby electromagnetic steel sheets, and the like, as in 39D in FIG. It is also possible to have a bent electromagnetic steel sheet configuration. The dust core has a small eddy current in any direction through the magnetic flux. Moreover, it can improve so that an eddy current may not become excessive, even if the magnetic flux which passes in the surface direction of an electromagnetic steel plate changes with time by making a slit to radial direction in each electromagnetic steel plate. As described above, when the magnetic flux passing in the surface direction of the electrical steel sheet changes with time, some countermeasure against eddy current is required.

Next, improvement of the insulating paper 387 and 386 shown in FIG. 37 will be described. In the case of FIG. 37, the cross-sectional shape of the slot is the same from the part of the stator core 381 to the other part in the rotor axial direction. Therefore, the terminal treatment of the insulating paper 386 and 387 often protrudes about 10 mm from the end of the stator core 381 in the rotor axial direction. The reason for this is that if the end of the stator core 381 is extended along the end surface of the rotor core along the end surface, the insulation distance between the insulating paper cut and the corner 388 of the stator core is shortened, or If the insulating paper is divided in the vicinity, there is a problem that the effective sectional area of the slot decreases and the winding space factor decreases. In the configuration of FIG. 40, as the slot shape approaches the end in the rotor axial direction, the slot cross-sectional area increases, and the insulating papers 398 and 397 are overlapped so as to secure an insulation distance from the stator core 391 while ensuring both insulation. Paper can be connected. Since the insulating paper 397 can be disposed along the slot shape 39B of the stator core 391, the length LCE2 of the coil end 395 in the rotor axial direction can be shortened.
Similarly, in the configuration of FIG. 41, the slot cross-sectional area is increased, and both insulating papers are overlapped along the slot shape 407 while being overlapped like the insulating papers 405 and 404 to ensure an insulating distance from the stator core 40B. It can be connected.

It is possible to improve the winding method and the space factor. The windings of the conventional motor as shown in FIG. 68 are difficult to manufacture because the windings intersect each other. A coil is first manufactured by a method called a so-called inserter system, and then the coil is inserted into a slot by a mold, and finally a winding is formed. With this method, production facilities become large and there is a problem in productivity. The coil end becomes long and there is a problem of the winding space factor.
In the configuration shown in FIGS. 38, 40, 41, 42, and 43 of the present invention, since the windings do not cross each other, for example, by arranging and using a simple winding guide jig, the windings It is also possible to wind while applying tension. This can be easily understood from the winding arrangement of FIG. 38 and the coil shapes of FIGS.

Specifically, first, the windings 37D, 37F, and 37H in FIG. 38 are formed into shapes like the windings 394 and 395 in FIG. 40 and the winding 401 in FIG. If the winding guide jig is formed along the substantially loop-shaped winding, the winding can be wound while tension is applied.
Next, the windings 37E, 37G, and 37J shown in FIG. 38 are formed into shapes like the winding 399 shown in FIG. 40, the winding 402 shown in FIG. 41, and the winding 417 shown in FIG. If the winding guide jig is formed along the substantially loop-shaped winding, the winding can be wound while tension is applied.
In addition, if a multi-pole motor is used to manufacture a coil outside the stator core and insert it into the slot, the radial difference between the inserted coils is small, and the coil can be easily inserted. If the coil is easy to manufacture and assemble, generally the space factor of the winding can be improved.

40 and 41, the coil end part is fixed and radiated by placing the motor flange part in close contact with the coil end so that the coil end is fixed by the flange part and at the same time the coil end part is heated. Can be effectively performed by transmitting to the flange.
The winding method shown in FIG. 38 is high in productivity because each winding has little physical interference with other windings and winding is easy, and the space factor of the winding is high. It is also possible to improve. As shown in FIGS. 40 and 41, the arrangement and configuration of the windings can be facilitated by improving the slot shape, and the protrusion LCE2 in the rotor axial direction of the coil end portions 395, 399, 401, 402 can be shortened. The effective utilization rate in the vicinity of the coil end, which has been a problem with conventional motors, is greatly improved. As a result, it can be said that the motor has been downsized.

(Example 10)
Next, the structure in which the stator core of the motor of the present invention is divided will be described with reference to FIGS. 44, 45, 46 and 47. FIG. FIG. 44 is a diagram modeling each winding when the stator of the 8-pole motor of the present invention is viewed from the rotor axial direction. 80E is an N pole permanent magnet of an 8-pole rotor, and 80F is an S pole permanent magnet. It consists of a stator core joined at four points of division points 42E, 42F, 42G, and 42H. Although the shape of these dividing points is indicated by double lines, it can be modified into various shapes. Since these divided stators are divided into 360 degrees in electrical angle, the windings of each phase can be wound around the same divided core, and the respective divided stators can be manufactured independently. The A-phase winding is wound from the slot 421 to the slot 424. The B-phase winding is wound from slot 423 to slot 426. The C-phase winding is wound from slot 425 to slot 422.

For example, when winding a winding around one of the four divided stator cores, since the arc shape is 90 °, the opening side of the slot is open, which is significantly easier than winding a winding around an annular stator core. There is a degree of freedom. Therefore, the winding can be directly wound by a winding machine, and the winding space factor in the slot can be improved.
The same effect can be obtained by dividing the stator core by an integer multiple of an electrical angle such as 360 ° or 720 ° in electrical angle. Also, for example, in the case of a stator core divided at 720 °, the shape of the stator or rotor can be deformed at a cycle of 720 °, and two in-phase rotor magnetic poles are 360 ° to reduce torque ripple. It is possible to adopt a method of canceling torque ripple with a slightly different pitch. In addition, this description is a condition necessary for making the circulating current in the stator core by a welding site described later zero.

  An example of a specific shape of the dividing points 42E, 42F, 42G, and 42H is shown in FIG. 431 is a side view of the stator core, and adjacent split cores are magnetically joined at a portion 434. In the portion 434, the electromagnetic steel plates are alternately combined in an uneven shape every one sheet or every several sheets. In this manner, the split cores 432, 438, 439, and 433 are joined by the joint portions 434, 435, 436, and 437 as shown in the lower cross section of the stator core in FIG. A partial enlarged view of the joint portion 434 is shown in FIG. For example, although the electromagnetic steel plates 440 and 441 are abutted, there is a gap Lgt when viewed microscopically. This gap Lgt makes it difficult for magnetic flux passing through the electromagnetic steel sheet in the circumferential direction. On the other hand, although the electromagnetic steel sheets are laminated in the rotor axial direction, a gap Lgs between the electromagnetic steel sheets is slightly generated. Here, since the opposing areas in the stacking direction are wide, the magnetic flux in the circumferential direction passes as shown by the arrows. In the portion 444 that is abutted in the circumferential direction, the magnetic flux in the circumferential direction is small, and a large amount of magnetic flux passes toward the direction 443 that easily passes in the circumferential direction. In this way, the difficulty of following the magnetic flux at the joint portion of the stator core can be reduced.

  For the mechanical fixing of each divided core, various methods such as pressure welding with a bolt, adhesive fixing with an adhesive, gripping with a casing, welding fixing with laser welding or the like are possible. FIG. 47 shows an example in which portions 451, 452, 453, 454, and 455 are fixed to each other in the stacking direction by laser welding. Since welding is an electrical connection between electromagnetic steel plates, heat is generated when a circulating current is generated in the magnetic body, and thus loss is increased. When only the outermost peripheral part and the innermost peripheral part of the motor are made conductive by welding or the like, there is no risk of circulating current because there is no magnetic flux linked to the path through which an electric current flows.

(Example 11)
Next, a configuration in which two sets of the motor of the present invention are arranged on the outer diameter side and the inner diameter side is shown in FIG. 48 and described. 48 shows an 8-pole motor. The motor shown in FIG. 1 has 8 poles, the rotor R1 is arranged on the outermost diameter side, the stator S1 corresponding to the rotor R1 is arranged on the inner side, and the inner diameter of the stator S1 is shown. The stator S2 is arranged on the side, and the rotor R2 that acts on the stator S2 is arranged on the inner diameter side of the stator S2. 466, 467, 468, 469, 46A are salient poles of the rotor R1. Reference numerals 461, 462, 463, 464, and 465 are salient poles of the rotor R2. The present invention can also be applied to the motor shown in FIG. A so-called inner rotor motor and outer rotor motor are arranged on the same surface. It can be said that another motor is arranged on the inner diameter side of the motor and the space is effectively used. As a combination of the arrangement of the stator and the rotor, the stator S3 can be arranged on the outermost side, the stator S4 can be arranged on the innermost side, and the rotors R3 and R4 can be arranged between S3 and S4.

In the configuration of FIG. 48, windings such as windings 46B, 46C, 46D, 46E, 46F, 46G, 46H, 46J, 46K, 46L, 46M, 46N, and 46Q are wound from the slot of stator S1 to the slot of stator S2. This makes it possible to simplify winding winding. In addition, since this winding structure is a space without using the side surface of the teeth in the rotor axial direction, it is easy to apply a magnetic saturation reduction measure as shown in FIG. 18, FIG. 19, and FIG. . Moreover, it is easy to apply a countermeasure for constant output control using magnetic saturation as shown in FIGS.
In addition, since the electromagnetic conditions of the motor on the outer diameter side and the motor on the inner diameter side are different, if the optimization of both motors is attempted, the currents of both motors have different values, and there is a problem that inconvenience occurs in the windings. . In order to solve this problem, windings 46X, 46R, 46S, 46T, 46U, 46V, and 46W as shown in FIG. 48 can be added and wound. By optimizing both motors in this way, higher output, smaller size, and lower cost can be realized.

  Next, a description will be given of places where fixing such as welding of the stator core and rotor core of the motor of the present invention is allowed. FIG. 49 shows the stator core and rotor core shown in FIG. 48, and is assumed to be fixed by laser welding or the like. Since the welded portion is also electrically connected, an eddy current flows in the electric circuit portion of the core and iron loss occurs, so the place where welding is allowed is limited. The outer peripheral side of the rotor R1, for example, 47B, 47G, 47C, 47H, 47D, 47J, 47E, 47K, and 47F, can be welded because no magnetic flux is generated in all the electric circuits formed therein. For the rotor R2, even if welding is performed at the same point on the circumference and an integral multiple of 360 ° in electrical angle, the number of magnetic flux linkages linked to all the electric circuits to be formed is zero, which is induced. The voltage is zero and there is no harmful effect. For example, 477, 478, 479.

  The stators S1 and S2 require attention because the cores of both stators are connected. First, it is necessary to select whether to weld the stator S1 side or the S2 side. When welding the S1 side, it is possible to weld the same point on the circumference, which is an integral multiple of 360 ° in electrical angle. For example, 474, 475, 476. In these places, since the number of magnetic flux linkages linked to all the electric circuits to be formed is zero, the induced voltage of each electric circuit is zero, which is not harmful. In particular, in the stator configuration in which the stators S1 and S2 are combined as shown in FIGS. 48 and 49, the slots 474, 475, and 476 on the outer peripheral side of the stator are suitable for laser welding because of the function of the stator. Yes.

Example 12
Next, a four-phase motor will be described with reference to FIG. FIG. 50 shows a two-pole motor in which the three-phase motor shown in FIG. Reference numeral 480 denotes a rotor, and 489, 48A, 48B, 48C, 48D, 48E, 48F, and 48G are stator salient teeth. A phase winding is wound from positive A481 to negative A phase 485, B phase winding is wound from positive B482 to negative B phase 486, C phase winding is negative from positive C483 to negative Winding around the C phase 487, the D phase winding is wound from the positive D484 to the negative D phase 488. The width Hm of the rotor salient pole can be increased, and smoother operation is possible. However, there are disadvantages in that the structure is somewhat complicated and the number of transistors in the inverter is increased. Further, the present invention can be similarly applied to the motor shown in FIG.

The present invention can also be applied to a five-phase motor shown in FIG. 490 is a rotor, 49B, 49C, 49D, 49E, 49F, 49G, 49H, 49J, 49K, and 49L are stator salient teeth. A-phase winding is wound from positive A491 to negative A-phase 496, B-phase winding is wound from positive B493 to negative B-phase 498, and C-phase winding is negative from positive C495 to negative Winding to C phase 49A, D phase winding from positive D497 to negative D phase 492, and E phase winding from positive E499 to negative E phase 494. The width Hm of the rotor salient pole can be increased, and smoother operation is possible. However, there are disadvantages in that the structure is somewhat complicated and the number of transistors in the inverter is increased. Further, the present invention can be similarly applied to the motor shown in FIG. Furthermore, a multiphase motor can be configured. In particular, in a 5-phase motor, a normal motor mechanism is dominated by components of the third harmonic and the seventh harmonic, and has an effect of canceling out these harmonic components, which is preferable in terms of vibration and noise. It is.
Further, in the motor utilizing the reluctance torque of the present invention shown in FIG. 2, FIG. 7, etc., vibration noise is a big problem. For the above reason, the number of pole pairs is an integer multiple of 5 or more prime numbers such as 5 or 7. The motor having the configuration has an effect of canceling out a relatively low-order vibration, and is preferable as a motor configuration.

(Example 13)
Next, a method of reducing unnecessary leakage magnetic flux by arranging a plate of copper, aluminum or the like or an annular conductor in the vicinity of the rotor salient pole will be described with reference to FIG. Reference numerals 501 and 502 denote rotor salient poles, and reference numerals 503 and 504 denote spaces between the stator and the rotor. The motor torque is effectively generated by the magnetic flux generated in the vicinity of the air gap between the stator and the rotor. However, when a large current is applied to the motor, the torque in the spaces 503 and 504 is not so effective. Incorrect magnetic flux is also generated. This ineffective magnetic flux causes magnetic saturation of the rotor salient poles, resulting in a problem that the maximum torque of the motor is reduced. As a countermeasure, by arranging a plate such as copper, aluminum or the like as shown in 505, 506, 507, 508 or an annular conductor, a current is induced in the conductor due to the time change of the ineffective magnetic flux, A magnetomotive force is generated to prevent the change of the magnetic flux, and the increase of the ineffective magnetic flux can be reduced. Further, the present invention can be similarly applied to the motor shown in FIG.

  In addition, the conductors 505, 506, 507, and 508 are actually used with the motor of the present invention having multiple poles as shown in FIGS. 2 and 7, so that the rotor recesses should be manufactured by die casting such as aluminum. And a multi-pole motor with good productivity can be realized. This is an effective method for increasing the number of motors and increasing the torque. The circulating current between the conductors between the recesses reduces the magnetic flux effective for torque generation and increases the Joule loss of the conductors. It is necessary to prevent the circulating current from flowing between them.

Next, a method for reducing magnetic flux that is not effective for torque generation using a permanent magnet will be described with reference to FIG. In this case, since the N and S magnetic pole directions of the permanent magnet are fixed, the magnetic pole direction of the rotor is also fixed, 691 being the N pole and 692 being the S pole. Therefore, it can be driven by a three-phase AC inverter that can pass positive and negative currents, but cannot be driven by an inverter that can pass only one-way current to each winding. By disposing the permanent magnets 693, 694, 695, and 696, the ineffective magnetic flux from the circumferential side surfaces of the rotor salient poles 691 and 692 can be reduced. By arranging the same permanent magnet on the side surface in the rotor axial direction, the ineffective magnetic flux can be reduced.
Further, by adding soft magnetic bodies 697 and 698, a magnetic flux in the direction opposite to the magnetic flux of the rotor as indicated by an arrow can be generated in the rotor, and the effect of reducing the magnetic saturation of the rotor can be obtained. Is possible. Further, the present invention can be similarly applied to the motor shown in FIG.

  Next, the number of motor pole pairs that can reduce the vibration and noise of the motor of the present invention will be described. A motor using so-called reluctance torque, such as the present invention, tends to increase the magnetic flux density at the corners of each salient pole, particularly when generating a large torque, and generates a large attractive force in the radial and circumferential directions. As a result, there is a problem that vibration and noise are likely to increase compared to a surface magnet synchronous motor as shown in FIG. As a countermeasure against this problem, by making the number of pole pairs of the motor a large prime number such as 5, 7, 11, etc., the effect of canceling the harmonic component of the force generated by each pole pair over the entire circumference of the motor As a result, low vibration and low noise can be achieved. The same effect can be obtained even when the number of pole pairs of the motor is twice the value of a large prime number, 10, 14, 22 or the like. The same is true for the tripled value. The specific motor shape is such a shape that the cross-sectional shape of the 4-pole motor shown in FIG. 7 is transformed into a 5-pole pair.

(Example 14)
Next, a motor capable of simplifying the configuration of the inverter will be described with reference to FIGS. 54 and 55. FIG. In the motor of FIG. 54, the winding of the motor of FIG. 6 is changed to a double winding called so-called bi-directional winding. The winding 111 in FIG. 6 is changed to windings 521 and 522 in FIG. 54, and the winding 114 is changed to windings 527 and 528 in FIG. The winding 521 is wound in series with the winding 527, and the winding 522 is wound in series with the winding 528. Since it is desired to make the magnetic fluxes linked to the windings 521 and 522 as common as possible, it is electromagnetically preferable to wind two electric wires in parallel. Similarly, for the other windings, the winding 113 in FIG. 6 is windings 525 and 526 in FIG. 54, the winding 116 is windings 52B and 52C, the winding 115 is windings 529 and 52A, and the winding 112 is winding. Lines 523 and 524 are replaced. The winding method of FIG. 54 can be driven by the inverter shown in FIG. 55, so that the inverter is simplified and the cost can be reduced. However, since the winding is thinned to about 1/2, the resistance value is increased, the copper loss is increased, and the motor efficiency is lowered.

  The inverter shown in FIG. 55 and each winding shown in FIG. 54 will be described. 53A is a direct current power source, which indicates a direct current power source obtained by converting a commercial alternating current power source of 50 Hz or 60 Hz into direct current, or a direct current power source such as a battery. VM indicates the positive side of the DC power supply 53A, and VL indicates the negative side. The winding 531 in FIG. 55 is a winding constituted by 521 and 527 in FIG. 54, and the winding 532 is a winding constituted by 522 and 528. At this time, the windings 531 and 532 are wound in parallel at a close distance so that most of the magnetic fluxes linked to both windings can be made to be the magnetic fluxes linked in common. Further, the windings 531 and 532 are arranged so that the polarities are reversed as indicated by the dots of the windings. When the windings 531 and 532 are spatially separated in the slot as shown in FIG. 54, the windings 522 and 528 arranged at the back of the slot are the regenerative winding 532 in FIG. It is preferable that more magnetic flux is interlinked with the winding at the back of the slot. Similarly, winding 533 is windings 525 and 52B, winding 534 is windings 526 and 52C, winding 535 is windings 529 and 523, and winding 536 is windings 52A and 524.

  The method of driving the motor shown in FIG. 54 with the inverter shown in FIG. 55 can be driven by the same method as shown in FIG. 8 when driving to the CCW. However, the method for regenerating the magnetic energy of the motor is different. Specifically, in the state of FIG. 8A, a current is supplied to the winding 531 by the transistor 537, and at the same time, a current is supplied to the winding 535 by the transistor 539. In the state of FIG. 8B, the transistor 539 is turned off and the magnetic energy of the magnetic flux linked to the winding 535 is consumed by regenerating to the power source 53A using the winding 536 and the diode 53D. . The windings 535 and 536 have a large mutual inductance, and most of the magnetic fluxes are linked together, so that energy can be supplied to each other. Before and after this time, a current is passed through the winding 533 by the transistor 538 to generate a magnetic flux indicated by an arrow in FIG. 8B to generate a CCW torque.

In the state of FIG. 8C, the transistor 537 is turned off, and the magnetic energy of the magnetic flux linked to the winding 531 is consumed by regenerating to the power supply 53A using the winding 532 and the diode 53B. . Before and after this time, a current is passed through the winding 535 by the transistor 539, and a magnetic flux indicated by an arrow in FIG. 8C is generated to generate a CCW torque.
When rotating to the state of FIG. 8D, the rotational position is a rotational position advanced by 60 ° in electrical angle from the state of FIG. 8A toward the CCW direction, and the subsequent rotation in the CCW direction is It will be driven by a similar algorithm.
The inverter shown in FIG. 55 is a simple inverter composed of three transistors and three diodes, and can be reduced in cost. At the same time, the forward voltage drop of the current and the voltage drop at the diode during regeneration are about ½ compared to the ordinary three-phase AC inverter as shown in FIG. 69, which is efficient and generates little heat. In this respect, the inverter can be downsized.

  FIG. 56 shows an example of reducing the instantaneous overvoltage applied to the transistor of FIG. When the current flows through the transistor 537, the magnetic energy of the magnetic flux linked to the winding 531 is regenerated to the power source 53A by the winding 532 and the diode 53B in a simple principle. The leakage magnetic flux component of the winding 531 cannot be regenerated because it is a magnetic flux component that is not linked to the winding 532. These leakage magnetic flux components have a problem in that when the transistor 537 is turned off, an instantaneous excessive voltage is generated in the winding 531 and the transistor 537 is damaged. The diodes 541, 542, and 543 in FIG. 56 can rectify and collect excessive voltages generated in the windings 531, 533, and 535, and can absorb them to the resistor 546, the capacitor 544, and the Zener diode 545 which are overvoltage absorption circuits. Since the total amount of magnetic energy such as the leakage magnetic flux component is small, it can be solved by a relatively small overvoltage absorption circuit. The overvoltage absorption circuit can be variously modified.

  Next, the case where the motor shown in FIGS. 1 and 6 is a small capacity motor of about several W will be described with reference to FIG. Reference numerals 551, 552, and 553 denote three-phase windings, 554, 555, and 556 denote current-carrying transistors, 557, 558, and 559 denote diodes, and 55A, 55B, and 55C denote resistances that absorb the magnetic energy of the motor. When the output capacity of the motor is small as described above, the circuit can be simplified by absorbing the magnetic energy of the motor with a simple circuit. Here, the resistors 55A, 55B, and 55C can be combined with a capacitor, a Zener diode, and the like.

  Next, in the case where the motor shown in FIGS. 1 and 6 is a motor having a medium capacity or more, an example of the configuration of the inverter is shown in FIG. 58 and will be described. Reference numerals 561, 562, and 563 denote three-phase windings, 564, 565, and 566 denote transistors that conduct current, and 567, 568, and 569 denote diodes. The regenerative current collected by the diodes 567, 568, and 569 is charged to the capacitor 56C, converted into a voltage by the transistor 56A, the choke coil Ldcc, and the diode 56B, which are DC-DC converters, and charged to the DC power supply 53A. This DC-DC converter has a common configuration that is often used. The electric charge charged in the capacitor 56C is supplied to the choke coil Ldcc by the transistor 56A, and the transistor 56A is turned off in this state. At this time, the magnetic energy accumulated in the choke coil Ldcc can be charged to the DC power supply 53A in the form of a current Irc using the diode 56B. In this way, the kinetic energy of the motor and the magnetic energy of the motor are regenerated and efficiently regenerated to the DC power source 53A. Note that the regenerative voltage VH in FIG. 58 can be set according to the required regenerative characteristics. Further, the DC-DC converter can be modified.

  In an electric vehicle, a hybrid vehicle, etc., two or more motors are often used for driving a vehicle. Usually, the fuel consumption in the city driving mode, that is, the driving efficiency is a problem. Although it depends on the type of automobile, it is often less than or equal to ½ of the maximum motor torque in the urban driving mode. Accordingly, a motor-side power generation capacity during regeneration, that is, a regeneration capacity, that is 1/2 or less of the maximum output capacity during rapid acceleration is sufficient. When an automobile must decelerate rapidly, it can be considered that it is sufficient to use a mechanical brake function together from the viewpoint of safety. From such a viewpoint, the DC-DC converter as shown in FIG. 58 can be shared by a plurality of motors, and a driving circuit for driving one motor has three transistors 564, 565, 566. And three diodes 567, 568 and 569. Therefore, the inverter shown by the broken line in FIG. 58 is a simple inverter that can drive one motor by three transistors and three diodes, and can reduce the cost. At the same time, the forward voltage drop of the current and the voltage drop at the diode during regeneration are about ½ compared to the ordinary three-phase AC inverter as shown in FIG. 69, which is efficient and generates little heat. In this respect, the inverter can be downsized.

Next, another circuit for driving the motor shown in FIGS. 1 and 6 will be described with reference to FIG. Compared with FIG. 58, a transistor 571 and a diode 572 are added, and the negative connection point of the capacitor 573 is a common line VL. By adding the transistor 571, the regenerative voltage of the windings 561, 562, and 563 can be set to a large value of about (VH−VL).
Next, a method of driving the motor shown in FIGS. 1 and 6 with the inverter shown in FIG. 72 described above will be described. The three-phase winding A winding, B winding, and C winding in FIG. 6 correspond to the windings 87D, 87E, and 87F in FIG. For example, when a current is passed through the winding 87D, the transistors 871 and 872 can be controlled to apply a DC voltage, regenerate, and control the flywheel. However, only one-way current can be passed through the winding 87D. The same applies to the winding currents of other phases. The relationship between the rotational position θr of the motor and each phase current may be energized according to the relationships of FIG. 3, FIG. 4, FIG. 8, and FIG.

Next, a method of driving the motor shown in FIGS. 1 and 6 with the inverter shown in FIG. 69 will be described. The inverter in FIG. 69 is a commonly used three-phase AC inverter. Although the balanced three-phase current and voltage can be output, there is a restriction of the following equation that the sum of the three-phase currents is zero.
Iu + Iv + Iw = 0 ……………………………………………………………… (17)
The three-phase winding A winding, B winding, and C winding in FIG. 6 correspond to the windings 834, 835, and 836 in FIG. Depending on the winding arrangement, in the case of FIG. 8A, if the positive A winding is 111, the positive B winding is 113, and the positive C winding is 115, the rotational position θr of the rotor 8 and FIG. 9 cannot be energized.

  For example, the windings 111 and 112 in the adjacent slots cannot be set to the maximum value of the positive current and the maximum value of the negative current because of the relationship between the equation (17) and the winding arrangement. Further, when a positive current flows through the windings 111, 116, and 115, it can be controlled so that the current of the winding 116 is the maximum and the winding 111 or the winding 115 is the maximum positive current. Absent. However, in the state of FIG. 8A, a positive current can be passed through the windings 111, 116, and 115, and a negative current can be passed through the windings 112, 113, and 114. It is possible to generate torque in the CCW direction. However, at this time, a magnetic flux that passes from the tooth 11B to the tooth 118 through the auxiliary salient pole magnetic pole 162 is also generated, and the magnetic flux is a useless magnetic flux that does not generate torque at the rotational position θr. As described above, the relationship between the rotational position θr and each phase current is different from the relationship shown in FIGS. 3, 4, 8, and 9, but the connection between the inverter and each winding shown in FIG. The motor shown in FIG. 6 can be driven.

  Next, another circuit for driving the motor shown in FIGS. 1 and 6 will be described with reference to FIG. In the description of the motor driving method in FIGS. 3 and 8, a method has been shown in which a current in one direction is supplied to each winding and current is supplied only to the two-phase windings of the three-phase windings. This is to make the inverter simple and inexpensive. However, if optimum values for the direction and magnitude of the current can be selected for the three windings 561, 562, and 563, there are some points that can improve the driving characteristics of the motor. For example, if the direction of the magnetic flux passing through the rotor is constant, a configuration in which a permanent magnet is added to the rotor becomes easy. In addition, there is a direction and magnitude of current that can provide a large torque when the current for exciting the rotor is not supplied to the two windings but is supplied to the three windings. The inverter shown in FIG. 60 has a configuration in which the direction and magnitude of the current can be freely selected for the three-phase winding. For example, the transistors 581 and 58G can be controlled to pass a current in one direction to the winding 561, and the transistors 583 and 582 are controlled to pass a current in the opposite direction.

  Similarly, the current to winding 562 may control transistors 585 and 588 or transistors 587 and 586. Similarly, the current to winding 563 may control transistors 589 and 58C or transistors 58B and 58A. Also, the voltage between the winding terminals can be set to approximately zero volts, and so-called flywheel current control can be performed by controlling the corresponding transistor. For example, if the transistor 581 is turned on and the transistor 584 is turned off while the transistors 581 and 584 are controlled to flow through the winding 561, the current in the winding 561 passes through the diode 58F. A circulating current that returns to the winding 561 through the transistor 581 can also flow.

  Next, a method of driving a motor as shown in FIGS. 6 and 7 from low speed rotation to high speed rotation will be described with reference to FIGS. 61 and 63. FIG. The current energization method and torque generation method at low speed rotation are shown in FIGS. In FIG. 9, the time width of each current can be changed by reducing torque ripple, motor vibration noise, and energization restrictions of the inverter. The amplitude of each current can also be varied according to the rotational position θr of the rotor, and each current can be supplied to the three phases. In particular, regarding the vibration noise of the motor, the radial suction force between the stator and the rotor is very large, and the rotation angle change rate of the radial suction force is often greatly affected. Of course, torque ripple is often a problem in terms of torque fluctuation applied to the load.

When the motor shown in FIGS. 6 and 7 is driven, the rotational direction position where the magnetic flux is generated moves about 90 ° in electrical angle, and the radial attractive force between the stator and the rotor changes greatly, causing vibration. Cause. At low speed rotation, there is little problem of noise due to the relationship between the natural frequency of the motor and human hearing. However, as the motor speed increases, radial suction force vibration gradually becomes a problem. Further, from the viewpoint of inverter frequency restriction and motor efficiency, there is a problem that energizing the main salient pole magnetic pole 161 and the auxiliary salient pole magnetic pole 162 alternately is a burden.
As a countermeasure for this, at low speed rotation, as shown in the time chart of FIG. A driving method that reduces the driving at the auxiliary salient pole magnetic pole 162 and increases the ratio of driving at the main salient pole magnetic pole 161 is effective in terms of vibration noise and motor efficiency.

  Further, in the high speed rotation, the state shown in FIG. 8A is changed to the state shown in FIG. 8C without passing through the state shown in FIG. 8B, and the magnetic flux indicated by the arrow is obtained. The force also changes and is likely to cause vibration noise. Here, when the state shown in FIG. 8A is changed to the state shown in FIG. 8C, the currents of the windings 112 and 115 are not changed, and the currents of the windings 111 and 114 are gradually decreased to obtain the winding 113. 8 and 116, the magnetic flux indicated by arrows (a) and (c) in FIG. 8 can be gradually changed, and the change in the attractive force in the radial direction can be smoothed, and the vibration noise can be reduced. It is possible to reduce. For example, in the transient state changing from (a) to (c) in FIG. 8, the magnetic flux passes from the tooth 118 through the main salient pole magnetic pole 161 to the tooth 117, and at the same time, the magnetic flux passes from the tooth 11A to the main salient pole magnetic pole 161. There is a timing to pass to the passing teeth 11B. That is, the radial suction force between the stator and the rotor smoothly moves in the rotational direction, and the radial suction force can be driven without abruptly changing. In this way, operation with low vibration noise can be performed.

  FIG. 61 shows a motor control device. 591 is a motor, 592, 593 and 594 are windings, 595 is an encoder for detecting the rotational position, and 596 is an interface of the encoder 595. The rotational position 598 and the motor speed 597 are detected and output. 59A is an adder / subtractor that receives a speed command 599 and a motor speed 597 and detects a speed error. 59C is a compensator that receives a speed error signal 59B and performs PID compensation such as proportionality, integration, and differentiation, and outputs a torque command 59D. . 59G is a current command value creating means, which receives the torque command value 59D, the rotational position 598, and the motor speed 597, and reads information on the motor current and voltage from the motor drive information at that time from the motor information database 59P. The current command information 59H is output. The current command information 59H can include information such as a three-phase current command value and a motor voltage command value, which will be described later with reference to FIGS.

  Reference numeral 59J denotes current control means, which receives current command information 59H, detected current information 59N, rotational position 598 and motor speed 597, recognizes the operating state of the motor, and outputs motor voltage commands Vac, Vbc and Vcc. For example, as shown in FIG. 62, the specific calculation of the current control unit 59J is performed by subtracting the A-phase current detection value Ias from the A-phase current command value Iac by the adder 742 and adding the difference value Iae to the compensator 741. Compensation calculation is performed to obtain a voltage command value Vai, a feedforward voltage Vaf is calculated, a sum Vac of the voltage command value Vai and the feedforward voltage Vaf is obtained by an adder 743, and is output to the power converter 59L. The same applies to the B-phase and C-phase voltage command values Vbc and Vcc. 59L is a PWM modulation converter and inverter, and supplies the A-phase current Ia, the B-phase current Ib, and the C-phase current Ic to the motor 591.

  FIG. 63 is a diagram showing the drive ratio of the rotor salient poles. At low speed rotation N1, the torque value of the main salient pole magnetic pole 161 and the torque value of the auxiliary salient pole magnetic pole 162 are approximately the same, and the rotational speed is increased. N2 increases the torque value of the main salient pole magnetic pole 161, decreases the torque value of the auxiliary salient pole magnetic pole 162, and further increases the torque value of the main salient pole magnetic pole 161 when the rotation speed reaches N3. This is a characteristic in which the torque value at the magnetic pole 162 becomes zero. In the high-speed rotation region where the rotation speed is N3 or more, driving is performed only with the torque of the main salient pole magnetic pole 161. However, when the rotation frequency increases, the influence of torque pulsation at that frequency on the rotation fluctuation decreases. There are many applications that can be used without problems in practice. 63, the operating points R1, R2, R3, R4, and R5 are stored as data, and the interval between the operating points can be obtained by interpolation, so this operation can be performed with a small amount of data. The characteristics can be stored and realized.

  Next, a method for obtaining the current command values Iac, Ibc, and Icc for each phase from the torque command 59D and the rotational position 598 will be described. FIG. 64 is a data table in which the horizontal column is the rotor rotational position θr and the vertical column is the torque Tc. In one column Pmn of the table, the values of the three-phase current values Iac, Ibc and Icc corresponding to the rotor rotational position θr and the torque Tc are obtained in advance by magnetic field analysis using a finite element method. It is also possible to store actual measurement values in this data table. Since the data table is a current value corresponding to a finite discrete rotor rotational position θr and torque Tc, the values of the three-phase current values Iac, Ibc, and Icc corresponding to the intermediate values are interpolated. Can be obtained. The method of storing various data in a table and determining and controlling the final value by interpolation calculation is the so-called data table method or map method, and the method itself is a new method. Absent.

  In addition, information on the rotor rotational position and voltage can be included in each column of the table of FIG. By adding the voltage information, it is possible to output the feedforward voltage Vaf in addition to the current command value of each phase, and as shown in FIG. 62, more accurate current control is possible, and even higher rotation speeds can be achieved. It can be controlled with high accuracy. In particular, with regard to the feedforward voltage Vaf and the current command value Iac, by performing predictive calculation of the voltage value and current value based on the motor speed 597, the control accuracy at high speed rotation can be greatly improved.

Next, a method for obtaining the voltage of each winding will be described. For example, it is the value of the voltage Vaf of the A-phase winding shown in FIG.
The first method is a method in which voltage information under various conditions is also stored as data in the table of FIG. 64, and is obtained by performing interpolation calculation using discrete data as needed. The second method is a method having a data table of the flux linkage number Ψ as shown in FIG. FIG. 65 is a table of magnetic flux linkage number Ψ with the horizontal column representing the rotor rotational position θr and the vertical column representing the current I. The relationship between the number of flux linkages Ψ and the voltage V is expressed by the following equation.
V = dΨ / dt≈ΔΨ / Δt = (Ψ2-Ψ1) / Δt (18)
This flux linkage number Ψ is a function of the rotational position θr and also a function of current. Therefore, it is necessary to calculate the flux linkage number change ΔΨ from the change in the rotational position θr and the change in the current I during the minute time Δt, and to obtain the voltage using equation (18). Since the rotational position θr and the current I in the data table are discrete values, the value of the flux linkage number Ψ is calculated by interpolation calculation. The values Ψ1 and Ψ2 of the number of magnetic flux linkages before and after the minute time Δt are obtained.

Assuming that the current I in FIG. 65 is handled as shown in FIG. 8, each phase current can be determined relatively. However, the torque generated by the current does not have ideal torque characteristics as shown in FIG. 9, but often has characteristics distorted by nonlinearity such as leakage magnetic flux or magnetic saturation. About this distortion, what kind of correction may be performed as needed. In addition, although the vertical column in FIG. 65 is the current I, data can be configured as the torque T.
The data table creation method can be modified in various ways, and a method of using the data table of FIG. 64 and the data table of FIG. 65 in a balanced manner, or a method of including the other data in one of the data tables and so on.

Further, since the flux linkage number ψ, the number of turns Nw, the inductance L, and the current I are expressed by the following equations, the data table of FIG. Even if it is obtained as a table, the same answer can be obtained and is included in the present invention.
V = dΨ / dt = Nw × dΦ / dt = L × dI / dt (19)
When this equation is integrated over time, the following equation is obtained.
Ψ = Nw × Φ = L × I ……………………………………………………………… (20)

  When a magnetic field analysis is performed on the motor model by a finite element method or the like, the interlinkage magnetic flux φ interlinked with each winding can be easily obtained. The data table of the number of magnetic flux linkages Ψ shown in FIG. 65 can be calculated including calculation of conversion necessary for motor control by copying the data of the magnetic field analysis result to the spreadsheet software and using it. If a formula is defined in advance in the spreadsheet software, it can be used repeatedly thereafter. These calculations can be performed relatively easily using current magnetic field analysis tools and general-purpose software.

In the above, the example of the various forms regarding this invention was demonstrated. An object of the present invention is to reduce the size, increase the performance, reduce the cost, improve the quality, and the like of a motor drive system including a motor and its control device. In particular, in recent years, the prices of rare earth magnets and heavy metals have soared in relation to resource problems, and they are also motors that do not use rare earth magnets or the like, and motors that use a small amount. Hereinafter, the effects of the present invention will be described together.
One of the specific effects of the present invention is cost reduction due to not using expensive rare earth magnets or using a small amount, as shown in FIGS. The fact that it is less susceptible to price fluctuations related to global resource issues is also a major advantage.

The relax tank torque is a characteristic in which the torque is generated without depending on the direction of the magnetic flux, and this characteristic is advantageous and disadvantageous depending on the configuration of the motor. Since the control device for driving a motor that effectively utilizes the relax tank torque of the present invention has a configuration and a driving method that can be configured with a small number of transistors, the control device side can also be realized at low cost. Furthermore, when the current is applied to the winding, the number of semiconductors arranged in series with the winding can be reduced from the usual two to one, reducing the loss of the inverter, increasing the efficiency of the inverter, Miniaturization and cost reduction can also be realized.
As for the utilization method of the soft magnetic material, if the winding current is a unidirectional motor, the magnetic flux density changes dynamically using the fact that the magnetic flux direction is unidirectional at each location of the stator. A method has been proposed in which the width is doubled. Specifically, the method shown in FIGS. 18 to 22 is used to obtain a change width of magnetic flux density close to 4T with a normal silicon steel sheet, increase the maximum torque of the motor, and consequently reduce the size and cost of the motor. Is realized.

In the motor of the present invention, the maximum torque of the motor can be increased and the motor can be reduced in size by simultaneously realizing the multipolarization of the motor and the leakage flux reduction measure. This is based on the premise that the torque of the motor can be increased in proportion to the number of poles, while the leakage magnetic flux that is an adverse effect of multipolarization is reduced by the leakage magnetic flux reduction measure. Here, the current density of the winding is ignored because it discusses the maximum motor torque in a short time.
It should be noted that the stator magnetic flux can be unidirectional at each location not in the vicinity of the maximum torque of the motor but rather in a light load operation state, so that the change width of the magnetic flux density is a normal permanent magnet type AC motor. It is possible to operate with a stator iron loss reduced to about 1/4.
Further, since the motor rotor shown in FIGS. 1 and 6 can be manufactured very firmly, it can be used at high speed. A high output can be realized, and the output per volume, that is, the output density can be improved.

In addition, the noise of the motor using the relax tank torque may be a big problem. In the motor of the present invention, the direction of the radial suction force between the stator and the rotor can be smoothly moved in the rotation direction along with the rotation of the rotor, and the variation in the magnitude of the radial suction force can be reduced. Therefore, quiet motor operation is possible.
With respect to the winding factor of the stator, the length of the coil end of the winding, and the productivity of the winding, the motors shown in FIGS. 66 and 67 have left room for improvement. In the motor according to the present invention, the whole stator winding, concentrated winding, and the structure in which the windings of each phase are not crossed, the stator is simplified, and the slot shape, winding shape and insulating paper shape are further devised. High space and shortening of coil ends are possible. Another characteristic is that the winding resistance is smaller than that of a conventional switched reluctance motor.

As for the method of constant output control of the motor of the present invention, in addition to the conventional method of reducing the magnitude of magnetic flux at high speed rotation, a method of effectively performing constant output control by utilizing the magnetic saturation of the soft magnetic material is proposed. did.
It was also shown that the motor characteristics can be further improved by adding a slit to the rotor and adding a magnet. Then, two motors of the present invention are arranged on the outer diameter side and the inner diameter side to simplify the windings, and further, windings of different configurations are added to the slots of the motors on the outer diameter side so that each motor characteristic Showed how to optimize.
Moreover, the concrete control method of the motor of the present invention was shown, and it was shown that the motor and the control device can be realized with a simple configuration.

As mentioned above, although the example of the various form regarding this invention was demonstrated, this invention can be variously deformed and is included in this invention. For example, there is no restriction on the number of poles of the motor, and the type of rotor has been described as a surface magnet type rotor, but it can be applied to rotors of various structures. Various torque ripple reduction techniques can also be applied to the motor of the present invention. For example, a method of smoothing the shape of the stator magnetic pole and rotor magnetic pole in the circumferential direction, a method of smoothing in the radial direction, a method of moving some rotor magnetic poles in the circumferential direction and canceling torque ripple components, etc. There is.
Various forms of motors are possible. Expressed by the shape of the air gap between the stator and rotor, the inner rotor type motor, the outer rotor type motor, and the air gap shape are discoid. It can be transformed into an axial gap type motor or the like. The linear motor is also a modification.

In addition, a motor shape in which the air gap shape is a slightly tapered shape from the cylindrical shape is also possible. In addition, a plurality of motors including the motor of the present invention can be combined and manufactured. Moreover, the structure which abbreviate | omitted and abbreviate | omitted a part of motor of this invention is also possible. As the soft magnetic material, in addition to using a normal silicon steel plate, it is possible to use an amorphous magnetic steel plate, a powder magnetic core obtained by compression molding powdery soft iron, and the like. In particular, in a small motor, a three-dimensional shape part can be formed by punching, pressing and forging a magnetic steel sheet to form a part of the above-described motor of the present invention.
As for the current to be supplied to the motor, an example in which the current of each phase has a specific shape has been described, but it is also possible to control with a current of various waveforms such as control with a sine wave current. With respect to these variously modified motors, the modified technology intended to be the subject of the motor of the present invention is included in the present invention.

It is a figure which shows the cross section of the three-phase two-pole motor which is embodiment of this invention, and the winding arrangement | positioning of each phase. It is a figure which shows the cross section of 3 phase 8 pole motor which is embodiment of this invention, and arrangement | positioning of each slot. It is a figure which shows the electric current conduction state, magnetic flux, and torque generation of the motor of FIG. It is a figure which shows the relationship between the rotor rotational position (theta) r of the motor of FIG. 1, and the electric current and torque of each winding. It is a cross-sectional view of the motor which improved the rotor shape of the motor of FIG. It is an embodiment of the present invention, and is a view showing a cross section of a motor in which a main salient pole magnetic pole and an auxiliary salient pole magnetic pole are arranged on a rotor, and an arrangement of each slot. It is a figure which shows the cross section of the motor which deform | transformed the motor of FIG. 6 into 3 phase 8 pole, and arrangement | positioning of each slot. It is a figure which shows the relationship between the rotor rotational position (theta) r of the motor of FIG. 6, and the electric current and torque of each coil | winding. It is a figure which shows the relationship between the rotor rotational position (theta) r of the motor of FIG. 6, and the electric current and torque of each coil | winding. It is a figure which shows the relationship of the shape of each part of the motor of FIG. 6, an electric current, and magnetic flux. It is a figure which shows the relationship of the shape of each part of the motor of FIG. 6, an electric current, and magnetic flux. It is a table | surface which shows the shape of each part of the motor of FIG. 6, and the relationship between an electric current and magnetic flux. It is a figure which shows the example of a deformation | transformation of each part shape of a stator and a rotor. It is a figure which shows the cross section of a motor with a rotor shape asymmetrical, and a characteristic which differs in CCW and CW, and arrangement | positioning of each slot. It is a figure which shows the relationship between an electric current and a torque, especially the maximum torque of a motor. It is a figure which shows the magnetic saturation of a tooth | gear when a motor generate | occur | produces a big torque. It is a figure which shows the magnetic saturation of the auxiliary salient pole when a motor generates a big torque. (A) is a longitudinal cross-sectional view of a conventional motor, and (b) is a longitudinal cross-sectional view showing a method for reducing magnetic saturation of teeth. (A) is a figure which shows the method of reducing the magnetic saturation of a tooth | gear, (b) is the elements on larger scale. It is a figure which shows operation | movement of the magnetic flux of a tooth | gear. (A) is a figure which shows the method of reducing the magnetic saturation of a tooth | gear, (b) is the elements on larger scale. It is a figure which shows the method of reducing the magnetic saturation of a tooth | gear. (A) is a longitudinal cross-sectional view which shows the magnetic saturation of a rotor, (b) is the side view. It is a figure which shows the method of utilizing the magnetic saturation of a soft magnetic body, when driving | operating a motor by high speed rotation. It is a figure which shows the method of utilizing the magnetic saturation of a soft magnetic body, when driving | operating a motor by high speed rotation. It is a figure which shows the constant output characteristic of the high-speed rotation area | region of a motor. It is a figure which shows the magnetic characteristic in the case of implement | achieving the constant output characteristic of the high-speed rotation area | region of a motor using the magnetic saturation of a soft magnetic body. It is a figure which shows the cross section of a 4 pole motor by the structure which improves anisotropy, without reducing the magnetic flux density of a rotor. It is a figure which shows the cross section of a 4-pole motor by the structure which improves anisotropy, without reducing the magnetic flux density of a rotor. It is a figure which shows the cross section of the motor of 4 poles which is the structure which improves anisotropy without reducing the magnetic flux density of a rotor, and is equipped with an auxiliary salient pole. It is a figure which shows the cross section of the motor which added the permanent magnet to the rotor. It is a figure which shows the cross section of the motor which gave concentrated winding for every tooth. It is a figure which shows the relationship between the winding current in the state of a certain specific operation | movement of a motor, magnetic flux, and magnetic saturation. It is a figure which shows the relationship of the magnetic flux and magnetic saturation when the shape of a slot and the density of an electric current are improved. It is a figure which shows the utilization example of a material with a high shape of a slot, a current density, and a saturation magnetic flux density. It is a figure which shows the state of the winding of a stator of 3 phases, 4 poles, and 12 slots. It is a figure which shows the shape of the inside of a slot, the shape of a coil end, and the rotor axial direction edge part vicinity of a stator core. It is a figure which shows the winding arrangement | positioning structure with little mutual crossing and interference of each winding | winding by the stator of 3 phases, 4 poles, and 12 slots. It is a figure which shows the winding arrangement | positioning structure with little mutual crossing and interference of each winding | winding by a stator of 3 phases, 6 poles and 18 slots. It is a figure which shows the stator core of a coil end part vicinity, slot shape, winding shape, and insulating paper shape. It is a figure which shows the stator core of a coil end part vicinity, slot shape, winding shape, and insulating paper shape. It is a figure which expands and shows a stator core shape, slot shape, and coil | winding shape linearly. It is a figure which shows the shape of the electromagnetic steel plate of the rotor axial direction end of a stator core. It is a figure which shows the cross section of the motor by which the stator core was divided | segmented every 360 degrees by the electrical angle. It is a figure which shows the structure of the junction part of the divided | segmented stator core. It is a figure which shows the clearance structure of the junction part of the divided | segmented stator core, and distribution of magnetic flux. It is a figure which shows the structure of the junction part of the divided | segmented stator core, and the coupling | bonding by welding. FIG. 5 is a diagram showing a winding relationship of a winding in a composite structure motor in which two motors are arranged on an outer diameter side and an inner diameter side. It is a figure which shows the example of the site | part which can be welded to a rotor axial direction by the stator core and rotor core of a motor of the composite structure which arrange | positions two motors to an outer diameter side and an inner diameter side. It is a figure which shows the cross section of the motor of 4 phases, 2 poles, and 8 slots. It is a figure which shows the cross section of a motor of 5 phases, 2 poles, and 10 slots. It is a figure which shows the cross section of the motor which arrange | positions the conductor plate or the conductor which formed the closed circuit in the circumferential direction end of a rotor salient pole, and reduces a leakage flux. It is a figure which shows the cross section of the motor which has arrange | positioned the permanent magnet and the magnetic body to the circumferential direction end of the rotor salient pole. It is a figure which shows the cross section of the motor which arrange | positions two windings of the same phase wound in parallel in each slot. It is a figure which shows the inverter which drives the motor of FIG. It is a figure which shows the inverter which drives the motor of FIG. It is a figure which shows the inverter which drives the motor of FIG. It is an inverter that drives a three-phase motor controlled by a one-way current. It is an inverter that drives a three-phase motor controlled by a one-way current. This is a three-phase inverter capable of current control with currents in both directions and without relative restrictions on each current. It is an example of the control apparatus which controls this invention motor. FIG. 62 is a detailed view of a current control portion of the control device of FIG. 61. It is a figure showing the drive ratio of the main salient pole magnetic pole and auxiliary salient pole magnetic pole in the case of driving the motor of the present invention shown in FIGS. 6 and 7 from low speed to high speed rotation. FIG. 62 is a diagram showing a format having data such as rotor rotational position θr, torque T, current I, voltage V and the like in the control device of FIG. 61. FIG. 62 is a diagram illustrating a format having data such as a rotor rotational position θr, a current I, and a flux linkage number Ψ in the control device of FIG. 61. It is a figure which shows the longitudinal cross-section of the conventional motor. It is a figure which shows the cross section of the conventional motor. FIG. 68 is a winding diagram of FIGS. 66 and 67; It is a figure which shows the connection of a structure of a three-phase alternating current inverter, and a three-phase alternating current motor. It is a figure which shows the cross section of a 3-phase alternating current, 2 pole, 12 slot synchronous reluctance motor provided with a multi-flux barrier type | mold rotor. It is a figure which shows the cross section of the switched reluctance motor which has a stator with six teeth symmetrically and four salient poles. FIG. 72 is an inverter that drives the switched reluctance motor of FIG. 71. FIG.

Explanation of symbols

110 Stator 117, 118, 119, 11A, 11B, 11C Teeth 111, 114 A phase winding 113, 116 B phase winding 115, 112 C phase winding 161 Main salient pole magnetic pole 162 Auxiliary salient pole magnetic pole 53A DC power supply 564 A Phase drive transistor 565 B phase drive transistor 566 C phase drive transistor

Claims (39)

A stator core having a slot for winding a three-phase full-pitch winding and having six teeth between electrical angles of 360 °;
Full-pitch winding, concentrated winding three-phase winding arranged in each slot of the stator,
A motor comprising a rotor having a main salient pole magnetic pole made of a salient pole-shaped soft magnetic material.   2. The rotor according to claim 1, wherein the rotor has two main salient poles with an electrical angle of 360 °, and includes a protrusion of a soft magnetic material in a circumferential direction in the vicinity of the tip of the rotor. motor.   The motor according to claim 1, wherein the rotor includes two main salient poles and two auxiliary salient poles at an electrical angle of 360 °. The main salient pole magnetic pole of the rotor has a width Hm in a circumferential direction of a portion facing the stator of an electrical angle of 35 ° to 60 °,
4. The motor according to claim 3, wherein the auxiliary salient pole magnetic pole of the rotor has a circumferential width Hh of an electrical angle smaller than the circumferential width Hm at a portion facing the stator.   The shape of the stator teeth, the main salient pole magnetic pole of the rotor, or the auxiliary salient pole magnetic pole of the rotor and the relative circumferential positional relationship are asymmetric in the circumferential direction. The motor described in.   The motor according to any one of claims 1 to 5, wherein a soft magnetic material is added to a rotor axial end of the stator teeth.   6. The motor according to claim 1, wherein a soft magnetic material and a permanent magnet are added to the stator tooth or the rotor axial end of the rotor magnetic pole.   The motor according to any one of claims 1 to 5, wherein an excitation winding for exciting a soft magnetic material and a magnetic flux is added to a tooth of the stator or a rotor axial end of the rotor magnetic pole.   The motor according to any one of claims 1 to 5, wherein a permanent magnet is arranged in the slot of the stator so that the magnetic flux passes between adjacent teeth.   The shape of the stator teeth, the main salient pole of the rotor, or the auxiliary salient pole of the rotor is such that the circumferential width on the back yoke side is larger than the circumferential width of the tip. The motor according to any one of claims 3 to 5, wherein the motor is characterized.   6. The motor according to claim 3, wherein a soft magnetic material is added to a rotor axial end of the auxiliary salient pole magnetic pole of the rotor. Add an excitation winding or permanent magnet to excite a soft magnetic material and magnetic flux at the rotor axial end of the stator tooth or the rotor magnetic pole,
6. The motor and control device according to claim 1, wherein when the motor is operated at high speed, the motor current of each slot is controlled in a direction in which the magnetic flux of the teeth is strengthened. A permanent magnet is disposed in the slot of the stator so that the magnetic flux passes between adjacent teeth,
The motor is controlled so that the direction of the magnetic flux excited by the permanent magnet and the direction of the magnetic flux excited by the motor current are the same when performing motor operation at high speed rotation. The motor and the control device according to claim 5.   The motor according to any one of claims 1 to 5, wherein a slit is provided in a direction from one magnetic pole to another magnetic pole in the main salient pole of the rotor.   The motor according to claim 14, wherein the main salient pole of the rotor includes a slit in a direction from one magnetic pole to another magnetic pole, and a permanent magnet in the slit.   The motor according to any one of claims 1 to 5, wherein the means according to claims 6 to 15 are combined. 6 teeth between the electrical angle of the stator 360 °,
Concentrated winding with short-pitch winding wound around each tooth,
Two main salient poles of the rotor which are arranged between the electrical angle of the rotor 360 ° and the circumferential width Hm is an electrical angle of 35 ° to 60 °;
A motor comprising two auxiliary salient poles that are respectively disposed between the two main salient poles and have a circumferential width Hh smaller than the circumferential width Hm.   18. The motor according to claim 1, wherein a cross-sectional area SBT of a winding disposed on the opening side of the slot is smaller than a cross-sectional area SBB of a winding disposed on the back yoke side.   18. The soft magnetic material SMA on the tooth slot opening side has a saturation magnetic flux density relatively higher than the saturation magnetic flux density of the soft magnetic material SMB on the back side of the tooth slot. The motor described in.   The motor according to any one of claims 1 to 17, wherein an amorphous metal is used for the soft magnetic material of the rotor. A three-phase A phase, B phase, C phase, Nn pole motor,
The number of slots is 3 × Nn,
At least 12 slots are wound around the negative A-phase slot S4, which is arranged through the outer diameter side at the rotor axial end of the slot S1 around which the A-phase winding is wound, and separated by 3;
A C-phase winding is wound from the slot S1 to the fourth slot S5, arranged at the rotor axial end through the outer diameter side, and wound around three negative C-phase slots S8,
A B-phase winding is wound from the slot S1 to the eighth slot S9, arranged at the rotor axial end through the outer diameter side, and wound around three negative B-phase slots S12,
A B-phase winding is wound from the slot S1 to the second slot S3, arranged at the rotor axial end through the side surface of the rotor axial end of the slots S4 and S5, and three negative B-phases separated from each other Is wound around the slot S6,
A phase A winding is wound from the slot S1 to the sixth slot S7, arranged at the rotor axial end through the side surface of the rotor axial end of the slots S8 and S9, and is separated by three negative A phases. The motor according to any one of claims 1 to 3, wherein the motor is wound around a slot S10.
Here, Nn is an even integer of 4 or more, and slots S1 to S12 are slots arranged in the circumferential direction in the order of the number. In the vicinity of the axial end of the stator core, the shape of the slot is widened to the back yoke side,
In the vicinity of the axial end of the stator core, the shape of the slot is bent from the axial direction to the circumferential direction,
It is characterized in that the physical interference between the windings wound in the respective slots is reduced, and a stator core shape is formed in which each winding can be wound with a gentle curve from each slot to the side surface in the rotor axial direction of the stator core. The motor according to any one of claims 1 to 3 and claim 21.   The motor according to claim 22, wherein a back yoke portion of the stator core is disposed on an outer peripheral side or an inner peripheral side of a coil end portion of the stator.   24. The motor according to claim 22, wherein the insulating material ZZ1 in the slot and the insulating material ZZ2 in the vicinity of the end in the axial direction in the slot are overlapped at a portion where the slot shape is widened. The circumferential width of the soft magnetic body of the stator is composed of a combination of divided cores divided into an integral multiple of 360 ° in electrical angle,
4. The motor according to claim 1, wherein a winding to be wound around the divided core is wound around the width of an integral multiple of 360 ° in terms of an electrical angle of the stator.   26. The composite motor according to claim 1, wherein the first stator and the rotor are disposed on the outer diameter side of the motor, and the second stator and the rotor are disposed on the inner diameter side of the motor. Motor. Place the first rotor on the outer diameter side of the motor,
Place the second rotor on the inner diameter side of the motor,
Disposing a first stator and a second stator between the first rotor and the second rotor;
A first set of windings of both motors is wound between the slot of the first stator and the slot of the second stator,
27. The motor according to claim 26, wherein a second set of windings are wound between the slots of the first stator.   The motor according to any one of claims 1 to 27, wherein the interior of the motor core is joined by welding or the like at an electrical angle of an integral multiple of 360 °.   The motor according to any one of claims 1 to 17, wherein the stator has a stator configuration having four or more phases.   The motor according to any one of claims 1 to 6, wherein a permanent magnet or a conductor is disposed around the rotor magnetic pole.   31. The motor according to claim 1, wherein the number of pole pairs of the motor is an integral multiple of a prime number of 5 or more. A plurality of windings of the same phase are wound in parallel in each slot of the three-phase stator,
The diode D1 is connected in series to the winding on one side of the parallel winding and connected to the power source, and the other two-phase windings are similarly connected to the power source by connecting diodes D2 and D3 in series.
17. The motor and control device according to claim 1, wherein the other winding is connected to transistors TR1, TR2, and TR3 that control the current of the motor. Connect one terminal of each three-phase winding of the motor to the power supply,
Each other end of the three-phase winding is connected to each driving transistor TR4, TR5, TR6,
The diodes D4, D5, and D6 are connected to connection points between the other ends of the three-phase windings and the driving transistors TR4, TR5, and TR6, respectively. Motor and control device. Connect one terminal of each three-phase winding of the motor to the emitter of transistor TR7,
17. The motor and control device according to claim 1, wherein each other end of the winding is connected to a collector of a transistor TR8, TR9, TR10. One terminal of each three-phase winding of the motor is connected to each emitter of the transistors TR11, TR12, TR13,
16. The motor and control device according to claim 1, wherein the other ends of the windings are connected to collectors of transistors TR14, TR15, TR16. One terminal of the motor winding WU is connected to the emitter of the transistor TR17 and the collector of the transistor TR18;
One terminal of the winding WV is connected to the emitter of the transistor TR19 and the collector of the transistor TR20,
One terminal of the winding WW is connected to the emitter of the transistor TR21 and the collector of the transistor TR22,
The motor and the control device according to claim 1, wherein the other ends of the windings WU, WV, and WW are connected to each other. The transistor TR23 and the transistor TR24 are connected in series between the positive voltage and the negative voltage of the power supply, one end of the winding WU is connected to the middle point, and the other end of the winding WU is connected to the positive voltage and the negative voltage of the power supply. Between the transistor TR25 and the transistor TR26 arranged in series with each other,
The transistor TR27 and the transistor TR28 are connected in series between the positive voltage and the negative voltage of the power supply, one end of the winding WV is connected to the intermediate point, and the other end of the winding WV is connected to the positive voltage and the negative voltage of the power supply. Between the transistor TR29 and the transistor TR30 arranged in series with each other,
The transistor TR31 and the transistor TR32 are connected in series between the positive voltage and the negative voltage of the power supply, one end of the winding WW is connected to the intermediate point, and the other end of the winding WW is connected to the positive voltage and the negative voltage of the power supply. 16. The motor and control device according to claim 1, wherein the motor and the control device are connected to an intermediate point between a transistor TR33 and a transistor TR34 arranged in series with each other.   In the low-speed rotation, the main salient pole magnetic pole and the auxiliary salient pole magnetic pole of the rotor are driven alternately. 5. The motor and the control device according to claim 3, further comprising motor control means for reducing a current for driving with the auxiliary salient pole.   The motor and the control device according to any one of claims 1 to 15, further comprising a data table in which current information of each winding corresponding to the torque and rotation position of the motor is tabulated.

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