JPS59110201A - Comb line type band-pass filter - Google Patents

Abstract

PURPOSE:To obtain a comb line type BPF which has superior dielectric strength characteristics and stable electric characteristics by using resonance elements which have axial length nearly a quarter a resonance frequency and specifying intervals between the resonance elements. CONSTITUTION:The resonance elements 21-2n have axial length a quarter resonance wavelength in terms of electric length and are arrayed in a line with the same polarity. Further, input/output coupled circuit elements 30-3n+1 are provided and the center interval between resonance elements is specified by equations I . In this case, Cdk.(k+1) is the center interval between the resonance elements, k=1-n, and (n) is the degree of a filter; and (d) is the diameter of the resonance elements, W is the width of a housing 1, and Lck.(k+1) is reactance loss between resonance elements. Further, Mk.(k+1) is the coefficient of the magnetic field coupling between the resonance elements, and lambdac=2W(epsilon)<1/2>, where lambdac is cutoff wavelength, lambda is the wavelength of a transmit signal, and epsilon is a dielectric constant. Consequently, the comb line type BPF which has superior dielectric strength characteristics and stable electric characteristics is obtained.

Description

Detailed Description of the Invention - / Conventionally, ink-digital BPFs or hole-coupled BPFs have been mainly used as BPFs using coaxial resonators. are arranged alternately with opposite polarity, and the outer ends of the screws for resonant frequency fine tuning alternately protrude outward from the earth wall and bottom wall of the casing, making the shape relatively complex and large.
During design and manufacturing, there is a large error between the theoretical value and the experimental value, so it is necessary to confirm each case experimentally and correct the error, which has the disadvantage of requiring a lot of time and effort.

Since the hole combination type BPF requires drilling of the connection hole, etc., it is unprofitable if the manufacturing cost of the casing increases, and it has the disadvantage that adjustment requires a lot of time and labor.

The comline type BPF has the advantages of being relatively small, inexpensive to manufacture, and relatively easy to adjust, but has the following drawbacks.

Figure 1 shows a combline type BP consisting of a conventional coaxial resonator.
A sectional view showing an example of F (BB sectional view in Fig. 2),
The figure is a sectional view taken along the line A-A in Figure 1, and Figure 3 is its equivalent circuit diagram, where 1 is an electromagnetic shielding case, 2 is a resonant element made of a bar-shaped conductor, and 3 (input/output coupling made of a bar-shaped conductor). circuit element, 4
is an input/output coaxial terminal, 5 is an electrode plate that forms a load capacitance,
In this BPF'l, the axial length of the resonant element 2 is selected as fil of the resonant wavelength, and resonance is achieved by the capacitance between the electrode plate 5 attached to the open end of each resonant element and the housing 1. In addition, the resonant elements are coupled by an interstage coupling capacitance C (2, cai'...).However, the load capacitance formed between the electrode plate 5 and the housing 1 is relatively large. As a result, each gap between the electrode plate 5 and the housing 1 becomes narrower, deteriorating the pressure resistance characteristics. The electrostatic capacitance formed between them changes significantly depending on the change in power, and it has drawbacks such as not being able to obtain stable electrical characteristics, making it extremely unsuitable for high-power transmission BPP, etc. .

Further, as described above, the resonant elements are configured to be coupled by electric field coupling. Therefore, when a dielectric resonator is formed by interposing a dielectric between the resonant element and the housing, the coupling coefficient between the legs changes depending on the dielectric constant of the dielectric, making the design difficult.

The present invention has excellent withstand voltage characteristics, has stable electrical characteristics without being affected by changes in ambient temperature, and is suitable for high power applications. It is an object of the present invention to realize a combline type BPF for very high frequency waves or microwaves that does not cause a change in the interstage coupling coefficient due to a change in the ratio and is therefore easy to design and manufacture.

FIG. 4 is a sectional view showing one embodiment of the present invention (B--B in FIG. 5).
B sectional view), FIG. 5 is a sectional view taken along line AA in FIG. Ya1
are input/output coupling circuit elements % 4a and 47L? + is an input/output coaxial terminal. Then, the width W of the housing 1 and the resonant element 2
When the diameter d of 1 to 2n is small compared to the resonant wavelength (for example, the diameter d is about hO to iO of the resonant wavelength),
The axial lengths of the resonant elements 2I to 2□ are formed to be electrical lengths that correspond to the resonant wavelength, and the width W of the housing 1 and the resonant elements 2I to 2n are
When the diameter d of the resonant element 2. is relatively large with respect to the resonant wavelength, the resonant element 2. The axial length is set to 2 to 2 electrical lengths, which is appropriately shorter than the resonance wavelength of 4, so that the mechanical free end of each resonant element is kept electrically open.

FIG. 6 is an equivalent circuit diagram of the BPF of the present invention shown in FIGS. 4 and 5, where R1 to Rn are resonator circuits consisting of resonant elements 2- to 2n and casing 1, and M+a % MJj
%...Mζ stop,) hole is magnetic field coupling coefficient, Co
, I and C7, Cess, ) are input/output coupling capacitances. In the BPF of the present invention configured as described above, when a resonant current flows through the first-stage resonant element 21, the resonant element 2. A TEM mode wave is generated between the casing 1 and the casing 1, and its magnetic field component is formed by the casing 1 between the resonant elements 2I and 22, and the cut-off wave is determined by the horizontal width W of the casing 1. The wave tube section is excited (the element 21 serves as a probe or antenna and excitation is performed), resulting in an L+ mode wave having an electric field component E shown in FIG. 7 and a magnetic field component H shown in FIG. 8. Excite two resonant elements in the stage. 1. Signal transmission is performed in the same manner.

Hl in the above cut-off waveguide section and the cutoff wavelength λ of the mode wave are expressed by the equation (1), and the reactance loss Lc between the resonant elements is calculated by the equation (2). Analysis of the results shows that the experimental formula (3) is more accurate because there is disturbance in the electromagnetic field near the resonant element.

λ, = 2'W (i...
... (1) ... (3) ε: A value determined by the material interposed between the housing 1 and the resonant element,
In the case of air, ε=1; in the case of a dielectric, it is its dielectric constant.゛λ: Extra free space length of transmission signal cd = Center distance between adjacent and opposing resonant elements Signal transmission between resonant elements Actance loss Lc From magnetic field coupling coefficient between resonant elements M+, a % M, +, j...・・・M/
7L-ri, 71 (collectively expressed as M) can be found, but since the strength of the magnetic field is proportional to the resonance current a → magnitude, the actance loss LQ between each resonant element and the magnetic field coupling coefficient M can be calculated. The following relationship holds true between them.

LQ=-2ologM・・・・−・(4) Once M=IO”・・・・・・(5) Housing 1
width W, and center distance Cd between the resonant elements.

Diameter of the resonant element d2 Given ε, which is determined by the material interposed between the resonant element and the housing, and the wavelength equation of the transmission signal, the magnetic field coupling coefficient M can be obtained from equations (3) or 5), and the magnetic field coupling coefficient is When M is given, the center distance Cd between the resonant elements can be found from equations (3) to (5).

Next, when designing a distributed constant BPF, find the geometric coefficients of the scaled low-pass filter, and based on this, the BPF
The general design method is to obtain the required transmission characteristics by determining the circuit constants of . fc
A case will be explained in which a BPF with a Chebyshev characteristic in the passband and a Wagner characteristic in the attenuation region is obtained based on the geometric coefficients of a Chebyshev-type normalized low-pass filter that shows a curve diagram of the transmission characteristic at the cutoff frequency). do.

First, if the voltage standing wave ratio (VSWR) within the passband allowed in the design of the BPF is S, then the allowable ripple Lar within the passband is expressed by the following equation.

From the above formula, 1. , ar and determine the circuit order n,
The geometric coefficient g1 is obtained from equation (7), and the geometric coefficients g2 to g, 1 are obtained from equation (8).

...... (8) In the above formula, ...... (9) Y = Illinll (-L) -...
・(IQ)n β=in (coth±“-) ・・・・・・(+
+) 17.37 ... (12) In Fig. 9, RL is the load resistance, and when the circuit order n is an odd number, RL = I, and when n is an even number, as shown in Fig. 6. Magnetic field coupling coefficient M12% M2.3 s...
・M <7z-+3.71 together and M k4N spear
If each geometric coefficient is collectively expressed as g8, then BP
The characteristic impedance Z6C of the resonator constituting F is determined by the width W of the housing 1 and the diameter d of the resonant element, regardless of the center spacing of the resonant elements, so M, (+7) is expressed as 3) . However, 13wr: Pass band width f that gives the allowable ripple and ar
6: Actance loss f Lc between center frequency resonant elements 21 and 21 of BPF'
The actance loss between 2.2 U and 23 is LC2,3...
...The actance loss between 2n-1 and 21 is expressed as LC(
n-, n, □, and put them together as LeK, (no,
), and the center distance between the resonant elements 21 and 2a is C.
The center distance between d14.2 and 23 is ca,,l...
・The center distance between 21-5 and 27L is Cd (*-11,71
, and if we put this together and express it as Cd, (3)
From the formula, Cd de, (1o+1 = 0.3d+“11”...
・(14)54.6η k = 1 , 2 , ......n input = 300/l (GHz) = 300000Δ(MH
z) -...-(+6) Also, (4), (5) and (13)
From the equation, we get the following equation:

While substituting the required values for BWr and fa in equation (17), the geometric coefficient g or g obtained from equations (7) and (8)
? By substituting each value of t into equation (17), the actance loss between each stage ``K'', (K+Q) is determined, and this LQK, (K+
+) into equation (14), and the respective values of the diameter d of the resonant element, the width of the housing 1 (width of the resonator) W, the dielectric constant ε, the cut-off wavelength cleaning, and the wavelength of the transmission signal are expressed as ( 14) Substitute into the formula to find the center distance Cd formula (K and l) for each adjacent resonant element, and calculate the actual center distance of each resonant element stone or 21t by (
By matching Cd obtained from equation 14) with iK+1), the transmission characteristic of the BPF of the present invention can be made to be the above-mentioned transmission characteristic. Similar techniques can be used to achieve the objective when obtaining other transmission characteristics.

When the pass band is Chebyshev characteristic and the attenuation band is Wagner characteristic, the transmission characteristic of the BPF of the present invention is expressed by the following formula. X(+
In the case of 7h(x) = cos (n co8x)...
・"・(20)x)l, 7;, (x) = cosh (n cosh-'x)
・・=−(21)x=()
...... (22) BWr fa f Figure 11 is a characteristic curve diagram of the BPF of the present invention that reduces the above characteristics, where the horizontal axis is the transmission frequency t (MH2), the vertical axis is the attenuation ATT (dB), fn and f are the allowable ripple La
These are the upper and lower limit frequencies of the passband width Bwr that gives r.

The above is the most basic coupling configuration, but in this configuration, for example, when the load Q is high, the center spacing of the resonant elements becomes large, which may be disadvantageous in terms of mounting design.

FIG. 12 is a sectional view (BB sectional view in FIG. 13) showing the main parts of an embodiment that can eliminate such defects, and FIG.
In the A-A sectional view of Figure 2, 6Il and 6JI to 6Rn
-+) and 6a (n-r) are apertures, which are located between the adjacent resonant elements at each upper edge, each outer edge, and each lower edge of the housing l.
It is attached closely to the top wall, side wall, and bottom wall (there may be some gaps), parallel to the axial direction of the resonant element, and perpendicular to the side wall of the housing 1. FIG. 13 shows, for example, a resonant element 2. and 22 has an aperture 61. Although this example shows a case in which 6□ to 6IcK-1 are provided facing each other, one of them may be omitted and, for example, 6□ to 6IcK-1) may be provided on one side as shown in FIG. Even in this case, the upper edge of the aperture does not necessarily have to be brought into close contact with the earthen wall of the housing 1.

Now, the load Q when the throttle is not provided as shown in FIGS. 4 and 5 and the load Q when the throttle is provided as shown in FIGS. 12 to 14 are made to match each other. Resonant element 2. When adjusting each center interval of 27I to W@,
If a diaphragm is provided, the center spacing is Cd1)<,lxo,)
, the reactance loss between each resonant element is LOlK,
(H,), and the magnetic field coupling and coefficient between each resonant element are MiK, (H4), then these relationships are expressed by the following equation.

...(23) LQll<, 1 to 10 = 207!0gM1H
, (H++) ...... (24
)1, cJ, (ht M1 de, (do middle r) = IO”...
-(25) η is given by equations (15) and (I6).

MiK of formula (26), 4. ) and MK of equation (18)
(The aperture width can be approximately determined from the W orientation and the horizontal width (resonator width) of the housing 1 from the following equation.

M This material is 9 W (26) MiK bar) If the value of the aperture width obtained from the above formula is smaller than W/2, one on both sides as shown in Figure 13. A diaphragm is provided, and the value of D is W.
If it is larger than /2, the diaphragm 9 may be provided only on one side as shown in FIG.

The above example illustrates the case where an inductive aperture is provided, but the 15th
The main part sectional view (BB sectional view in Fig. 16) is shown in the figure.
As shown in the A-A sectional view of FIG. 15, the resonant element 2
The aperture 6.1 to 2n is brought into contact with the bottom wall of the housing wall at each opposing intermediate portion, that is, the housing wall and side wall on the ground side of the resonance element. When the radius -7 is provided, the electric field energy of the H1+ mode wave in the rectangular waveguide section of the BPP is concentrated and accumulated between the diaphragm and the earthen wall of the casing, and acts as a capacitive diaphragm. Therefore, when the magnetic field coupling amount (inductive coupling amount) between the resonant elements receives the canceling effect of the capacitive aperture, the coupling becomes tighter.

The embodiments shown in FIGS. 12 to 14 are suitable for forming a BPF with a high load Q because the degree of coupling can be made sparse by adjusting the inductive aperture width. On the other hand, the embodiments shown in FIGS. 15 and 16 are suitable for forming a BPF' with a low load Q because the coupling can be made dense by the capacitive diaphragm. Since the distance between the centers of the elements 21 and 21 and the length and width of the housing 1 can be kept constant and the degree of coupling between stages can be freely adjusted by adjusting the aperture width, this is not only advantageous in terms of design and manufacturing, but also improves transmission characteristics. You can get used to it.

FIG. 17 is a cross-sectional view of a main part (B-8 cross-sectional view in FIG. 18) showing another example of the coupling adjustment element, and FIG. 18 is a cross-sectional view taken along A-A in FIG.
In the cross-sectional view, 7. to 7n-1 are coupling adjustment screws, and the resonant elements 2. An earthen wall of the casing 1 in each middle part of the to 2 liters,
That is, it is formed so that it is aligned with a screw hole provided in the housing wall opposite to the open end of the resonant element, and the insertion length of the screw into the housing can be freely adjusted by operation from outside the housing. The figure shows an example where one screw is attached to each middle part of the resonant element, but it is also possible to attach multiple screws to each middle part as appropriate. Instead of mounting the screw parallel to the resonant element 2. The screws may be inserted in the direction perpendicular to the resonance elements from the upper side wall of the housing 1 in each of the opposed gaps 2 to 24, that is, the side wall of the portion near the open end of each of the resonance elements 2r to 2FL, or the screws may be inserted from one side wall. Instead of inserting the screws into the housing, the screws may be inserted from both side walls.

Since the part of the housing 1 between the resonant elements acts as a cut-off waveguide, when providing a coupling adjustment screw in this part, the coupling adjustment screw 7t (or 7
The electric field E concentrates on this screw according to the length of insertion of a to 7n a) into the housing, and the electric field strength becomes stronger. As the electric field strength becomes stronger, the magnetic field strength also increases and the magnetic field coupling becomes denser. . Therefore, when designing the BPF, the intervals between the resonant elements are determined so that the load Q is at its highest, that is, the magnetic field coupling coefficient is minimized, within the range allowed by the dimensions of the housing, and the coupling adjustment screws are operated from outside the housing. By changing the insertion length, the magnetic field coupling coefficient is increased appropriately to achieve the required load Q, and a frequency adjustment screw (not shown in the figure, but conventionally known) is installed opposite the open end of each resonant element. It is possible to create any load Q at any frequency by operating it from outside the housing. That is, by designing and manufacturing BPFs with the same shape and dimensions, it is possible to have any load Q at any frequency by adjusting from outside the housing.

FIG. 20 is a sectional view (B-8 sectional view in FIG. 21) showing the main part of another embodiment of the present invention, and FIG. 21 is a sectional view taken along A-A in FIG. 20.
[iii] In this embodiment, the apertures 611 to 62 (n-I) explained in FIGS. 12 to 14 are provided in the opposing gaps of the resonant elements 2 to 2n, and the apertures 611 to 62 (n-I) described in FIGS. The coupling adjustment screws 71 to 7shi1 shown in the figure are attached, and the coupling effects of the resonance element, the aperture, and the coupling adjustment screws can be used in combination, so the BPF
By miniaturizing B, it is almost independent of the size of the load Q.
It is possible to standardize the PF housing. In other words, the width and total length of the housing are
Since the PP can be configured and the coupling adjustment screw can be adjusted from outside the housing, adjustment is easy and good electrical characteristics can be obtained in a short time.

Fig. 22 is also a sectional view (BB sectional view in Fig. 23) showing the main parts of an embodiment in which a diaphragm and a coupling adjustment screw are also provided, and Fig. 23 is a sectional view taken along AA in Fig. 22. In this embodiment, a capacitive aperture 6. I to 63 (nt) in case 1
The resonance element 2 or 2 is installed in close contact with the earthen wall of
Connecting adjustment screw 7 from the top of the side wall of body 1 near the open end of a.
I to 771-1 are inserted perpendicularly to the resonant element, and both the aperture and the coupling adjustment screw are provided in areas where the electric field is strong, so the electric field is lower than in the embodiments shown in Figs. 15 and 16. Since the degree of concentration is increased and the coupling can be made denser, it is particularly suitable for configuring a BPF with a low load Q.

FIG. 24 is also a sectional view (second
5), and FIG. 25 is a sectional view taken along A-A in FIG.
Inductive shorting rods 811.8Jl to 811□,), 8
2. The coupling adjustment effect is almost the same as that of an inductive aperture. Although the figure shows an example in which two inductive shorting rods are provided, one or more than two inductive shorting rods may be provided as appropriate, and the amount of coupling is determined depending on the number of installed rods, the installation position, and the diameter.

Inductive shorting rod? At the same time as providing capacitive coupling adjustment screws 7 to 7 on the earth wall of the housing in the opposing gap of the resonant element.
By attaching L-, and adjusting its insertion length from outside the housing, the amount of inductive coupling of the inductive shorting rod can be freely adjusted and changed, and good electrical characteristics can be obtained.
F can be manufactured in a small size and economically. Note that the coupling adjustment function can be enhanced by attaching the capacitive coupling adjustment screw as centrally as possible on the top wall of the housing in the opposing gap between the resonant elements.

Next, we will discuss the input/output coupling circuit in the BPF of the present invention.

In the embodiments shown in FIGS. 4 and 5, the input/output coupling circuit elements 3° and 3° and
According to the experimental research results of the present inventor, the coupling in this embodiment is a combination of electric field coupling and magnetic field coupling, and the electric field coupling coefficient is M6% magnetic field coupling coefficient. M)4, and the composite coupling coefficient of both is JM, then lhM is given by the following equation.

(35) ... (28) ... (29) Cdo, +: input/output coupling circuit element 3° and resonance element 2.
Center distance η: (+5) and given by equation (16).

It is difficult to uniformly determine the coupling coefficient from the above theoretical formulas, and therefore it is also difficult to determine Cda and + from each of the above formulas, so it is practical to use Fig. 26 in combination to determine the required coupling coefficient. be. In the same figure, the horizontal axis is caa, +J
(mm) The % vertical axis is MGI/l, Mg and HM.

First, given the required transmission frequency f1 transmission wavelength input, the width w5 of the housing l, the output coupling path element 3°, and the center spacing Cd·,l of the resonant element 2I, we obtain (28) and (29).
), calculate the coupling coefficient and the size of MM, and from FIG.
Find tyj and calculate 0do4. The configuration of the input/output coupling circuit element 3n and the resonance element 2n side is composed of elements 3o and 2.
Since it is generally configured in exactly the same way as the 1st side, Cdo, + obtained as above is 3 W +
The center spacing is 1 and 2□.

The coupling of this input/output coupling circuit is a combination of electric field coupling and magnetic field coupling as described above, but the result is capacitive coupling as shown in FIG. This is often convenient when configuring a duplexer using BPF.

The input/output coupling circuit elements 3° and 3-IL+l in FIGS. 4 and 5 are connected to lumped constant circuit elements such as coupling capacitors or electrode plates that form distributed capacitance by facing the resonant elements 2I and 2n, respectively. In the case where the input/output coupling ports are configured as a capacitive coupling type as shown in FIG. 6, the input/output coupling capacitances ell, l and 07? An-*l) as the geometric coefficient g
, and g71, etc., in each of the above formulas, Xa, 1 and xicW+1) ' c, , and Cm,
+ total r) ') J) F41 conversion') 7 ctance, that is, the characteristic impedance Z of BPF' + CIL, tTLhr) normalized capacitance Xa, + and Xn, (nya υ: center of BPF' 06.I and On at angular frequency ω. (7L-u) Glue actance As shown in Figure 27, input/output coupling ports F11!r elements 3° and 3□O are set to have the same diameter as resonance elements 2. and 2rL. When provided with proper characteristics, a magnetic field coupling type input/output coupling circuit is constructed as shown in the equivalent circuit diagram in Fig. 28.Other symbols in Figs. 27 and 28 refer to Fig. 4. and the same as in FIG.

The amount of input/output coupling in this example, that is, the coupling coefficient M o and Mn in the characteristic impedance ZaC of the resonator constituting the BPF. +l) can be found by (33).

Moreover, the input/output coupling circuit element 3 when the width of the housing 1 is W. and 3□yu, and resonant element 2. The coupling loss of the input/output coupling transformer consisting of LWa, 7 and "n, (r
L+I) then −LW,,, =LWn, (***)= 20JOg
Ma, + (dB) ”= (34), element 3
Center distance between ° and 21 Cdc+, I and element 24 and 3 milk+
The center distance Ctln, tn+o of 1 is obtained by the following equation.

η is a value given by equations (15) and (16).

FIG. 29 is a sectional view (B-'B sectional view in FIG. 30) showing the main part of another embodiment of the present invention, and FIG.
In the A sectional view, in this embodiment, the human output coupling circuit elements 36 and 3□+1 (31Y, not shown, but having the same configuration as 3) are formed using strip lines, and the width thereof is Although it is difficult to determine the thickness, the distance between the centers of the opposing resonant elements 21 and 2n, etc. by theoretical calculation in a microwave circuit, it is possible to experimentally determine the required dimensions relatively easily. When a strip line is used as the input/output coupling circuit element as in this embodiment, the space factor is good, the whole can be made compact and economical, and a BPF with relatively high power and similar losses can be constructed. This is particularly effective when configuring a duplexer. Note that other symbols in FIGS. 290 and 30 are the same as in FIGS. 4 and 5.

FIG. 31 is also a cross-sectional view of a main part of another embodiment of the present invention (third
2), and FIG. 32 is a sectional view taken along A-A in FIG. 31. In this embodiment, the input/output coupling circuit element 3o
and 37N+ (3, FLu is not shown) formed using thin wire, and its diameter, resonant elements 2I and 2
Although it is not easy to theoretically calculate the distance between the centers and n, the required dimensions can be determined experimentally relatively easily. Although the thin wire in this example has a smaller power capacity than the strip line in the previous example, it has sufficient power capacity as an input/output coupling circuit element of a BPF used in general communication devices or duplexers. , the entire BPF can be formed compactly and economically, and it is easier to obtain than strip lines, and installation and adjustment are easier. Other symbols in FIGS. 31 and 32 are the same as in FIGS. 4 and 5.

In the embodiments shown in FIGS. 29 to 32, the input/output coupling circuit is formed into a capacitive coupling type by providing an input/output coupling circuit element made of a strip line or a thin wire with a polarity opposite to that of a resonant element. However, by providing the input/output coupling circuit element with the same polarity as the resonant element, a magnetic field coupling type input/output coupling circuit can be formed.

FIG. 33 is a cross-sectional view of main parts (third embodiment) showing another embodiment of the present invention.
(B-B sectional view in Figure 4), Figure 34 (A-A in Figure 33)
In the cross-sectional view, this embodiment (;L\te(input/output coupling circuit cord 3. and 37L+I(3ν11, 11L in the diagram)
\) is formed with a loop and each of the magnetic field coupling ports 3 is folded into squares, and the other symbols are the same +i as in FIGS. 4 and 5.

Input/output magnetic field coupling coefficient M in this example. , I and M
n, tn) is approximately given by the following equation.

f: Transmission frequency (Hz) Sq: Surface impact where the input/output coupling loop cuts the magnetic flux (ml) μ
, = 4π x 10-'' (Hen 1) A) r: Position radius of the loop, that is, J between the inside 70 of the loop and the inside and outside axes of the resonant element facing the loop!: Bottom wall of the housing 1 in the loop from
Highest input/output magnetic field coupling coefficients Ma, 1 and M7! ,
To adjust rn+, ), it is necessary to adjust the position radius r of the loop and the area Sq where the loop cuts the magnetic flux.If the loop is made of a thin plate, it is easy to change the area sq. If the capacity is large and the loop is made of a thick plate, it is not easy to finely adjust the area Sq.
.

In such a case, the input/output magnetic field coupling coefficient can be finely adjusted by using a rotating loop as shown in cross section in FIG. The same figure (in this case) shows the end wall of the casing, 4. is the input/output coaxial terminal, outer conductor 9, inner conductor 1
The base of the external conductor 9 is rotatably inserted into a mounting hole bored in the end wall of the housing 1. 1
2 is a ring-shaped presser metal fitting with a groove f provided on its inner peripheral surface.
2. The flange-shaped protrusion provided on the outer periphery of the base of the external conductor 9 is loosely fitted. 3. denotes each input/output coupling port S element consisting of a loop, the inner end of which is fixed to the inner end of the inner conductor 10 with a set screw 13, etc., and the outer end of the loop is fixed to the base of the outer conductor 9 by welding or the like. be. When the set screw 14 is loosened and the input/output coaxial terminal 4# is rotated around the central axis, the loop 3- also rotates together with the terminal 46, and the set screw 1
4 T- When tightened, the inner surface of the groove provided on the inner circumferential surface of the ring-shaped presser metal fitting 12 is crimped against the collar-shaped protrusion, and the input/output coaxial terminal 4. and loop 3. is fixed at an arbitrary rotation angle.

Therefore, as shown in FIG. 36(a) is a cross-sectional view similar to that in FIG. 33, and as shown in FIG. The rotation angle is θ.

Since the input and output magnetic field coupling coefficients in the case of tosult and θ1=0 are Mo and r, the input and output magnetic field coupling coefficient Ma6. t is Men, I = M
a, + 00 days θ "...
... (37), and the area where the loop cuts the magnetic flux can be easily finely adjusted.

In each of the above embodiments, since the resonant elements 2° to 2n are all arranged in a row, the overall shape becomes horizontally elongated, which may be disadvantageous in terms of mounting design. Furthermore, in the case of passive transmission characteristics, coaxial cables or strip lines and connections between these transmission lines and the resonant elements are used to indirectly couple the resonant elements separated by two or an integral multiple of the resonant elements. Since a coupling loop or a capacitive element is required, the indirect coupling circuit becomes relatively complex and long.

Fig. 37 is a sectional view (BB sectional view in Fig. 38) showing an embodiment configured to eliminate such drawbacks;
FIG. 8 is a sectional view taken along the line A-A in FIG. 37, where 1 is an electromagnetic shielding case, 15 is a partition made of a conductor, and the partition wall partitions the inside of the case 1 into a U-shape. 21...2
．． m% 2M'l++...2rL is a resonant element made of a rod-shaped conductor, and is disposed in a U-shape in a casing 1 partitioned into a U-shape. 3° and 3? Lya is an input/output coupling circuit element; 4. and 47L-r+ are input/output coaxial terminals. Also in this embodiment, the side wall of the housing 1 and the partition wall 15
interval W and resonant element 2. When the diameter d of the resonant element 2. or 2
The axial length of rL is selected as the electrical length of the resonant wavelength, and if the width W and diameter d are relatively larger than the resonant wavelength, the resonant element 2. The axial length of 2rL to 2rL is made to be appropriately shorter than the resonant wavelength of 4 in terms of electrical length.

The basic coupling action between the resonant elements also differs from the embodiment shown in FIGS. 4 and 5 (hereinafter referred to as the first This is exactly the same as in Example (abbreviated as Example). As shown in FIG. 39, the electromagnetic field between the resonant elements 2m and 2a++ in the folded portion is disturbed by the end of the partition wall 15, so it is difficult to perform accurate physical examination or find a theoretical calculation formula. Although it is difficult, in FIG. 39, the resonance elements 21 and 21□
Assuming that the cutoff wavelength is approximately determined by the electromagnetic field mode on the right side of the drawing from the line A-A that connects the centers of -A position symmetrical to the side wall of the housing 1 with the A line as the center (indicated by a dashed line)
By imagining a side wall and assuming that the electromagnetic field mode between this virtual side wall and line A-A is symmetrical with the electromagnetic field mode on the right side of line A-A, resonant elements 2 and 2,
(1) Actance loss, magnetic field coupling coefficient, etc.
It can be approximately determined using the formula or formula 5).

The resonant element spacing other than the spacing between the resonant element 24 and 2'FI'L+1 can be changed to adjust the magnetic field coupling coefficient as described in the first embodiment;
The center spacing of 2 and 2 almost matches W, and if W is determined (the thickness of the partition wall 15 is greater than W), 2 and 2a++
The center spacing of the BPF is also determined automatically and cannot be changed, which imposes constraints on the design of the BPF.

In order to eliminate this restriction, the end of the partition wall 15 is regarded as the inductive aperture described in connection with FIG. 14, and by changing the ratio of the aperture width and W, the resonant elements 2nL and 26. .

The degree of coupling between them can be adjusted. Note that the partition wall 15 separates the resonance elements 2m- and 2. Direct connection between W1 and C and 2
Since it is necessary to prevent direct coupling between □ and 2.1'tL, there is a limit to the aperture width.
Adjustment of the degree of coupling between WLs 1 and 2 and WL 5 is limited to adjustment in the direction from the degree of coupling at the maximum aperture width to sparse.

Now, when it is assumed that the aperture consisting of the end of the partition wall 15 does not exist, that is, when it is assumed that D=w, the actance loss L between the resonant elements 2□ and 2yyi4I is
i y , lW (+ r) can be found from the same way as equation (23), and the required aperture width can be found by substituting the obtained values into equations (24) or 26). I can do it.

As shown in a sectional view of the main part in FIG. 40, a capacitive coupling adjustment screw 71. By inserting L into the housing 1 and changing its insertion length, the resonant element 2. , the degree of coupling between L and 2m and 5 can be adjusted in the direction where it becomes denser, so by using both the adjustment by the aperture width and the adjustment by the coupling adjustment screw 7□, the coupling between other resonant elements can be improved. Similar to the degree adjustment, it is possible to freely adjust the degree of connection between 2□ and 21, and the degree of openness.

FIG. 41 is a sectional view showing another configuration of the folded portion (No. 42).
FIG. 42 is a cross-sectional view taken along A-A in FIG. and 2% or 1, and resonant element 2. A coupling hole 16 is formed centered at a point on the line of intersection with the partition wall 15 in the plane containing the respective central axes of 2 and 1. ) to the center of the coupling hole 16 is selected to be the length of an arbitrary harmonic of the transmission signal, for example, from the node of the third harmonic to the voltage wave antinode, the third harmonic IH at the periphery of the coupling hole 16 is selected. Therefore, the resonant elements 21 and 2□
Since there is no coupling of the third harmonic between , and, and only the fundamental wave is coupled through the coupling hole 16, a PF with good harmonic characteristics can be constructed. To determine the diameter of the coupling hole 16, first, let the required magnetic field coupling coefficient be Mm, (2nd edition), then the coupling susceptibility bLyu, (,
Wl and ) are therefore: However, r: Radius of the resonant element The state in which the center distance between the resonant elements 2□ and 27jL++ is kept at one foot by adjusting the size of the coupling hole 16, which is determined from equation (40), to an appropriate size. You can freely adjust the degree of coupling.

Next, the input/output coupling circuit element 3 in the embodiment shown in FIGS. 37 and 38 (hereinafter abbreviated as the second embodiment)
゜ and 3n Ya 1 can also be implemented by applying all the various embodiments explained in the first embodiment as they are, and between the resonance elements 2FrL and 2ffl+1 (39
) are coupled through the coupling hole 16, this opposing section is excluded (all resonant elements 2.n and 211 are connected in the opposing section of the
As in the case of the first embodiment, a coupling adjustment element of any one of a diaphragm, a coupling adjustment screw, or an inductive shorting rod is interposed in the opposing sections of the elements without interposing a coupling hole therebetween. By interposing these in appropriate combinations, the bonding characteristics can be freely adjusted.

The method of making the transmission characteristics of the first embodiment into the required characteristics based on the geometric coefficients of the standardized image pass filter is to apply this directly to the second example to obtain the required transmission characteristics. I can do it.

However, for example, a BPF with a Chebyshev characteristic in the passband and a Wagner characteristic in the attenuation range has poorer attenuation characteristics than a 7I8i polar type BPF, so the polarization of the BPF of the present invention will be explained below.

FIG. 43 is a cross-sectional view (C-C cross-sectional view in FIG. 44) showing an example in which the circuit order is 6 in the second embodiment and is formed to have polar characteristics, and FIG. 44 is A in FIG. -A sectional view, FIG. 45 is a BB sectional view of FIG. 43, and in each figure, 1 is the housing, 2. None\2BL is a resonant element arranged in a U-shape. 3° and 3□1 input/output coupling circuit elements, 4. and 4 are input/output coaxial terminals, 15 is a partition wall, and 17 is an indirect coupling hole, the center of which is approximately at an appropriate height on the intersection line of the partition wall 15 with a plane containing the central axes of the resonant elements 21 and 2&. . In place of the indirect coupling hole 17, an inductive coupling element such as a loop may be used. 18 is a capacitive indirect coupling element, for example, a lumped constant circuit element such as a capacitor or a resonant element 2. A counter electrode plate forming a coupling force between the resonant element 2. and a counter electrode plate forming a coupling capacitance therebetween, which are connected via a connecting line. The capacitive indirect coupling element 18 is inserted through a hole approximately centered at an appropriate height on the intersection of the partition wall 15 and a plane containing the center axes of the resonant elements 2a and 25, and is insulated from the partition wall 15. It is shaped to be held and supported.

In this example, the case where the circuit order is selected as 6 is illustrated, but it is of course possible to increase or decrease the circuit order as appropriate. The configuration of the output coupling circuit element and the like are the same as those in the first and second embodiments.

Fig. 46 is an equivalent circuit diagram of this embodiment, where C□ and C1 are input/output coupling capacitances, R, to R6 are resonant circuits, M, 1
．． Or M, 6 is the crotch magnetic field coupling coefficient, Mlh is the indirect magnetic field coupling coefficient, and OaG is the indirect coupling capacitance.

One section of the resonant circuit in FIG. 46, for example, the R5 section, can be rewritten as shown in FIG. 47.

In the figure, LR is the inductance in the resonant circuit, C is the capacitance in the resonant circuit, and MITL is the magnetic mutual inductance.

As shown in FIG. 48, the equivalent circuit diagram shown in FIG. 47 can be divided into a circuit portion R that performs a resonance action and a circuit portion PC that constitutes a phase circuit. In the same figure, CI, l
is the capacitance of the equivalent phase circuit, and the other symbols are the same as in FIG. 47. A circuit portion P that constitutes a phase circuit.

The [F] matrix is: (41) However, as shown in Fig. 49, the characteristic impedance z0 and the characteristic admittance Ya are 1, and the distance is the electrical length and the tube wavelength λJ is The [F'l matrix of cable No. 4 is...
(42) If the equation (42) is established in the equation (41), the right side of the equation (41) becomes equivalent to the r2A cable, and FIG. 46 can be rewritten as shown in FIG. 50. In the figure, P to R7 are 90' phase paths, and other symbols are the same as in FIG. 46.

During transmission through the main circuit between points a and d in Figure 50, the resonant signal produces a phase difference of +90ゝIn circuits R and R6, a phase difference of ±9σ×6=±540′ is generated, and both routes are t:9o” phase rotation +9C1”X5=+
Since a phase difference of 450" is generated, the result is +450
'Produces a phase difference of ±5401. Therefore 450
' +540°=990@-36 (1@X 3=-90
° or 450°-540'=-90'. That is, the attenuated signal produces a phase difference of 190° both at frequencies higher than the resonance frequency and at frequencies lower than the resonance frequency.

Points a and d are indirectly coupled (magnetic field coupling, i.e., current coupling) by coupling holes or loops indicated by 17 in FIGS. 43 and 44 and Ml and & in FIGS. 46 and 50. Therefore, through this indirect coupling element, a
The signal transmitted from point to point d has a phase difference of +90°, and at point d, an attenuated signal that is transmitted through the main circuit and reaches point d, and an attenuated signal that is transmitted through an indirect coupling element and reaches point d. The amount of coupling between the resonant circuits R5 and R& through the indirect coupling hole (or loop) 17 varies closely in relation to the load Q of the BPF; The amount of attenuation that occurs while the signal at the frequency that is intended to cause an attenuation pole (hereinafter abbreviated as attenuation pole signal) is transmitted through the main circuit from point 6 to point d by setting the magnitude of By determining the diameter of the indirect coupling hole 17 or the area where the loop cuts the magnetic flux so that the coupling attenuation amounts in the indirect coupling elements are equal to each other, an attenuation pole can be generated at a desired frequency position.

Next, while transmitting the main circuit between point b and point C, the resonance signal is transmitted in phase circuit P or R5 +90'X3=
+270° phase difference, whereas the attenuated signal is ±9♂×4=t in the resonant circuit R
The phase circuit P generates a phase difference of h36σ.

or at t +90°X '3= +270"
This results in a phase difference of +27♂+
3601 phase difference, i.e. +27σ+360 = +
63r; -360@X2=-9♂ or +2
A phase difference of 70" -360°=-90° is produced. On the other hand, a capacitive indirect coupling element (the fourth
3 and 18) in Fig. 45, a phase difference of +90'' is generated, so there is a difference of 1 at point C between the attenuated signal transmitted through the main circuit and the attenuated signal transmitted through the capacitive indirect coupling element.
This produces a phase difference of 80°. Therefore, by determining the coupling capacitance of the capacitive indirect coupling element so that the amplitudes of the attenuation pole signal transmitted through the main circuit and the attenuation pole signal transmitted through the capacitive indirect coupling element are equal to each other at point C, the target frequency can be determined. Attenuation poles can be generated at certain positions.

In addition, FIG. 51 is an equivalent circuit diagram of the indirect coupling element 17 consisting of a coupling hole or loop, etc., and FIG. 52 is an equivalent circuit diagram of the indirect coupling element 17 consisting of a coupling hole or a loop.
In the equivalent circuit diagram of 8, the respective IF, l, and lFc1 matrices are as follows: bL = induced septance Imaginary part R1: Real part expressed as a complex number The phase characteristic of the indirect coupling element consisting of the coupling hole 17 or loop is: eL: The phase characteristic of the phase angle capacitive indirect coupling element 18 is ec=tan-
'-'' ...... (48) θ6 If the two phase angles bL and Xc are sufficiently large, the phase angles θ and θ of elements 17 and 18 will be the same value, for example, S = +00
In this case, θ, =88.85'Xc:100
If θ. = 88.85".

When using a magnetic field coupling element such as a coupling hole or a loop as an indirect coupling element, the phase relationship is based on the current, and the circuit is a parallel circuit (admittance circuit) shown in Figure 5+.
When a capacitive coupling element is used, the phase relationship is based on voltage, and the circuit is handled using a series circuit (impedance circuit) shown in FIG.

Even if the indirect coupling element 17 shown in FIGS. 43 to 46 is replaced with a capacitive coupling element, and the equivalent circuit is shown in FIG. 53, the resonance circuits R9 and R6 are indirectly coupled by the indirect coupling capacitance Crb. Similarly, it can be made to have polar characteristics. Also, as shown in the equivalent circuit in Fig. 54, R5
and R6 and between R3 and R9 may be indirectly coupled by magnetic field coupling (M, & and Mar). In particular, in this case, the input/output coupling circuit is made of a rod-shaped conductor as shown in Fig. 4. By using a transformer configuration in which the input/output coupling circuit element is provided with polarity opposite to that of the resonant element, high power BP
F can be formed.

In FIGS. 53 and 54, only one of the resonant circuits R5 and R6 or R2 and R5 may be indirectly coupled, and generally the number of resonant circuits to be indirectly coupled can be arbitrarily selected. Generally speaking, the resonant circuits to be indirectly coupled are two resonant circuits or an integer multiple of the resonant circuits separated from each other.

Furthermore, all indirect coupling elements are formed using either magnetic coupling elements or capacitive coupling elements, and any number of indirect coupling elements among the plurality of indirect coupling elements are formed using magnetic coupling elements, and the remaining indirect coupling elements are The present invention can also be implemented by configuring the capacitor to be formed using a capacitive coupling element.

The transmission characteristics of the BPF of the present invention in which the pass band has Chebyshev characteristics and an attenuation pole is generated in the attenuation region as described above can be obtained from the following equation.

When the order n is an odd number, When the order n is an even number, ...... (53) f-: Frequency that produces an attenuation pole In the equation (,50), 1 and lL mean the imaginary part, (
Re in formula 51) means taking the real part.

In each of the above formulas, p, 1,000 css, m; :1
(19) which expresses the transmission characteristic where one pass band has the Chebyshev characteristic and the attenuation zone has the Wagner characteristic. Figure 55 is a characteristic curve diagram of the BPF of the present invention having the above-mentioned polar type characteristic, and f-j is the frequency that produces the attenuation pole, and the other signs are the first
It is the same as Figure 1.

Of course, the first embodiment can also have polar characteristics, but in this case, since the resonant elements are arranged in a row, the intervals between the resonant elements to be indirectly coupled are large.
A coaxial cable or strip line is required as an indirect coupling circuit, and a magnetic coupling element or capacitive coupling element is required as a coupling element between the coaxial cable or strip line and the resonant element, so the configuration is relatively complex and large. There is a drawback.

In each of the above embodiments, a resonant element (
A dielectric resonator in which a dielectric material is interposed between an internal conductor) and a housing (external conductor) may be used; in this case, since the coupling between the legs in the BPF of the present invention is magnetic field coupling, the dielectric constant of the dielectric material is Since the magnetic permeability can be treated as 1 in the same way as in the case of air, design and manufacture is as easy as in the case of air.

The inventor has developed a BP of order 4.6 and 8 for each example.
As a result of making a prototype of F and comparing the theoretical values and actual values regarding transmission characteristics, dimensions of component parts, etc., the two agreed extremely well, proving the correctness of the theoretical configuration of the present invention.

As is clear from the above explanation, in the BPF of the present invention, the distance between the open end of the resonant element and the housing wall can be made relatively large, so it has excellent voltage resistance characteristics and is not affected by changes in ambient temperature The optical characteristics are stable and good, the entire structure can be made compact, and various transmission characteristics can be provided while maintaining the overall shape and dimensions.

[Brief explanation of the drawing]

1 and 2 are cross-sectional views showing a conventional combline type bandpass filter, FIG. 3 is an equivalent circuit diagram thereof, and FIGS. 4 and 5 are cross-sectional views showing an embodiment of the present invention. , Figure 6, Figure 28
Figures 46, 53, 548, 19 and 36 are explanatory diagrams of the operation of the filter of the present invention, Figures 9 and 1.
Figure 0 is an explanatory diagram of the design method of the filter of the present invention, Figures 11 and 55 are curve diagrams showing examples of transmission characteristics of the filter of the present invention, Figures 12 to 18, and Figures 20 to 55. Figure 25,
29 to 34 and 39 to 42 are sectional views showing essential parts of other embodiments of the present invention, and FIG. 26 is a diagram for explaining the input/output coupling circuit of the filter of the present invention. Curve diagram, 2nd
7, 37, 38, 43 to 45 are cross-sectional views showing other embodiments of the present invention, and FIG. 35 shows an example of the input/output coupling circuit element of the filter of the present invention. Sectional view, No. 47
Figures 52 to 52 are equivalent circuit diagrams for explaining the operation of the filter of the present invention, in which 1: housing, 2 and 21 to 2□: resonant element, 3.3° and 3□, 1 = input/output stand circuit element, 4.
4゜ and 4□, : input/output coaxial terminal, 5: electrode plate-61
1 and 6kol to 6trn-r) and 62C1-υs
6/or 6□-1=aperture, 7. to 7n-1' coupling adjustment screw, 81- and sat to B/In-tl and 8 Q (n-ly: inductive shorting rod, 9: outer conductor, 1
0: internal conductor, 11: insulator, 12: presser fitting, 13 and 14: set screw, 15: partition wall, 16: coupling hole, 17 and 18: indirect coupling element. Figure 1 Figure 2 Figure 3 Figure 4 Figure 5 Figure 6 Figure 7 Figure 8 Figure 9 Figure 10 Figure 11 Figure 12 Figure 13 Figure 14 Figure 15 Figure 16 Figure 19 Figure 17 Figure 18 Figure 20 Figure 29 Figure 30 / 0 Figure 31 Figure 32 / 0 Figure 33 Figure 34 Figure 35 Figure 36 [A] (B) Figure 37 Figure 39 Figure 40 Figure 41 Prisoner Figure 42 Figure 44 Figure 43 Figure 45 Figure 46

Claims (28)

[Claims] (1) A plurality of resonant elements having an electrical length and an axial length of approximately A of the resonant wavelength and arranged in a line with the same polarity, and an input/output coupling circuit element, and the plurality of resonant elements Let each center spacing be Lo, (to = -201ogM (to +1>CdX, (X+1): to the center spacing of the resonant element = 1, 2, ... nn: order of the filter d: Diameter of the resonant element W: Width LQ of the housing (outer conductor of the resonator)
2W g Cleaning: Cutoff wavelength Input: Wavelength of transmission signal ε: A combline type bandpass filter characterized by determined by dielectric constant. (2) A plurality of resonant elements having an axial length approximately equal to the resonant wavelength in terms of electrical length and arranged in a line with the same polarity; It is provided with an indirect coupling circuit that indirectly couples the resonant elements separated by a number of resonant elements, and an input/output coupling circuit element, and the distance between the centers of the plurality of resonant elements is Cd, (to (vl): Center spacing of resonant elements = 1.
2,...n n: Order of the filter d: Diameter of the resonant element W: Width of the housing (external conductor of the resonator) [, C, (subtracted): Glue actance between the resonant elements loss M
, (Kt+): magnetic field coupling coefficient between resonant elements λG=2W
g ＾C: Cutoff wavelength input: Wavelength of transmission signal ε: A combline type bandpass filter characterized by being determined by dielectric constant. (3) Has an electrical length that is approximately the same as the resonant wavelength, and has the same polarity? a plurality of resonant elements arranged in a line; a coupling adjustment element provided in each opposing gap between the plurality of resonant elements;
and an input/output coupling circuit element, and the distance between the centers of each of the plurality of resonant elements is Cd, (◆I): the distance between the centers of the resonant elements=1. 2
,・Fist...n n: Order of the filter d: Diameter of the resonant element W: Width Lc of the housing (outer conductor of the resonator), (K+ll: Actance loss M between the resonant elements, ( →, ): Magnetic field coupling coefficient between resonant elements c = 2
W/'H A combline type bandpass filter characterized by: cutoff wavelength input: wavelength of transmission signal ε: determined by dielectric constant. (4) A plurality of resonant elements that emit an axial length approximately equal to the resonant wavelength in electrical length and are arranged in a line with the same polarity, and two or an integral multiple of the resonant elements of the plurality of resonant elements are arranged in a row with the same polarity. an indirect coupling circuit that indirectly couples the resonant elements separated by a plurality of resonant elements; a coupling adjustment element provided in each opposing gap of the plurality of resonant elements; and an input/output coupling circuit element; Let the center spacing of each resonant element be Cd, (B1) = -20OgMK, (Kn
) Cdx, Or++): Center spacing of resonant element = 1.
2,...n n: Order of the filter d: Diameter of the resonant element W: Width of the casing (outer conductor of the resonator) LQX, (A, S: Actance loss M between the resonant elements)
H, (Kya,): c=2W to magnetic field coupling coefficient between resonant elements
, /H λC: Cutoff wavelength λ: Wavelength of transmission signal ε: A combline type bandpass filter characterized by being determined by dielectric constant. (5) A signal transmission line formed by a plurality of resonant elements having an electrical length and an axial length approximately equal to the resonant wavelength and having the same polarity and arranged in a U-shape, and the plurality of resonant elements. in,
A combline type band-pass filter characterized in that it includes a conductor partition provided between an outgoing path and a returning path (excluding the folded portion), and an input/output coupling circuit element. (6) A signal transmission path formed by a plurality of resonant elements arranged in a U-shape with the same polarity and having an axial length of approximately A of the resonant wavelength in electrical length, and the plurality of resonant elements. in,
Excluding the folded part ・The conductor partition wall provided between the outgoing path and the returning path, and two of the plurality of resonant elements that are in a cascade connection relationship.
1. A combline type band-pass filter comprising: an indirect coupling element that indirectly couples resonant elements separated by a number of resonant elements or an integral multiple of the number of resonant elements; and an input/output coupling circuit element. (7) A signal transmission path formed by a plurality of resonant elements arranged in a U-shape with the same polarity, emitting an axial length approximately equal to the resonant wavelength in electrical length, and the plurality of resonant elements. in,
A conductive partition wall provided between the outgoing path and the incoming path excluding the folded portion, a coupling adjustment element provided in each opposing gap in the cascade connection direction of the plurality of resonant elements, and an input/output coupling circuit element. Features a combline type bandpass filter. (8) A plurality of resonant elements having an electrical length and an axial length approximately equal to the resonant wavelength and having the same polarity and arranged in a U-shape, and a signal transmission line formed by the plurality of resonant elements. in,
A conductive partition wall provided between the outgoing path and the incoming path excluding the turning portion, and resonant elements that separate two resonant elements in a cascade connection among the plurality of resonant elements or an integral multiple of the resonant elements. A combline type bandpass filter comprising: an indirect coupling element for coupling; a coupling adjustment element provided in each opposing gap in the cascade connection direction of the plurality of resonant elements; and an input/output coupling circuit element. vessel. (9) A plurality of resonant elements having an electrical length approximately equal to the length of the resonant wavelength and arranged in two rows with the same polarity, and a plurality of resonant elements arranged between the rows of the plurality of resonant elements. a conductive partition wall, a coupling hole bored in one end of the conductor partition wall and coupling mutually opposing resonant elements at the ends of the plurality of resonant elements arranged in the two rows; A combline type bandpass filter characterized by comprising an input/output coupling circuit element. (10) A plurality of resonant elements that emit an axial length approximately equal to the resonant wavelength in terms of electrical length and are arranged in two rows with the same polarity; a coupling hole bored in one end of the conductor partition wall for coupling mutually opposing resonant elements at the ends of the plurality of resonant elements arranged in the two rows; Among the plurality of resonant elements, an indirect coupling element that indirectly couples two or an integral multiple of the resonant elements in a cascade connection with each other, and an input/output coupling circuit element are provided. A combline type bandpass filter featuring features. (11) A plurality of resonant elements having an electrical length and an axial length of approximately A of the resonant wavelength and arranged in two rows with the same polarity, and a plurality of resonant elements arranged between the rows of the plurality of resonant elements. a coupling hole bored in one end of the conductor partition wall for coupling mutually opposing resonant elements at the ends of the plurality of resonant elements arranged in the two rows; It is characterized by comprising a coupling adjustment element and an input/output coupling circuit element provided in each of the opposed gaps in the cascade connection direction of a plurality of resonant elements, except for the opposed gap in which the coupling hole is provided. Combline type bandpass filter. (12) A plurality of resonant elements having an electrical length approximately equal to the resonant wavelength and arranged in two rows with the same polarity, and a plurality of resonant elements arranged between the rows of the plurality of resonant elements. a coupling hole bored in one end of the conductor partition wall for coupling mutually opposing resonant elements at the ends of the plurality of resonant elements arranged in the two rows; An indirect coupling element that indirectly couples two or an integral multiple of two resonant elements in a cascade connection among the plurality of resonant elements, and A combline type bandpass filter comprising a coupling adjustment element and an input/output coupling circuit element provided in each of the opposed gaps except for the opposed gap in which the coupling hole is provided. (13) The actance loss between the resonant elements is LO, (
For mains: Reactance loss Cd -(K++): Center spacing of resonant element k = 1
, 2, ......n n: Order of bandpass filter d: Diameter of resonant element W: Width of the housing (outer conductor of the resonator) ^. =2w/i C: cutoff wavelength λ: wavelength of transmission signal ε: determined by dielectric constant The combline type bandpass filter according to any one of claims 1 to 12. (14) The center distance of the resonant element excluding the folded portion is Lc
, (K+Linny 201OEM)c, (IcsuI)c
d'r(H+'>' center spacing of resonant element = 1,
2, ......nn: Filter order d: Diameter of the resonant element W: Width of the housing (outer conductor of the resonator) TJQK, (Niya): Actance loss between the resonant elements M% , (H+r): magnetic field coupling coefficient between resonant elements ζ=
2W/T Bottom: Cutoff wavelength Input: Transmission signal wavelength ε: Determined by dielectric constant The combline type bandpass filter according to any one of claims 5 to 12. (15) The input/output coupling circuit element is composed of rod-shaped conductors facing each other with opposite polarity to the first and final stage resonant elements, and the input/output coupling capacitance formed between the opposing resonant elements is Ga, +: input/output Coupling capacitance zo: Characteristic impedance of the bandpass filter ω1: Center angular frequency Xo, + of the bandpass filter: 06. !
A combline type bandpass filter according to any one of claims 1 to 12, which is determined by a normalized reactance of . (16) The input/output coupling circuit element is composed of bar-shaped conductors facing each other with the same polarity as the first and final stage resonant elements, and the magnetic field coupling coefficient between them and the opposing resonant elements is Lwa, -F- M,,I= IO Me,r: Magnetic field coupling coefficient Lwa, +: Combline type bandpass filter according to any one of claims 1 to 12, determined by input/output coupling loss. (17) The input/output coupling circuit element is composed of rod-shaped conductors facing each other with opposite polarity to the first and final stage resonant elements, and the input/output coupling coefficient force (, M9.: input/output coupling coefficient M , : Electric field coupling coefficient M - Magnetic field coupling coefficient 06 m, 1 : Center distance W between the input/output coupling circuit element and the resonant element facing it: Width d of the casing (outer conductor of the resonator): Width of the resonant element Diameter included, = 2w, /i λC: Cutoff wavelength included: Wavelength of transmission signal 7 ε: Determined by dielectric constant Combline type bandpass filter according to any one of claims 1 to 12 . (18) The combline type bandpass filter according to any one of claims 1 to 12, wherein the input/output coupling circuit element comprises first-stage and final-stage resonant elements and opposing strip lines. (19) The combline type bandpass filter according to any one of claims 1 to 12, wherein the input/output coupling circuit element comprises first-stage and final-stage resonant elements and thin wire conductors facing each other. . (20) The combline type bandpass filter according to any one of claims 1 to 12, wherein the input/output coupling element comprises first-stage and final-stage resonant elements and fixed loops facing each other. (21) Claim 1 in which the input/output coupling circuit element comprises first-stage and final-stage resonant elements and rotating loops facing each other.
The combline type bandpass filter according to any one of Items 1 to 12. (22) Claims 3, 4, and 7, in which the coupling adjustment element is an inductive aperture provided in parallel with the resonant element;
The combline bandpass filter according to any one of Items 8, 11, and 12. (23) Claims 3, 4, and 7 in which the coupling adjustment element is a capacitive aperture provided perpendicularly to the resonant element.
8. The combline bandpass filter according to any one of Items 8, 11, and 12. (24) The comb according to any one of claims 3, 4, 7, 8, 11, and 12, wherein the coupling adjustment element is a variable insertion length capacitive screw. Line type bandpass filter. (25) Claims 3, 4, and 7 in which the coupling adjustment element comprises an inductive throttle and a variable insertion length capacitive screw.
8. The combline bandpass filter according to any one of Items 8, 11, and 12. (26) Claims 3, 4, and 7 in which the coupling adjustment element comprises a capacitive diaphragm and a variable insertion length capacitive screw.
8. The combline bandpass filter according to any one of Items 8, 11, and 12. (27) Clauses 3, 4, 7, and 8, in which the coupling adjustment element comprises an inductive shorting rod and a variable insertion length capacitive screw;
The combline bandpass filter according to any one of Items 11 and 12. (28) The combline bandpass filter according to any one of claims 1 to 12, wherein the resonant element is a component of a dielectric resonator.