JPH1141931A - Power converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate the necessity of installing rush current suppression circuits in all cell inverters at the time of initial charging and thereby reduce the number of circuit constituent elements by turning on AC power supplies for the cell inverters having no rush current suppression circuits and then starting an operation after completion of charging of capacitors of all the cell inverters. SOLUTION: First, only a power switch 2a is turned on and cell inverters 4U1, 4V1, 4W1 having rush current suppression circuits are initially charged. Nextly, with a power switch 2b being in an off state, the remaining cell inverters 4U2a-4W3a are charged using the initially charged cell inverters 4U1, 4V1, 4W1. By this method, it is not necessary to necessarily install rush current suppression circuits in all the cell inverters at the time of initial charging.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power converter capable of obtaining a high-voltage polyphase AC output.

[0002]

2. Description of the Related Art The configuration shown in FIG. 40 is known as a power converter capable of obtaining a high voltage output.
The operation of the power converter with the configuration of PWHammond “A New
Approach to Enhance Power Quality for Medium Volta
ge AC Drive ", IEEE trans.on IA, 1997.

In FIG. 40, a primary winding of a transformer 3 having nine sets of secondary windings is connected to a three-phase AC power supply 1 via a power switch 2 such as a contactor or a circuit breaker for turning on and off the AC power supply. Is connected to the secondary winding of the transformer 3.
A phase multiplex power converter 4 is connected, and a load 5 such as an AC motor is connected to the three-phase multiplex power converter 4.

A three-phase multiplex power converter 4 is a single-phase multiplex converter 4 in which three cell inverters are connected in series.
U, 4V and 4W are connected by Y connection. Cell inverter 4U1, 4U2, 4U3, 4V1, 4V2, 4V
Reference numerals 3, 4W1, 4W2, and 4W3 are basic components of the three-phase multiplexing conversion device 4.
Shown in

In FIG. 41, 101R, 101S, 1
01T is an AC input terminal of a three-phase power supply, 102 is a known diode converter composed of six diodes D1 to D6, 103 is a filter capacitor for smoothing DC power, and 104 is four transistors Q1 to Q4. , And 105P and 105N are AC output terminals of the inverter 104.

Reference numeral 106 denotes an inrush current suppression circuit at the time of initial charging comprising a resistor R and a short-circuit switch SW. The inrush current to the capacitor 103 when the AC power is turned on at the start of operation is suppressed by the resistor R. When charging of the battery 103 is completed and a DC voltage is established, the short-circuit switch SW is turned on to short-circuit the resistor R. It is an unnecessary auxiliary circuit during operation, but an indispensable circuit.

The three-phase alternating current input to the terminals 101R, 101S, and 101T is converted into direct current by the diode converter 102 and smoothed by the filter capacitor 103, so that a substantially constant direct current voltage Vc is obtained. The following voltage is obtained between the output terminals 105P and 105N by the combination of the transistors of the inverter 104 which are turned on and turned on two by two at the same time.

Q1 and Q4 are on: + Vc Q1 and Q2 are on or Q3 and Q4 are on: 0 Q2 and Q3 are on: -Vc As described above, the cell inverter of FIG.
It is possible to obtain three types of output voltages of 0 and -Vc.

In FIG. 40, three-phase AC power is supplied to each of cell inverters 4U1 to 4W3 from nine sets of insulated secondary windings of transformer 3, and output terminals of three cell inverters of each phase are connected in series. The three-phase multiplex converter 4 is realized by a configuration in which the phase multiplex converters 4U, 4V, and 4W are connected in a Y-connection. The AC motor 5 of the load is supplied with power from the three-phase multiplex converter 4. Each phase output voltage of the single-phase multiplex converters 4U, 4V, 4W is a voltage obtained by adding the output voltages of the cell inverters described above, and is + 3Vc, + 2Vc, + Vc, 0,-.
It is possible to obtain seven types of outputs Vc, -2Vc and -3Vc.

As described above, the output of the power converter is 7
It has the advantage that a voltage of the level can be obtained and a voltage with less harmonics can be supplied to the load. In FIG. 40, three sets of single-phase multiplex converters 4U, 4V, and 4W are Y-connected to form a three-phase power converter 4, but three sets of single-phase multiplex converters are Δ-connected to form a three-phase power converter. It is also possible to configure the power converter 4 of FIG. Further, the three-phase power converter 4 can be configured by other connection such as V-connection of two sets of single-phase multiplex converters.

Another feature is that a voltage obtained by adding the outputs of the cell inverters is obtained, and a high-voltage power converter can be constructed. Although FIG. 40 shows a configuration of three series in each phase, the number of series can be increased and an output of higher voltage can be obtained. Conventionally, in a case where a boosting transformer is used in order to match the voltage rating of a power converter with that of a load, the configuration shown in FIG. 40 eliminates the need for the boosting transformer.

Another advantage is that the power supply side harmonics can be reduced by shifting the voltage phase of the secondary winding as in the secondary winding of the transformer 3 shown in FIG. In FIG. 40, it is most effective to reduce the power supply harmonics by shifting the phases of the power supply voltages of the three series-connected cell inverters of each phase by 20 degrees. In general, it is known that the phase shift amount in the case of N multiplexing is effective to be 60 degrees / N. Although the inverter in FIG. 41 uses a transistor, it is apparent that a similar configuration can be realized by using another semiconductor switch such as a GTO.

[0013]

However, the conventional configuration has the following problems. First, there are many auxiliary circuits. The inrush current suppression circuit is a circuit necessary for suppressing the inrush current to the capacitor at the time of initial charging, but is an unnecessary circuit after the voltage of the capacitor is established.

One of the problems is to reduce the inrush current suppression circuit as much as possible, to simplify the circuit configuration, and to provide an inexpensive device. Second, power regeneration cannot be performed.

The cell inverter shown in FIG. 41 uses a diode converter 102 to obtain a DC voltage. Since a diode converter cannot supply a negative current, it cannot perform power regeneration as is well known. The power converter shown in FIG. 40 is often used for so-called energy saving applications in which a fan, a pump or the like is driven at a variable speed. Fans and pumps basically do not need to regenerate power, but can cause problems when shutting down the device.

When the vehicle is decelerated for stopping, it is necessary to generate a deceleration torque (negative torque) on the electric motor in order to speed up the deceleration. However, the state where the electric motor is generating negative torque is the generator operating state, and electric power is regenerated to the power supply side. Since the electric power regeneration cannot be performed in the configurations of FIGS. 40 and 41, a negative torque cannot be generated in the electric motor, and when the motor stops, the motor is decelerated by the load torque.

The load torque of the fan or pump is 2
Since it is proportional to the power, the load torque is large and rapidly decelerates at a high rotation speed. However, at a low rotation speed, the load torque becomes small and the time until stopping is increased. The prolonged stop time may be a problem in operating the apparatus.

Third, a large-capacity DC circuit filter capacitor is required. The third problem is that the DC power of the cell inverter greatly fluctuates, which is a problem common to single-phase power converters. Since the output of the inverter is single-phase, the instantaneous power of the output, which is the product of the single-phase voltage and current, fluctuates at a relatively low frequency twice the output fundamental frequency.

It is necessary to process this power fluctuation in a DC circuit, and in order to smooth the fluctuating power and keep the DC circuit voltage almost constant, it is necessary to use a large-capacity filter capacitor.

Further, since the diode converter which cannot regenerate power is used as described above, if the load power factor is low, it is necessary to increase the capacitance of the capacitor for suppressing the fluctuation of the DC voltage.

FIG. 42 shows the relationship between the load power factor and the single-phase instantaneous power waveform. The voltage V and the current i are sine waves. Here, (a) has a power factor of 1 and (b) has a power factor of 0.1.
8, (c) is when 0. As shown in FIG. 42, as the power factor decreases, the period during which the polarity of the voltage V and the polarity of the current i are different increases, and the period during which the instantaneous power P becomes negative increases. Since the DC circuit voltage is substantially constant, the DC circuit current is negative during the period when the instantaneous power is negative. However, since the diode converter cannot supply a negative current, all of the negative current is absorbed by charging the filter capacitor.

Although the capacitor voltage changes due to this charging, the only way to reduce the voltage fluctuation is to increase the capacitor capacitance. It is a disadvantage of the configurations of FIGS. 40 and 41 that it is necessary to increase the capacitance of the capacitor as described above, which also increases the outer shape of the device.

Fourth, the secondary winding of the power transformer is complicated and expensive. It has been described that shifting the phase of the secondary winding of the power transformer 3 in the configuration of FIG. 40 is effective in reducing power harmonics. The greater the number of multiplexes and the smaller the shift angle, the greater the reduction effect. However, on the other hand, as the number of multiplexes increases, the winding of the secondary winding becomes complicated, and the transformer becomes expensive. The challenge is to reduce power supply harmonics by using an inexpensive transformer with a simple winding method.

Therefore, the present invention eliminates the need for providing an inrush current suppression circuit at the time of initial charging, which is unnecessary after the establishment of the capacitor voltage, in all the cell inverters, and reduces the number of circuit components to provide an inexpensive power converter. One purpose is to provide.

Another object of the present invention is to provide a power converter capable of regenerating power without making the device expensive. Also, to provide a power converter capable of increasing the operating power factor of each cell inverter in order to suppress the fluctuation of the capacitor voltage which rises due to charging with a negative current when the load power factor is low. Is the third purpose.

It is still another object of the present invention to provide a power converter capable of reducing power supply harmonics and improving power supply power factor by using an inexpensive transformer having a simple winding method.

[0027]

In order to achieve the above object, in a power converter according to a first aspect of the present invention, at least one of a plurality of cell inverters constituting a power converter is initially charged. A rush current suppressing circuit for suppressing a rush current to the capacitor at the time. Then, at the start of operation, first, the AC power supply of the cell inverter having the rush current suppression circuit is turned on, and the capacitor of the cell inverter having the rush current suppression circuit is charged via the rush current suppression circuit. When the charging of the capacitor is completed and the DC voltage is established, the capacitor of the cell inverter having no inrush current suppression circuit is charged using the cell inverter having the inrush current suppression circuit as a current source. When the charging of the capacitors of all the cell inverters is completed, the AC power supply of the cell inverter without the inrush current suppression circuit is also turned on, and the operation is started.

In the power converter according to the second aspect of the present invention, at least one initial charging power source is provided. And
At the start of operation, the capacitor of the cell inverter is charged by the initial charging power supply. When charging of all the cell inverters is completed, the AC power supply of the cell inverter is turned on and the operation is started.

In the power converter according to a third aspect of the present invention, an inrush current suppressing circuit for suppressing inrush current to a capacitor at the time of initial charging of at least one of the plurality of cell inverters constituting the power converter. Is provided. Then, at the start of operation, first, the AC power supply of the cell inverter having the rush current suppression circuit is turned on, and the capacitor of the cell inverter having the rush current suppression circuit is charged via the rush current suppression circuit. When the charging of the capacitor is completed and the DC voltage is established, the capacitor of the cell inverter having no inrush current suppression circuit is charged using the cell inverter having the inrush current suppression circuit as a current source. When the charging of the capacitors of all the cell inverters is completed, the AC power supply of the cell inverter without the inrush current suppression circuit is also turned on, and the operation is started.

In the power converter according to claim 4 of the present invention, at least one initial charging power source is provided. And
At the start of operation, the capacitor of the cell inverter is charged by the initial charging power supply. When charging of all the cell inverters is completed, the AC power supply of the cell inverter is turned on and the operation is started.

In the power converter according to a fifth aspect of the present invention, at least one invertor of each of the plurality of cell inverters constituting the power converter has a rush current for suppressing a rush current to the capacitor at the time of initial charging. A suppression circuit is provided. Then, at the start of operation, first, the AC power supply of the cell inverter having the rush current suppression circuit is turned on, and the capacitor of the cell inverter having the rush current suppression circuit is charged via the rush current suppression circuit. When the charging of the capacitor is completed and the DC voltage is established, the capacitor of the cell inverter having no inrush current suppression circuit is charged using the cell inverter having the inrush current suppression circuit as a current source. When the charging of the capacitors of all the cell inverters is completed, the AC power supply of the cell inverter without the inrush current suppression circuit is also turned on, and the operation is started. This makes it possible to perform the initial charging in a shorter period of time and in a better balance than in the power converter according to the third aspect.

In the power converter according to claim 6 of the present invention, at least one initial charging power source for each phase is provided. Then, at the start of operation, the capacitor of the cell inverter is charged by the initial charging power supply. When charging of all the cell inverters is completed, the AC power supply of the cell inverter is turned on and the operation is started. This makes it possible to perform the initial charging in a shorter period of time and in a better balance than in the power converter according to the fourth aspect.

In a power converter according to a seventh aspect of the present invention, a multi-phase inverter having a rush current suppressing circuit is provided by integrating the cell inverters of each phase on the star connection connection point side into one. Then, at the start of operation, first, the AC power supply of the polyphase inverter having the inrush current suppression circuit is turned on, and the capacitor of the multiphase inverter having the inrush current suppression circuit is charged via the inrush current suppression circuit. When the charging of the capacitor is completed and the DC voltage is established, the capacitor of the cell inverter having no inrush current suppression circuit is charged using the multi-phase inverter having the inrush current suppression circuit as a current source. When the charging of the capacitors of all the cell inverters is completed, the AC power supply of the cell inverter without the inrush current suppression circuit is also turned on, and the operation is started. Thus, as compared with the power converter according to claim 3, the output voltage capability is reduced by using the polyphase inverter, but the capacitance of the capacitor can be reduced, and the winding structure of the transformer can be simplified. it can.

In the power converter according to claim 8 of the present invention, the multi-phase connection method of the single-phase multiplex converter is star connection. In the power converter according to the ninth aspect of the present invention, the neutral point of the polyphase load is connected to the neutral point where the single-phase multiplex converter is star-connected. Thus, the current of each phase can be controlled independently.

In the power converter according to a tenth aspect of the present invention, a star-connected impedance element is connected to the output side of each single-phase multiplex converter, and a neutral point of the star-connected impedance element is connected to the neutral point. The phase multiplexing converter is connected to a star-connected neutral point. This allows
The current of each phase can be controlled independently.

[0036] In the power converter according to claim 11 of the present invention, the multi-phase connection method of the single-phase multiplex converter is a loop connection. Thus, the current of each phase can be controlled independently.

In the power converter according to the twelfth aspect of the present invention, the multi-phase connection method of the single-phase multiplex converter is V-connection. Thus, the current of each phase can be controlled independently.

In the power converter according to the thirteenth aspect of the present invention, the rush current suppressing circuit is provided in the cell inverter having the highest DC circuit voltage. As a result, a high charging voltage can be obtained, and the charging time can be reduced.

In the power converter according to claim 14 of the present invention, the magnitude of the charging current is controlled by varying the output voltage of the cell inverter having the inrush current suppressing circuit or the initial charging power supply.

In the power converter according to claim 15 of the present invention, the output voltage of the cell inverter having the rush current suppressing circuit or the initial charging power supply is set to a substantially constant voltage, and the on / off state of the semiconductor switch constituting the charged cell inverter is turned on / off. The magnitude of the charging current is controlled by controlling the ratio and controlling the equivalent capacitor voltage. Thereby, even if the output voltage of the charging power supply reaches the maximum output voltage, the charging current can follow the command value.

In the power converter according to claim 16 of the present invention, the converter of at least one of the plurality of cell inverters constituting the power converter is a converter capable of regenerating power.

In the power converter according to a seventeenth aspect of the present invention, the cell inverters of each phase on the star connection node side are combined into a multi-phase inverter having a converter capable of regenerating power.

In the power converter according to the eighteenth aspect of the present invention, the polyphase load is an AC motor, and when generating a deceleration torque in the AC motor, the cell inverter having the converter having no power regeneration capability is short-circuited, and The AC motor is controlled by a cell inverter having a regenerable converter or a polyphase inverter having a power regenerable converter. As a result, a larger brake torque can be generated as the rotation speed of the AC motor decreases, and the stop time can be reduced.

In the power converter according to the nineteenth aspect of the present invention, when the power factor of the multiphase load is a lagging power factor or a leading power factor, the output power of the multiphase load is higher than the phase of the output composite voltage fundamental wave applied to the multiphase load. Advance or delay the output voltage fundamental phase of a cell inverter having a regenerable converter or a polyphase inverter having a power regenerable converter. As a result, the voltage utilization factor of the power converter decreases, but the operating power factor of the cell inverter without power regeneration capability increases. Therefore, the negative period of the single-phase power handled by the cell inverter having no power regeneration capability is shortened, and the negative current absorbed by the capacitor is reduced, so that the capacitance of the capacitor can be reduced. In addition, the cell inverter having the power regeneration capability has a lower operating power factor, but since the power can be regenerated, the DC circuit voltage does not increase.

According to a twentieth aspect of the present invention, in the power converter of the nineteenth aspect, the output voltage of the cell inverter having the power regenerable converter or the multi-phase inverter having the power regenerable converter is provided. The fundamental wave phase is controlled so as to be orthogonal to the load current. Thus, the reactive power of the polyphase load is supplied from the cell inverter having the regenerative ability, so that the operating power factor of the cell inverter having no regenerative ability can be increased.

In the power converter according to claim 21 of the present invention, the converter of at least one cell inverter of each phase among the plurality of cell inverters constituting the power converter is deleted. Then, as in the power converter according to the twentieth aspect, the output voltage fundamental wave phase of the cell inverter from which the converter is deleted is controlled so as to be orthogonal to the load current. When only the reactive power of the load is supplied, it is not necessary to have a converter, and accordingly, a power source is not required, and the secondary winding of the transformer can be reduced.

In a power converter according to a twenty-second aspect of the present invention, the cell inverters of each phase on the star connection node side are combined into a multi-phase inverter without a converter. Then, as in the power converter according to claim 20, control is performed such that the output voltage fundamental wave phase of the multi-phase inverter from which the converter is deleted is orthogonal to the load current. When only the reactive power of the load is supplied, it is not necessary to have a converter, and accordingly, a power source is not required, and the secondary winding of the transformer can be reduced.

According to a power converter according to claim 23 of the present invention, in the power converter according to claim 21 or 22, the output voltage fundamental wave phase of the cell inverter without a converter or the multi-phase inverter without a converter is changed. Control so as to be orthogonal to the load current. As a result, when supplying only the reactive power of the load, it is not necessary to have a converter, and accordingly, a power source is not required, and the secondary winding of the transformer can be reduced.

According to a power converter according to claim 24 of the present invention, in the power converter according to any one of claims 19, 20 and 23, if the polyphase load is an induction motor, When the load is small, the voltage of the induction motor is controlled to be lower than the rating. Since the output torque is proportional to the product of the voltage and the effective current, in order to obtain the same output torque, when the voltage is lowered, the effective current increases accordingly. Thus, when the voltage is reduced at a low load, the reactive current is reduced and the effective current is increased, so that the power factor can be increased.

In the power converter according to claim 25 of the present invention, the voltage of the induction motor is set according to the operation frequency of the induction motor. Since the load torque of a fan, a pump, or the like is proportional to the square of the rotation speed, the operating power factor of the induction motor can be increased by lowering the voltage as the rotation speed decreases.

In a power converter according to a twenty-sixth aspect of the present invention, the converter capable of regenerating power is a PWM converter capable of controlling a power supply side instantaneous current waveform. In the power converter according to claim 27 of the present invention, the PWM converter controls the power supply current of the cell inverter having the PWM converter to have a sine wave, and the same single-phase multiplex power converter has no power regeneration capability. Controls to cancel the harmonics of the power supply current of the inverter. As a result, the combined power supply current can be controlled to a sine wave, and the secondary winding of the power supply transformer does not need to be phase-shifted to reduce harmonics, so that a transformer having a simple configuration can be provided.

In the power converter according to claim 28 of the present invention, the reactive current of the power supply current of the PWM converter is controlled so that the reactive current of the combined power supply current of the single-phase multiple power converter becomes zero. Thereby, the power supply power factor can be set to 1.

[0053]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram showing an embodiment of the present invention for achieving the first object. Reference numeral 1 denotes a three-phase AC power supply, and reference numerals 2a and 2b denote power supplies such as contactors and circuit breakers for turning on and off the AC power supply. A switch 3, a transformer 4, and a three-phase multiplex power converter 4 are single-phase multiplex converters 4 each having three cell inverters connected in series.
U, 4V and 4W are connected by Y connection. Reference numeral 5 denotes a load such as an AC motor supplied from the three-phase multiplex power converter 4.

Of the cell inverters constituting the multiplex power converter 4, 4U1, 4U2, 4U3 have the same configuration as the conventional one shown in FIG. 41, and the remaining cell inverters 4V1a, 4V2
a, 4V3a, 4W1a, 4W2a, and 4W3a have the configuration shown in FIG.

In FIG. 2, 101R, 101S, 10R
1T is an AC input terminal of a three-phase power supply, 102 is a known diode converter composed of six diodes D1 to D6, 103 is a filter capacitor for smoothing DC power, and 104 is four transistors Q1 to Q4. , And 105P and 105N are AC output terminals of the inverter 104. The rush current suppressing circuit 106 at the time of initial charging is different from the cell inverter of FIG.
There are no other components, and the other configurations are the same.

That is, the configuration of FIG. 1 is different from the conventional configuration of FIG. 40 in that the power switch 2 is divided into 2a and 2b, the transformer 3 is divided into 3a and 3b, and the initial charging is performed from some cell inverters. The only difference is that the inrush current suppression circuit 106 is removed.

In the embodiment shown in FIG. 1, the functions and operations during normal operation are exactly the same as those of the conventional configuration shown in FIG. 40, except for the filter capacitor 10 of the DC circuit of the cell inverter.
Only 3 charging functions.

In FIG. 40, when the power switch 2 is turned on, all the cell inverters charge while limiting the charging current by the inrush current suppression circuit 106. On the other hand, in the configuration of the embodiment of FIG. 1, only the power switch 2a is first turned on, and the cell inverters 4U1 and 4V having the inrush current suppressing circuit 106 are turned on.
1,4W1 is initially charged as in the conventional case. Next, with the power switch 2b turned off, the remaining cell inverters 4U2a to 4W3a are charged using the cell inverters 4U1, 4V1, and 4W1 that have already been initially charged. This charging method will be described in detail below.

First, the principle of charging in the configuration in which the current source 6 is connected to the AC output terminals 105P and 105N of the cell inverter of FIG. 3 will be described. The path of the current is supplied from the current source 6 can be controlled by turning on and off the transistors Q1 to Q4 of the inverter 104 in FIG.

First, when all the transistors Q1 to Q4 are turned off, a charging mode is set. In this case, four freewheeling diodes connected in anti-parallel to the transistors Q1 to Q4 constitute a full-wave rectifier circuit, and the current supplied from the current source 6 flows through the freewheeling diode to charge the capacitor 103. Then, regardless of the polarity of the current from the current source 6, the capacitor 103 is charged with the polarity shown.
Therefore, the current supplied from the current source 6 may be DC or AC. However, the voltage Vt between the inverter output terminals 105P and 105N changes depending on the polarity of the current is. Assuming that the direction of the current is flowing from the current source 6 to the terminal 105P is positive as shown in FIG. 3, the output terminal voltage Vt is as follows.

[0061]

Is> 0: Vt = Vc is <0: Vt = −Vc That is, the terminal on which the current flows from the current source has a positive voltage.

Next, when the transistors Q1 and Q2 are turned on or the transistors Q3 and Q4 are turned on, a short-circuit mode is set.
In this case, the output terminals 105P and 10P of the inverter 104
A state equivalent to a short circuit of 5N is obtained. Therefore,
The current is of the current source 6 is bypassed by the inverter 104 and does not flow to the capacitor 103. Naturally, the voltage Vt between the output terminals of the inverter becomes 0 regardless of the magnitude of the capacitor voltage Vc.

As described above, by controlling the on / off state of the transistors Q1 to Q4 of the inverter, it is possible to switch between the capacitor charging mode and the short-circuit mode. This is a point in charging the cell inverters connected in series.

FIG. 4 is an equivalent circuit for the U phase for explaining the charging principle in more detail. FIG. 4A shows a circuit configuration considered as a single phase, where Rm and Lm are impedances of the load 5. Cell inverter 4U already charged
1 is used as a variable voltage power supply, and the remaining cell inverters 4U2a and 4U3a are charged. The output voltage of the inverter 4U1 can be varied by pulse width modulation.

FIG. 4B is an equivalent circuit when only one cell inverter is charged. One of 4U2a and 4U3a is set to the short-circuit mode to bypass the current, and the other is set to the charging mode to charge. The output terminal of the inverter to be charged has the voltage Vc of the capacitor C, and is represented by a series circuit of the capacitor C and Rm and Lm of the load 5 as shown in the figure. The current is from the inverter 4U1 to this series circuit is
To charge the capacitor C. As is apparent from the figure, the voltage Vc of the capacitor C can be charged up to the output voltage Vi of the inverter 4U1. When the charging of one inverter is completed, the mode may be switched to charge the other inverter.

Further, both the cell inverters 4U2a and 4U3a are set to the charging mode, and the charging time can be shortened by charging them simultaneously. FIG. 4C is an equivalent circuit when both inverters 4U2a and 4U3a are charged simultaneously. A circuit in which capacitors of both inverters are connected in series is basically the same circuit as when charging one inverter. Since the capacitors of both inverters are connected in series, each capacitor can be charged only up to Vi / 2. Therefore, capacitor voltages Vc2 and Vc3 are low, and inverter 4U
A plurality of cell inverters can be charged simultaneously while there is room for one output voltage Vi. However, when some cell inverters are charged to some extent, it is necessary to set one of them to the short-circuit mode.

FIG. 4 shows an example in which a single-phase multiplex converter is constituted by three cell inverters, and one of the cell inverters is used for charging another cell inverter. However, in the above description, it can be easily understood that when the number of cell inverters is arbitrary, any of the cell inverters may be used for charging. As the ratio of charging inverters increases, the number of inverters to be charged decreases, and a high voltage can be used for charging, so that charging time can be shortened. However, the effect of the present invention that the inrush current suppressing circuit is reduced to simplify the circuit and reduce the cost of the device is reduced.

4 (b) and 4 (c) show a well-known RLC series resonance circuit.
When m is small, the current becomes oscillating, and it is necessary that the current does not exceed the allowable current value of the capacitor or the inverter element. For these reasons, it is desirable to control the current during charging.

FIG. 5 shows an example of a normal control circuit configuration for controlling the charging current. 7 is a current control circuit for controlling the output current is of the inverter 4U1, and 8 is the feedback of the current is to the current control circuit 7. A current detector for performing the operation. The current control circuit 7 includes a subtractor 71, a current controller 72 such as PI control, and a PWM circuit 73.

The output current is of the inverter 4U1 is compared with the current command is * by the subtractor 71, and the amount obtained by amplifying the difference by the current controller 72 is determined by the output voltage command V of the inverter 4U1.
i *. The voltage command Vi * is pulse width modulated by the PWM circuit 73, and the output voltage Vi is controlled by turning on and off the transistor of the inverter 4U1. FIG. 6 is a block diagram of a current control loop when the combined transfer function of the PWM circuit 73 and the inverter 4U1 is set to 1. As described above, a known circuit configuration controls the voltage Vi as a result of the feedback control of the current is and causes the output current is of the inverter to follow the command is *.

When the current is controlled by the output voltage Vi of the inverter 4U1 as shown in FIG. 5, when the capacitor voltage Vc becomes high and approaches the maximum output voltage Vim of the inverter 4U1, the current is that follows the command value is * cannot flow. . As a result, the charging time becomes longer. In such a case, the current control configuration shown in FIG. 7 can ensure the current control capability.

FIG. 7 shows an example in which the cell inverter 4U3a is charged. A subtractor 74 is added to the configuration of FIG. The output of the PWM circuit 73 is used for controlling the cell inverter 4U3a to be charged, and the output Vi of the charging inverter 4U1 is fixed at the maximum output voltage Vim. As can be seen from the block diagram of FIG.
The output current is of the inverter 4U1 is determined by the difference ΔVi between the inverter output voltage Vi and the capacitor voltage Vc in the following equation.

[0073]

## EQU2 ## In the configuration of FIG. 5, when the inverter output voltage command Vi * exceeds the maximum output voltage Vim, the inverter cannot output a voltage higher than the maximum output voltage Vim, so that the difference according to the command value Vi * is obtained. The voltage ΔVi cannot be secured, and the current is also unable to follow the command value is *.

In the configuration of FIG. 7, when the inverter output voltage command Vi * exceeds the maximum output voltage Vim, the equivalent capacitor voltage V is equivalent to the excess voltage (Vi * -Vim).
By reducing c *, the difference voltage Δ according to the command value Vi *
It is configured to secure Vi. That is, the equivalent capacitor voltage Vc * is controlled so that the following equation is satisfied.

[0075]

## EQU3 ## From this equation, the equivalent capacitor voltage Vc * for securing the difference voltage .DELTA.Vi in accordance with the command value Vi * is obtained from the following equation: .DELTA.Vi = Vi * -Vc-Vim-Vc *

[0076]

Vc * = Vim + Vc-Vi * Since the capacitor voltage Vc is charged to a voltage close to Vim, it can be approximated as Vim = Vc as follows.

[0077]

Vc * = 2Vim-Vi * The above equation is calculated by the subtractor 74 in FIG. 7 to obtain a necessary equivalent capacitor voltage Vc *, and the cell inverter 4U3 during charging is obtained.
Pulse width modulation is performed on the transistor a. That is, FIG.
As described above, the equivalent capacitor voltage Vc * is controlled by turning on / off the transistor of the cell inverter 4U3a and switching between the charging mode and the short-circuit mode. In this way, by performing pulse width modulation on the cell inverter to be charged, it becomes possible to cause the current is to follow the command value is * even after the charging inverter 4U1 reaches the maximum output voltage Vim.

The basic principle of the present invention has been described with reference to the single-phase equivalent circuit shown in FIG. However, actually, it has a three-phase configuration as shown in FIG. Next, how to charge in the case of the three-phase configuration will be described using the equivalent circuit of FIG. 8 as in the case of FIG.

FIG. 8A shows a three-phase circuit configuration.
(B) is a general equivalent circuit thereof. In FIG. 8B, the capacitor capacitances Cu, Cv and Cw of each phase are determined by the number of cell inverters in the charging mode.

As described above, since the capacitor voltage is always positive on the side where the current flows, it is not always the polarity shown in the figure. With this configuration, there are various methods for charging the capacitor, but a representative method will be described.

Cell Inverter 4U as Power Supply for Charging
Although 1,4V1 and 4W1 are provided for each phase, since the sum of the three-phase currents is 0, only two sets of current control circuits for controlling the charging current are required. The cell inverter as the charging power supply for the phase that does not perform current control is voltage controlled,
The charging speed can be reduced.

FIG. 9 shows an example of the charging current control, in which the feedback control is performed on the U-phase and V-phase currents isu and isv. Both phases are controlled using the same command value isc * / 2, and Vu *, Viv * are output as outputs of the controller.
Get. The W phase is voltage control, and FIG. 9 also shows an example of how to determine each phase voltage command. U-phase current controller output Viu * is set to 1
The output voltages Viu and Viw of the U-phase inverter 4U1 and the W-phase inverter 4W1 are controlled so that the voltage becomes 1/2. However, as shown in the figure, the U-phase and the W-phase have voltages of opposite polarities. The V phase is the output voltage Viv of the inverter 4V1.
Controls the voltage to be a voltage obtained by subtracting Viu * / 2 from the V-phase current controller output Viv *. By doing so, it is possible to use up to a voltage obtained by connecting two inverters in series as a charging voltage.

As described above, the U-phase and V-phase currents are
When the current is controlled to / 2, the W-phase current isw becomes -isc. Therefore, the magnitude of isc may be determined so as to be equal to or less than the allowable current of the device. Since the current of the W phase is twice as large as that of the U and V phases, the charging of the capacitor is also performed at twice the speed. Therefore, the U-phase and the V-phase simultaneously charge two cell inverters, and the W-phase charges only one cell inverter. As a result, as shown in the figure, the U-phase and V-phase capacitor capacities are equal to the capacitor capacity C of one cell inverter.
/ 2. Therefore, the three-phase capacitor voltage Vc
The voltage increase rates of u, Vcv, Vcv become equal. When the voltage of each capacitor approaches the maximum output voltage of the charging inverters 4U1 to 4W1, a current that follows the command value cannot be supplied, and the charging time becomes longer. If the current cannot follow the command value, the cell inverter can be charged more efficiently by setting the cell inverter to the short-circuit mode and reducing the number of inverters to be charged.

In the embodiment of FIG. 1 described above, the cell inverter with an inrush current suppression circuit used as a charging power source is arranged at the neutral point side of the Y-connected converter 4. The cell inverter with the rush current suppressing circuit may be located at any position of the cell inverters connected in series, and the effect does not change even if the position is different for each phase. Further, as is apparent from the single-phase operation diagrams of FIGS. 4 and 5 for describing the basic principle of the present invention, the present invention is also applied to a single-phase power converter in which cell inverters are connected in series. It is possible. That is, the present invention is not limited to the three-phase power converter.

The charging control of the cell inverter shown in FIG. 9 is merely an example, and it is apparent that there are various ways of charging the inverters. As described above, according to the present invention as in the embodiment of FIG. 1, only some of the cell inverters can be provided with an inrush current suppression circuit, and the remaining cell inverters can be made without the inrush current suppression circuit. As a result, the number of circuit components is reduced, and an inexpensive power converter can be provided.

Next, another embodiment will be described.
As is clear from the description of the embodiment in FIG. 1, the present invention can be realized if there is a charging power supply connected in series with the cell inverter. In the embodiment of FIG. 1, the cell inverter constituting the single-phase multiplex converter is used as the power supply for charging, but a configuration using a power supply dedicated for charging is also possible.

FIG. 10 shows an embodiment in which a power supply dedicated to charging is used. Reference numerals 4UB, 4VB, and 4WB denote charging power supplies, which are connected in series with the cell inverters of each phase. Each of the cell inverters 4U1a to 4W3a has a configuration without an inrush current suppression circuit. The charging power supply only needs to have a function capable of changing the output voltage and shorting the output after the initial charging to bypass the load current. For example, a cell inverter having the same configuration as the cell inverter with an inrush current suppression circuit shown in FIG. 41 can be used as a power supply.

Further, it is possible to use a power supply having the configuration shown in FIG. 11 in which the configuration on the output side is simplified as compared with FIG. 41.
In FIG. 11, when the transistor Q1 is turned on and Q2 is turned off, the voltage Vt between the output terminals 105P and 105N becomes the capacitor voltage Vc. When Q1 is turned off and Q2 is turned on, the output voltage Vt becomes zero. Therefore, the output voltage can be controlled by the pulse width modulation of Q1 and Q2. When Q1 is turned off and Q2 is turned on, the output is short-circuited. After the initial charging, the load current can be bypassed by setting this state.

In FIG. 11, the converter side is a three-phase input, but may be a single-phase input. Further, a thyristor converter may be used instead of the diode converter 102. By using a thyristor converter, the inrush current suppression circuit 106 can be omitted, and the output voltage Vt can be controlled by changing the DC circuit voltage Vc by controlling the firing phase angle of the thyristor converter.

FIG. 12 shows an example of a configuration in which a power supply integrating the three charging power supplies 4UB, 4VB, and 4WB in FIG. 10 is realized by a well-known three-phase inverter. FIG.
11 is different from FIG. 11 only in that output inverter 104 has a three-phase inverter configuration. It can be considered that the negative output terminal 105N in FIG. 11 is shared as the neutral point N of the Y connection, and the output voltage of each phase can be controlled by pulse width modulation as in the case of FIG. it can. In addition, in all three phases, the positive side transistors Q1, Q
2. Turn on Q3, or use negative side transistor Q for all three phases.
4, Q5, or Q6 is turned on to form a three-phase short-circuit mode, and the load current can be bypassed. However, since the charging power supply in FIG. 12 is Y-connected by the output inverter 104 as described above, the installation place is limited.

As is apparent from the above description, the case where a cell inverter with an inrush current suppression circuit is used as a charging power source and the case where a dedicated power source is used are equivalent. In the following other embodiments of the present invention, a case will be described in which a cell inverter with an inrush current suppression circuit is used, but the present invention can also be realized by using a dedicated power supply.

It is apparent that the three-phase inverter shown in FIG. 12 can be operated as an inverter not only at the time of initial charging but also in a normal operation state. FIG. 13 shows FIG.
This is an embodiment of the present invention in which two three-phase inverters are used as cell inverters with an inrush current suppression circuit, and 4UVW1 constituting the power converter 4 is a three-phase inverter having the configuration shown in FIG. That is, as compared with the configuration of the embodiment of FIG. 1, in FIG. 13, the cell inverters 4U1, 4V1, and 4W1 with the rush current suppressing circuit are replaced with the three-phase inverter 4UVW1.
The difference is that the number of converters is reduced and the secondary winding of the transformer 3a is reduced as a result.

As described above, the inverter of FIG. 12 can be regarded as a Y-connection of a single-phase inverter that can output only two types of output voltages, Vc and 0. The power converter 4 of FIG. 13 using the inverter has a different output voltage capability from that of a cell inverter capable of outputting all three types of voltages Vc, 0, -Vc as shown in FIG. Can be realized. Since the three-phase inverter does not fluctuate the power of the DC circuit, the DC capacitor can be reduced.
A simple winding structure can be obtained as in the transformer 3a of FIG.

Up to this point, the single-phase multiplex converters 4U to 4W
Although the embodiment of the present invention has been described with the main circuit configuration of FIG. 1 or FIG. 10 in which Y is connected, the present invention can also be applied to other three-phase multiplexed converters.

FIG. 14 shows that the single-phase multiplex converters 4U to 4W correspond to Δ
This is an example of a configuration in the case where the wires are connected.
It is a three-phase multiplex converter. The only difference from FIG. 1 is the method of connecting the single-phase multiplex converters 4U to 4W.

FIG. 15 shows an example of the charging current control in the configuration of FIG. FIG. 15 shows each of the single-phase multiplex converters 4U to 4U.
This is an example in which current control is performed for each W, and feedback control is performed on the inverter output currents isu, isv, and isw of each phase. As shown in the figure, an equivalent circuit in which the impedances Rm and Lm of the load 5 are connected to the connection points of the [Delta] connection, the load 5 becomes a bypass path for each of the single-phase multiplex converters 4U to 4W, and the Y connection in FIG. It is not necessary for the three-phase current to be zero as in the case of. Therefore, each phase current isu, i
sv and isw can be independently controlled to any size.

FIG. 16 shows two single-phase multiplex converters 4U and 4U.
This is an example of a so-called V connection system in which the three-phase multiplex converter 4b is configured with V. One of the output terminals of both single-phase multiplex converters 4U and 4V is connected, and the other output terminal is connected to two phases of load 5. The remaining one phase of the load is connected to the connection point N of the output terminals of the two single-phase multiplex converters 4U and 4V. If the fundamental phases of the output voltages of the two single-phase multiplex converters 4U and 4V are controlled so as to be different from each other by 60 degrees, the load has a 120-degree phase difference of 3 degrees.
A phase voltage is applied. As in FIG. 1 or FIG. 10, if each single-phase multiplex conversion device 4U, 4V is constituted by three cell inverters, the number of cell inverters is reduced by the absence of the W-phase single-phase multiplex conversion device. The number of lines can be reduced. Transformers 3c and 3d have a reduced number of secondary windings.

FIG. 17 shows an example of the charging current control in the configuration of FIG. As in the case of the Δ connection in FIG. 15, the currents isu and isv of the two single-phase multiplex converters 4U and 4V can be controlled independently to arbitrary magnitudes.

As described above, according to the present embodiment, it is possible to reduce the inrush current suppressing circuit at the time of initial charging, simplify the circuit, and reduce the cost of the device. However, it is necessary to separate the power supply voltage application timing of the cell inverter having the inrush current control circuit from the cell inverter having no inrush current control circuit.
In the embodiment of the present invention described above, this power supply separation is realized by providing two sets of power switches 2a and 2b and transformers 3a and 3b or 3c and 3d.

FIG. 18 shows an example of a configuration in which power switches 2c to 2k are provided on the secondary side of the transformer for each cell inverter. It is not rare to provide a power switch for each cell inverter from the viewpoint of maintenance, and in this case, it is not necessary to separate transformers. In the configuration of FIG. 18, the power-on timing of each cell inverter can be controlled, so that the power-on timing of the cell inverter having the inrush current control circuit required by the present invention can be separated from the cell inverter having no inrush current control circuit. Clearly what you can do.

The embodiment of the present invention has been described as an example in which a cell inverter having an inrush current control circuit used as a charging power supply is provided for each phase. Even if the inverter used for charging is not provided for each phase, the remaining cell inverters can be charged.

FIG. 19 shows an example of a configuration in which only one cell inverter 4U1 of the U-phase multiplex converter 4U has an inrush current suppression circuit, and the other cell inverters do not have an inrush current suppression circuit.

FIG. 20 shows an equivalent circuit at the time of charging in this configuration. FIG. 20 (a) is a general equivalent circuit, FIG. 20 (b) is for charging a U-phase cell inverter, and FIG.
This is an equivalent circuit for simultaneously charging the phase and the W phase, and varies the output voltage Viu of the inverter 4U1 as a result of feedback control of the U-phase current isu.

FIG. 20B is an equivalent circuit when all the V-phase and W-phase cell inverters are in the short-circuit mode.
A cell inverter without a U-phase inrush current suppression circuit is charged. Finally, by charging only one cell inverter in the U phase, it is possible to charge up to the maximum output voltage of the inverter 4U1.

FIG. 20C is an equivalent circuit when all the U-phase cell inverters are in the short-circuit mode. If the voltage drop of the resistor Rm is ignored, the equivalent capacitor voltage Vc of both phases is obtained.
The current shunts so that v and Vcw are equal. If the number of cell inverters to be charged in the V-phase and the W-phase is equal, the equivalent capacitor capacity of both phases is equal, and the charging current of each cell inverter is also equal. Of course, while the capacitor voltage of each cell inverter is low, it is possible to increase the number of series and charge many inverters at the same time.

As described above with reference to FIGS. 19 and 20, if at least one cell inverter with an inrush current suppression circuit is used for charging, the remaining cell inverters can be charged. Similarly, it is clear that even if an inverter with a rush current suppressing circuit used for charging is provided only in two of the three phases, the remaining inverters can be charged.

As described above, if at least one of the three-phase power converters connected in the Y-connection is provided with an inverter having a rush current suppressing circuit, the remaining cell inverters can be charged.

This means that the present invention is effective even if the single-phase converter for each phase connected in the Y-connection is not a serial connection of a plurality of cell inverters but is a single-stage single-phase converter. Means In other words, only one of the three phases needs to be provided with the rush current suppression circuit, so that it can be reduced to １ ／ of the case where the rush current suppression circuit is provided for all three phases. The single-phase inverter at the cell inverter output stage is not limited to the configuration shown in FIG. Neutral point clamp type 3
The present invention can also be applied to a level inverter or a single-phase inverter constituted by a multi-level inverter.

As shown in FIGS. 14 and 16, a three-phase multiplex converter in which a single-phase multiplex converter is Δ-connected or V-connected can control the charging current independently for each phase. On the other hand, in the three-phase multiplex conversion device configured by the Y connection in FIG. 1, an inverter with a rush current suppression circuit for charging is provided for each phase due to the restriction that the current sum of the three phases is zero. However, the charging current of each phase cannot be controlled independently. However, even in the case of the Y connection, the current of each phase can be controlled independently if the main circuit has a neutral line. 21 and 23 show an embodiment thereof. The configuration on the power supply side is the same as that in FIG. In both embodiments, the configuration of the three-phase multiplex conversion device 4 is the same as that in FIG. 1 or FIG.

FIG. 21 shows a case where the load 5 has a neutral point terminal NL, and the neutral point terminal NL is connected to the neutral point NC of the three-phase multiplex converter 4. FIG. 22 is an equivalent circuit of the embodiment shown in FIG. 21. Since the neutral point is connected, the current sum of the three phases does not need to be 0, and the current of each phase can be controlled independently. . In the case of FIG. 1 in which the neutral point connection is not made, the current of each phase is controlled to be equal to or less than the allowable current of the apparatus due to the restriction that the current sum of the three phases is 0,
Two of the three phases can only flow about 1/2 of the allowable current. However, in FIG. 21, it is possible to flow an allowable current for all three phases, and the charging time can be reduced. If it is desired to disconnect the neutral point during normal operation, a switch may be provided between the neutral points NL and NC, and the switch may be disconnected after the initial charging is completed.

FIG. 23 shows an embodiment in which the load 5 has no neutral point terminal NL. 9U, 9V and 9W are impedance elements, and 10 is a three-pole switch. Impedance elements 9U, 9V, 9W are Y-connected and connected to the output terminal of converter 4 via switch 10. The configuration is such that the neutral points of the impedance elements 9U, 9V, and 9W, which are Y-connected, and the neutral point NC of the conversion device 4 are connected.

FIG. 24 shows the impedance elements 9U, 9V,
FIG. 23 is an equivalent circuit of FIG. 23 when 9W is an RL series circuit and the switch 10 is turned on. In this case as well, the current sum of the three phases does not need to be 0, and it is clear that the current can be independently controlled for the three phases, and the charging time can be reduced as in the case of FIG. Switch 1 after initial charging
0 is turned off, and the impedance elements 9U, 9V, 9W are disconnected from the output of the converter 4.

In the above description, it is assumed that the DC circuit voltages of the cell inverters connected in series are equal. However, the DC circuit voltages do not have to be equal. Conversely, by setting different DC voltages, harmonics of the output voltage of the power converter can be reduced. As described above, when the DC voltage of the cell inverter is different, if the cell inverter having the highest DC voltage is provided with the rush current suppressing circuit and used as a charging power supply, a high charging voltage can be obtained, and the charging time can be reduced.

The cell inverter shown in FIG.
The description has been made assuming that it is configured by a diode converter as in FIG. However, the three-phase inverter 1 shown in FIG.
The present invention can also be applied to a device using a PWM converter having the same configuration as that of the device 04. Since a diode is connected in anti-parallel to each transistor in the PWM converter as in the case of the three-phase inverter 104, it is necessary to take measures to suppress an inrush current during initial charging. Also in this case, according to the present invention, the inrush current suppressing circuit can be omitted.

In the above description, the present invention has been described focusing on the three-phase power converter. The present invention can be similarly applied to a multi-phase power converter. A typical connection constituting a multi-phase power converter is a star connection in which one side of two output terminals of a plurality of single-phase power converters is a common connection point and the remaining terminals are connected to a load, and both output terminals. With both hands, there is a loop connection connection in which the hands are held together with the single-phase converters on both sides so as to form a large ring, and the connection point is connected to a load. In the case of three phases, the star connection is Y
Since the connection and the loop connection can be regarded as Δ connection, similarly, the power converter of the Y connection connection described in the embodiment of the present invention can be applied to a more polyphase star connection, and similarly, The Δ-connection power converter can also be applied to a multi-phase loop connection.

FIG. 25 is a block diagram showing an embodiment of a power converter capable of regenerating power without making the device expensive, which is the second object of the present invention.
Cell inverters 4U1b, 4V capable of regenerating power from one of three cell inverters connected in series of 4V, 4W
1B and 4W1b are different from the conventional FIG. 40 only. Cell inverter 4 capable of regenerating power
FIGS. 26 and 27 show configuration examples of U1b, 4V1b, and 4W1b.

FIG. 26 shows a known configuration example using a thyristor converter 102a.
2a is a three-phase bridge converter 10 composed of six thyristors T1p to T6p and T1n to T6n, respectively.
2p and 102n are connected in anti-parallel. Positive group converter 102p operates when the DC current is positive, and negative group converter 102n operates when the DC current is negative. When the output power of the inverter 104 is negative, the DC circuit current also becomes negative. At that time, by operating the negative group converter 102n, the power is regenerated to the AC power supply.

FIG. 27 shows six transistors Q7 to Q13.
This is an example of a configuration using a PWM converter 102b that uses the same. The PWM converter 102b has the same known configuration as a normal three-phase transistor inverter. The current on the AC power supply side can be controlled at a high speed, and the direction of power flow can be controlled by controlling the current phase with respect to the AC voltage, so that the power can be regenerated.

The converter input in FIGS. 26 and 27 is 3
Although shown in phases, both converters may be single-phase inputs. As in the embodiment shown in FIG. 25, by allowing some of the cell inverters to regenerate electric power, it becomes possible to generate a brake torque in the AC motor 5 as a load.

At this time, the cell inverter having no regenerative capability bypasses the load current in the short-circuit mode, and only the cell inverters 4U1b, 4V1b, 4W1b having regenerative capability operate as inverters. By setting the short-circuit mode, the output voltage of the cell inverter having no regenerative ability becomes zero, so that the output voltage range of the power converter 4 is reduced, and the power that can be handled is also reduced. Nevertheless, the braking force of the electric motor 5 that could not be obtained with the configuration of FIG. 40 can be obtained, and in particular, a larger braking force can be generated in a region where the rotation speed of the electric motor is lower. Generally, the output P of the motor is expressed as follows by the rotation speed ωr of the motor and the generated torque T.

[0121]

P = ωr · T When the brake torque is generated, T is negative, and the output P is also negative. If the loss of the motor is ignored, the electric power P of the above formula is regenerated to the power supply side during braking. If the regenerative power that can be processed by the power converter 4 is Pc, the maximum brake torque Tbm that can be generated by the electric motor 5 is as follows.

[0122]

Tbm = Pc / ωr That is, even if the regenerative power is small, a larger brake torque can be generated as the rotational speed decreases.

As described above, by making some of the cell inverters capable of regenerating electric power, a brake torque can be generated in the electric motor, and the stop time can be shortened. In particular, when a fan or a pump having a square load characteristic is driven, a large brake torque can be generated in a low rotation speed region where a load torque acting as a deceleration force is small, so that the effect is great.

The embodiment shown in FIG.
Although the three-phase power converter 4 is configured by Y-connection of U, 4V, and 4W, as described in the embodiment for achieving the first object, more multi-phase star connection or Δ It is clear that the same can be applied to loop connection such as connection or V connection. In addition, the single-phase power converter 4U,
The number of cell inverters constituting 4 V and 4 W in series is not limited to three, and the number of cell inverters having regenerative ability may be any number. The power switch may be on the secondary side of the transformer 2.

When the single-phase power converters 4U, 4V, 4W are star-connected to form a multi-phase power converter 4,
As in the embodiment of FIG. 13, the cell inverter on the star connection node side may be a polyphase inverter. FIG. 28 shows a three-phase power converter 4 using the three-phase inverter 4UVW.
And a three-phase inverter 4UVW
Are configured to be able to regenerate power.
That is, the three-phase inverter 4UVW has a configuration in which the diode converter 102 in FIG. 12 is replaced with the regenerable converter 102a or 102b shown in FIG. 26 or FIG.

In the embodiment shown in FIG. 28 as well, electric power can be regenerated, so that a braking torque can be generated in the electric motor 5 and the stop time can be shortened. In addition, since the three-phase inverter has no power fluctuation in the DC circuit, the DC capacitor can be reduced, and the secondary winding can be reduced as in the transformer 3 of FIG.

Next, a description will be given of an embodiment for realizing a third object of the present invention, which is to increase the operating power factor of the cell inverter. The present invention for this purpose is, for example, shown in FIG.
As shown in FIG. 28, the power conversion apparatus 4 includes a cell inverter having a regenerative ability.

In order to achieve the object, in this embodiment, the output voltage fundamental wave of the cell inverter having the regenerative ability and the output voltage fundamental wave of the cell inverter having no regenerative ability are controlled to have different phases. Even if the cell inverter having the regenerative ability has a low operating power factor, the DC circuit voltage does not increase because the power is regenerated. The present invention increases the operating power factor of a cell inverter having no regenerative ability by operating a cell inverter having a regenerative ability at a power factor even lower than the load power factor. Hereinafter, the configuration of FIG. 25 will be described in more detail as an example.

FIG. 29 shows a fundamental wave vector V0 of the combined voltage applied to the load 5 and a fundamental wave vector of the load current i. In FIG. 29, φ is the power factor angle, and FIG. 29A shows the case where the output voltages V1, V2, and V3 of the respective cell inverters are controlled to have the same phase, and FIG. Wave vector relationship.

If the cell inverters are controlled to have the same phase as shown in FIG. 29A, the output voltages V1, V2, and V3 of each cell inverter need only be V0 / 3 to apply V0 to the load 5. Since the cell inverters are connected in series,
The load current flows through each cell inverter, and the inverter power factor becomes equal to the load power factor. In addition, the power sharing of each cell inverter becomes equal.

On the other hand, when the phases are controlled to be different as shown in FIG. 29B, the voltages V1, V2 and V3 of each cell inverter for obtaining V0 need to be higher than in the case of FIG. , And the voltage utilization rate as the power converter decreases. However, if V1 is output from a cell inverter having regenerative ability, the output of the cell inverter without regenerative ability will be V2 and V3, and the operating power factor will be high.

FIG. 29 shows the case where the load 5 has a delayed power factor. When the load is at the advanced power factor, if the output V1 of the cell inverter having regenerative ability is delayed from the combined voltage V0, the cell inverter having no regenerative ability can be obtained. It is clear that the driving power factor of the vehicle increases.

As described above, by operating the cell inverter having the regenerative ability at a power factor lower than the load power factor, the operating power factor of the cell inverter having no regenerative ability can be improved.

As a result, the negative period of the single-phase power handled by the cell inverter having no regenerative ability is shortened, and the negative current absorbed by the capacitor is reduced, so that the capacitance of the capacitor can be reduced.

In the embodiment shown in FIG. 29, the cell inverter output voltage V1 having regenerative ability and the cell inverter output voltages V2, V3 having no regenerative ability can be controlled in various phase relationships.

One phase relationship is to control the cell inverter output voltage V1 having a regenerative ability so as to be orthogonal to the load current i. That is, this is a control method in which the reactive power of the load 5 is supplied from a cell inverter having a regenerative ability as much as possible, thereby increasing the operating power factor of the cell inverter having no regenerative ability.

As described above, if only the reactive power is supplied, it is possible to eliminate the converter of the cell inverter. FIG. 30 shows an embodiment of the present invention in which the converter section of the cell inverter is eliminated. The difference from the embodiment of FIG. 25 is that the cell inverters 4U1b, 4V1b,
4W1b is a cell inverter 4U1 without a converter
c, 4V1c, and 4W1c. Since there is no converter in the cell inverters 4U1c, 4V1c, and 4W1c, a power source is not required, and the secondary winding of the transformer is reduced.

FIG. 31 shows the configuration of the cell inverters 4U1c, 4V1c, 4W1c without a converter. FIG.
As is clear from comparing FIG. 1 with FIG. 26 or FIG.
The cell inverters 4U1c, 4V1c, 4W1c have a very simple configuration. Basically, the cell inverter 4 shown in FIG.
U1c, 4V1c and 4W1c handle only reactive power,
Controls the active power to maintain the voltage of the DC capacitor.

Similarly to FIG. 30, in the case where the star connection point in FIG. 28 is constituted by a three-phase inverter, the converter can be omitted by processing only the reactive power by the three-phase inverter. FIG. 32 shows a configuration example thereof. 4UVWa is a three-phase inverter without a converter, and FIG. 33 shows a configuration example thereof. The three-phase inverter 4UVWa is the single-phase inverter 4U1c, 4 in FIG.
As in the case of V1c and 4W1c, basically, only the reactive power is handled, and the active power is controlled to maintain the DC capacitor voltage.

As described above, the configuration of the power conversion device and the control device thereof will be described as an embodiment for realizing the third object of the present invention, which is to increase the operating power factor of the cell inverter having no regenerative ability. did. Further, an effective control device that achieves the object when the induction motor is driven as the load 5 will be described below.

FIG. 34 is a general vector diagram of an induction motor, where V is a terminal voltage and i is a current. FIG. 34 shows two types of current i when the output torque of the induction motor is large and small. As shown in the figure, the current vector i changes according to the output torque, and moves on a locus indicated by a chain line.
The current i is divided into a component ir (active current) in phase with the voltage V and a component ii (reactive current) orthogonal to the voltage V. The reactive current ii is an exciting current and is substantially constant regardless of the magnitude of the output torque, and the effective current ir is substantially proportional to the output torque. Therefore, as the output torque decreases, the effective current ir decreases, and as the output torque decreases, the power factor angle φ increases as shown in the figure, resulting in a low power factor.

However, FIG. 34 is a vector diagram when the motor is operated at the rated voltage, and it is not always necessary to operate the motor at the rated voltage. The lower the voltage, the smaller the exciting current and the smaller the reactive current ii. At a specific rotation speed, the output torque is substantially proportional to the product (active power) of the voltage V and the active current ir. Therefore, in order to obtain the same output torque, the effective current ir increases by an amount corresponding to the reduction in the voltage. That is, if the voltage is reduced at a low load, the reactive current ii decreases and the effective current ir increases, so that the power factor increases.

FIG. 35 is a comparison of vectors when the output torque is small. FIG.
(B) is a vector diagram at the time of operation at 1/2 of the rated voltage. In (b) where the voltage is reduced to １ ／ compared to (a), the reactive current is １ ／ and the effective current is doubled, and the power factor is greatly improved.

When this control for lowering the voltage at the time of light load is combined with controlling the output voltage of the cell inverter to a different phase as described above, it is possible to minimize the reduction in the voltage utilization rate of the power converter. Thus, the present invention can be realized more effectively.

When the magnitude of the load torque is known in advance, such as with a fan or a pump, control for lowering the voltage can be easily realized. FIG. 36 is a control configuration diagram showing an embodiment of the present invention. Reference numeral 11 denotes a control device for controlling the power conversion device 4 and a controller 10 for controlling the on / off of the transistor.
6 and a function circuit 107 for determining a voltage command V * for the frequency command F * of the output voltage of the power converter 4.

The controller 106 controls the on / off of the transistor so that the output voltage of the power converter 4 follows the given frequency command F * and voltage command V *. If the function set in the function circuit 107 is a straight line, the frequency command F * is proportional to the voltage command V *, and the exciting current becomes substantially constant even when the frequency changes, and the rated voltage operation is performed.
On the other hand, in the function circuit 107 of FIG. 36, the smaller the frequency command F *, the smaller the voltage command V
* That is, a function is set such that the V / F ratio decreases as the frequency becomes lower.

As described above, the load torque of the fan or the pump is proportional to the square of the rotation speed, and the lower the rotation speed, the smaller the load torque. Therefore, the lower the frequency, the more V
According to the embodiment of FIG. 36 in which the / F ratio is controlled to be small, it is possible to reduce the voltage at a light load and increase the operating power factor of the induction motor 5.

Although the embodiment for achieving the third object of the present invention has been described above, the number of cell inverters and the connection system of the power converter are not limited to specific ones. Obviously, more polyphase star, loop, or V connections are also effective.

A fourth object of the present invention is to provide a control device for a power conversion device capable of reducing power supply harmonics and improving power supply power factor by using an inexpensive transformer having a simple winding method. An embodiment of the present invention for achieving the above will be described.

In order to achieve this object, the aforementioned FIG.
5 or a power converter provided with a cell inverter having a regenerative ability shown in FIG. In addition, the cell inverter having the regenerative ability has a P inverter as shown in FIG.
It has a WM converter and is realized by a device capable of controlling the power supply current at high speed.

An embodiment in which this object is realized by the configuration of FIG. 37 will be described. FIG. 37 shows one phase for each phase as in FIG.
This is a configuration including two cell inverters with a PWM converter. However, the secondary windings of the transformer 3 are all Y-connected.

When each phase has a cell inverter with a PWM converter as shown in FIG. 37, a U-phase PWM converter is provided so that the sum of the power supply currents of U-phase cell inverters 4U1b, 4U2, and 4U3 becomes a sine wave. What is necessary is just to control the converter current of the cell inverter 4U1b. Similarly, the cell inverters 4V1b and 4W1b with V-phase and W-phase PWM converters also control the converter currents so that the sum of the power supply currents of the respective phases becomes a sine wave. That is, the PWM converters of the cell inverters 4U1b, 4V1b, and 4W1b are operated as a known active filter.

FIG. 38 shows an example of a control configuration of a U-phase PWM converter showing an embodiment of the control device of the present invention. The main circuit also shows only the U-phase portion. The V phase and the W phase have the same configuration. In FIG. 38, 8R1, 8S1, 8T1, 8R
2,8T2,8S2 are cell inverters 4U1b, 4U2
12U is a converter current control circuit for the cell inverter 4U1b.
In the current control circuit 12U, 110 is a subtractor, 111
Is a control amplifier such as a PI controller, 112 is a voltage detection circuit that detects the phase voltages eR, eS, eT of the power supply by performing insulation and filtering from the power supply voltage of the cell inverter 4U1b, and 113 is a DC current command ii *. , Ir * using the phase voltage signals eR, eS, eT to obtain an AC current command iR0 *.
, IS0 *, iT0 *, a coordinate rotator 114
R, 114S, 114T, 115R, 115S, 115
T is a subtractor, 116R, 116S and 116T are control amplifiers such as PI controllers, 117R, 117S and 117T are PWM circuits for pulse width modulation of the converter transistor of the cell inverter 4U1b, and 118 is a basic circuit based on three-phase input current. The fundamental wave removing filters 119R, 119S, and 119T that remove only waves and output only harmonic components are coefficient units that double the input signal.

The basic role of the converter of the cell inverter 4U1b is to control the DC circuit voltage Vc to a constant level.
A deviation from c * is obtained, and a signal amplified by the control amplifier 111 is set as an effective current command ir *. Reactive current command ii *
May be a constant value. The power supply voltage eR,
Using eS and eT, the AC current commands iR0 * and iS such that the component in phase with the power supply voltage is proportional to the active current command ir * and the component orthogonal to the power supply voltage is proportional to the reactive current command ii *.
0 *, iT0 *. The feedback of the converter phase currents iR1, IS1, and iT1 of the cell inverter 4U1b is performed, and the subtractors 115R, 115S, and 115T determine deviations so as to follow the current commands iR0 *, iS0 *, and iT0 *. 116S, 116
The deviation is amplified by T, and the PWM circuits 117R, 117S, 1
If the converter is controlled by a signal modulated by 17T, the power supply current of the cell inverter 4U1b is controlled to a sine wave.

However, this alone requires the cell inverter 4U.
1b can be controlled to a sine wave, but the harmonic current of the power supply current of the other cell inverters 4U2 and 4U3 is
The combined power supply current does not become a sine wave. In order to make the combined power supply current a sine wave, the AC current commands iR0 *, iS0
*, IT0 * are subtracted by subtractors 114R, 114S, 114T.
And the output iRH of the coefficient units 119R, 119S, and 119T,
Current commands iR1 *, iS1 * corrected by iSH, iTH
, IT1 *. Coefficient units 119R, 119S, 11
9T outputs iRH, iSH, iTH are output from current detector 8R
This is a signal obtained by extracting a harmonic component from the power supply current iR2, iS2, iT2 of the cell inverter 4U2 detected by 2, 8S2, 8T2 by the fundamental wave removing filter 118 and doubling the harmonic component. That is, the output of the fundamental wave removing filter 118 is a harmonic component of the power supply current of the cell inverter 4U2.

If the power sharing between cell inverters 4U2 and 4U3 is controlled equally, the power supply currents of both inverters have the same waveform. Therefore, the fundamental wave removing filter 118 which is a harmonic component of the power supply current of the cell inverter 4U2.
Multipliers 119R, 119S, 119T that double the output of
Output iRH, iSH, iTH are both cell inverters 4U
This is a composite value of harmonic currents of 2 and 4U3. The current command iR1 *, iS1 *, iT1 * obtained by subtracting the combined harmonic currents iRH, iSH, iTH is added to the cell inverter 4U.
By controlling the power supply currents iR1, iS1 and iT1 of the cell inverter 1b to follow, the power supply current of the cell inverter 4U1b cancels the power supply harmonic currents of the cell inverters 4U2 and 4U3, and the combined current of the power supply 1 becomes a sine wave. can do.

As described above, by providing the cell inverters 4U1b, 4V1b, and 4W1b with the PWM converter with the active filter function, the power supply harmonic current of the cell inverter with the diode converter can be canceled, and the combined power supply current can be reduced. It can be controlled to a sine wave. At this time, as shown in FIG. 37, the secondary winding of the power transformer 3 does not need to be phase-shifted to reduce harmonics, can be a transformer having a simple configuration, and provide an inexpensive device. It becomes possible.

In the embodiment shown in FIG. 38, the currents of cell inverters 4U1b and 4U2 are detected for three phases, respectively. However, since the sum of the currents of three phases is zero, only two phases are detected and the remaining one is detected. The phase current can also be calculated.
Also, the current of the cell inverter 4U2 is detected only in two phases,
The currents of the remaining two phases may be obtained by shifting the phase by 120 degrees each. Further, since the current waveform of the diode converter is determined by the power impedance, the magnitude of the DC current, and the like, a pattern previously formed as a function of the inverter output power can be used.

In the embodiment shown in FIG. 38, the current control is performed using the AC amount. However, in recent years, the detected AC amount is converted into a DC amount by coordinate conversion to perform feedback control, and the voltage of the control result is obtained. It is needless to say that the command may be converted again into an AC amount by coordinate conversion. In this case, it is necessary to perform the same coordinate conversion of the harmonic current superimposed on the current command as the detected current in order to cancel.

FIG. 38 shows an embodiment in which the secondary voltage of the transformer is equal and the DC circuit voltage of each cell inverter is also the same. This embodiment can be easily applied even when the power supply voltage of each cell inverter is different. FIG.
The secondary winding ratio (voltage ratio) of the transformer corresponding to the eight U-phase cell inverters 4U1b, 4U2, 4U3 is N1, N
2, N3, the power supply current of each cell inverter is 1 /
It is reflected on the primary winding at a ratio of N1, 1 / N2, 1 / N3. Therefore, for example, if the R-phase current command iR1 * is obtained from the harmonic current iRH2 of the cell inverter 4U2 and the harmonic current iRH3 of the cell inverter 4U3 as in the following equation, the harmonic current can be canceled.

[0161]

## EQU8 ## iR1 * = iR0 * -N1 (iRH2 / N2
+ IRH3 / N3) The same applies to the S phase and the T phase. 37 and 38
In the embodiment, all the secondary windings of the transformer 3 are Y-connected. However, the effect of the present embodiment does not change even if a secondary winding of another connection such as a Δ connection is included. For example, Δ
In the case of a connected secondary winding, the winding current cannot be detected. However, assuming that the circulating current is not flowing and the current sum of the three windings is 0, each winding current should be calculated from the phase current. Can be. The harmonic current can be canceled by calculating the harmonic current from the calculated winding current and superimposing it on the current command in consideration of the above-mentioned winding ratio.

In the embodiment shown in FIGS. 37 and 38, the harmonic current is canceled for each inverter output phase. However, when the three-phase inverter shown in FIG. It cannot be done every time. In this case, the power supply current of all the cell inverters other than the three-phase inverter may be detected or obtained by calculation, and the PWM converter may be controlled to cancel the harmonic current.

In the embodiment shown in FIG. 38, the reactive current command ii * is set to a constant value. However, this ii * is made negative to cancel the delay power factor of the diode converter and improve the power supply power factor. . Diode converters do not have a very low power factor, but have some delay power factor due to the overlap angle during commutation. ii * is set to a negative value, the fundamental wave current of the PWM converter is set as a leading power factor, and the delay of the diode converter is canceled. Although the power factor improving effect can be obtained even if ii * is set to a negative constant value, the power factor can be more positively controlled.

FIG. 39 shows an embodiment in which a power factor control circuit 13U is added to the embodiment of FIG. 38, and the other parts are exactly the same as those of FIG. In the power factor control circuit 13U, 120R, 120S and 120T are coefficient multipliers for doubling the input signal, 121R, 121S and 121T are adders, 1
22 is a coordinate rotator, 123 is a subtractor, and 124 is a control amplifier.

The power supply current of the cell inverter 4U2 is doubled by the coefficient units 120R, 120S, and 120T.
R, 121S and 121T add the power supply current of the cell inverter 4U1b for each phase. This added current iR
T, iST, and iTT are combined currents corresponding to the primary current of the transformer 3. This combined current iRT, iST, iTT
Is converted by the coordinate converter 122 using the phase voltage signals eR, eS, and eT of the power supply voltage detected by the voltage detection circuit 112 of the converter control circuit 12U to derive a reactive current component iiT orthogonal to the voltage of the combined current. I do. A signal obtained by amplifying the deviation from the command iiT * obtained by the subtractor 123 by the control amplifier 124 so that the reactive current component iiT is set to 0 is used as a reactive current command ii * of the power supply current of the cell inverter 4U1b as a PWM converter controller. Give to 12U.

As a result, the reactive current ii of the power supply current of the cell inverter 4U1b is controlled so that the reactive current iiT of the combined current becomes 0, and the power supply power factor always becomes 1. FIG.
In the above, the phase voltage signals eR, eS, and eT of the power supply voltage used in the coordinate converter 122 are obtained from the secondary voltage of the transformer 3, but may be obtained from the primary voltage of the transformer. If the primary voltage of the transformer is used, it is possible to prevent the influence of the impedance drop of the transformer. Further, the primary current of the transformer 3 may be used as the combined current.

In FIG. 39, the reactive current iiT of the combined current is set to 0.
Thus, the power supply power factor is set to 1; however, the ratio between the reactive current iiT and the effective current irT of the combined current can be controlled to be constant. In this way, it is possible to control to an arbitrary power factor.

[0168]

As described above in detail, according to the present invention, it is not necessary to provide an inrush current suppressing circuit in the initial charging, which is unnecessary after the voltage of the capacitor is established, in all the cell inverters. It is possible to provide an inexpensive power conversion device with a small amount.

Further, it is possible to provide a power converter capable of regenerating power without making the device expensive. Further, in order to suppress the fluctuation of the capacitor voltage which rises by being charged with the negative current when the load power factor is low, a control device of the power conversion device capable of increasing the operating power factor of each cell inverter is provided. Can be provided.

Further, it is possible to provide a control device of a power converter capable of reducing power supply harmonics and improving power supply power factor by using an inexpensive transformer having a simple winding method.

[Brief description of the drawings]

FIG. 1 is a configuration diagram of an embodiment for achieving a first object of the present invention.

FIG. 2 is a main circuit configuration diagram showing a detailed configuration example of the cell inverter of FIG. 1;

FIG. 3 is a main circuit diagram of a cell inverter for explaining a charging principle of the present invention.

FIG. 4 is an equivalent circuit diagram for explaining the charging principle of the present invention.

FIG. 5 is a configuration diagram of a control circuit of a charging current according to the present invention.

FIG. 6 is a block diagram of a charging current control loop of the present invention.

FIG. 7 is a diagram showing another control circuit configuration of the charging current of the present invention.

FIG. 8 is an equivalent circuit diagram of three phases for explaining the charging principle of the present invention.

FIG. 9 is a configuration diagram of a three-phase charging current control of the present invention.

FIG. 10 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 11 is a main circuit configuration diagram showing a configuration example of a charging power supply in FIG. 10;

FIG. 12 is a main circuit configuration diagram showing another configuration example of the charging power supply in FIG. 10;

FIG. 13 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 14 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 15 is a configuration diagram of a control circuit for a charging current according to the embodiment of FIG. 14;

FIG. 16 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 17 is a configuration diagram of a control circuit for a charging current according to the embodiment of FIG. 16;

FIG. 18 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 19 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 20 is an equivalent circuit diagram for explaining a charging current control principle of the embodiment of FIG. 19;

FIG. 21 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 22 is an equivalent circuit diagram for explaining a charging current control principle of the embodiment of FIG. 21;

FIG. 23 is a configuration diagram of another embodiment for achieving the first object of the present invention.

FIG. 24 is an equivalent circuit diagram for explaining a charging current control principle of the embodiment of FIG. 23;

FIG. 25 is a configuration diagram of an embodiment for achieving a second object of the present invention.

26 is a configuration diagram of the cell inverter with a converter capable of regenerating power in FIG. 25.

FIG. 27 is another configuration diagram of the cell inverter with a converter capable of power regeneration in FIG. 25;

FIG. 28 is a configuration diagram of another embodiment for achieving the second object of the present invention.

FIG. 29 is a vector diagram for explaining the principle for achieving the third object of the present invention.

FIG. 30 is a configuration diagram of another embodiment for achieving the third object of the present invention.

FIG. 31 is a configuration diagram of the cell inverter without a converter of FIG. 30;

FIG. 32 is a configuration diagram of another embodiment for achieving the third object of the present invention.

FIG. 33 is a configuration diagram of a three-phase cell inverter without a converter of FIG. 32;

FIG. 34 is a vector diagram for explaining the principle of another embodiment which achieves the third object of the present invention.

FIG. 35 is a vector diagram showing an effect of another embodiment that achieves the third object of the present invention.

FIG. 36 is a control configuration diagram of another embodiment that achieves the third object of the present invention.

FIG. 37 is a configuration diagram of an embodiment for achieving a fourth object of the present invention.

FIG. 38 is a control configuration diagram according to an embodiment for achieving the fourth object of the present invention.

FIG. 39 is a control configuration diagram of another embodiment for achieving the fourth object of the present invention.

FIG. 40 is a main circuit configuration diagram of a conventional power converter.

FIG. 41 is a main circuit configuration diagram showing a configuration example of a conventional cell inverter.

FIG. 42 is an instantaneous waveform diagram of single-phase power for explaining a conventional problem.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 ... AC power supply 2, 2a-2k ... Power switch 3, 3a-3d ... Power transformer 4, 4a, 4b ... Polyphase power converter 5 ... Load 6 ... Power supply 7 7a: current control circuit 8: current detector 9U to 9W: impedance element 10: switch 11: power converter control circuit 12U: converter current control circuit 13U: Power factor control circuit

Claims (28)

[Claims] 1. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A plurality of cell inverters, at least one of the plurality of cell inverters is provided with an inrush current suppression circuit that suppresses an inrush current to the capacitor during initial charging. When the AC power of the cell inverter having the inrush current suppression circuit is turned on and the initial charging of the capacitor is completed, the remaining capacitor of the cell inverter having no inrush current suppression circuit is charged by using the cell inverter as a current source, and all of the capacitors are charged. When charging of the capacitor of the cell inverter is completed, an inrush current suppression circuit is not provided. A power inverter for turning on the AC power of the cell inverter. 2. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. In a power conversion device configured by connecting a plurality of power supply units, at least one initial charging power supply is provided, and at the start of operation, the capacitors of the cell inverters are charged by the initial charging power supply, and the charging of the capacitors of all the cell inverters is completed. Then, the AC power supply of the cell inverter is turned on. 3. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplexing converter by connecting a plurality of single-phase output sides in series,
In a power converter for supplying power to a multi-phase load by connecting the single-phase multiplex converter in a multi-phase connection, an inrush current to the capacitor during initial charging of at least one of the plurality of cell inverters is suppressed. When an inrush current suppression circuit is provided, the AC power supply of the cell inverter having the inrush current suppression circuit is turned on at the start of operation, and when the initial charging of the capacitor is completed, the remaining inrush current suppression circuit is also set up using the cell inverter as a current source. A power converter comprising: charging a capacitor of a cell inverter having no inverter; and completing charging of capacitors of all the cell inverters, turning on the AC power of the cell inverter having no rush current suppressing circuit. 4. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplexing converter by connecting a plurality of single-phase output sides in series,
In the power converter for supplying power to a polyphase load by connecting the single-phase multiplex converter in a multi-phase connection, at least one initial charging power supply is provided, and at the start of operation, a capacitor of the cell inverter is charged by the initial charging power supply. When the charging of the capacitors of all the cell inverters is completed, the AC power supply of the cell inverter is turned on. 5. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplexing converter by connecting a plurality of single-phase output sides in series,
In the power converter for supplying power to a polyphase load by connecting the single-phase multiplex converter to a multi-phase connection, at least one cell inverter of each of the plurality of cell inverters is supplied with an inrush current to the capacitor at the time of initial charging. When the inrush current suppression circuit is provided, the AC power of the cell inverter having the inrush current suppression circuit is turned on at the start of operation, and the initial charging of the capacitor is completed, the remaining inrush current suppression circuit is set using the cell inverter as a current source. A power converter comprising: charging a capacitor of a cell inverter having no invertor; and completing charging of capacitors of all cell inverters, turning on the AC power supply of the cell inverter having no inrush current suppressing circuit. 6. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting DC power of the capacitor to AC power. A single-phase multiplexing converter by connecting a plurality of single-phase output sides in series,
In the power converter for supplying power to a polyphase load by connecting the single-phase multiplex converter in a multi-phase connection, at least one initial charging power supply for each phase is provided, and at the start of operation, the capacitor of the cell inverter is charged by the initial charging power supply. A power conversion device, wherein when the battery is charged and the capacitors of all the cell inverters are completely charged, the AC power of the cell inverter is turned on. 7. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplexing converter by connecting a plurality of single-phase output sides in series,
In the power converter that supplies power to a polyphase load by star-connecting the single-phase multiplex converter, a polyphase inverter having an inrush current suppression circuit is provided instead of the cell inverter of each phase on the star connection connection point side, and the operation is performed. At the start, when the AC power of the polyphase inverter is turned on and the initial charging of the capacitors of the polyphase inverter is completed, the capacitors of the cell inverters are charged using the polyphase inverter as a current source, and the capacitors of all the cell inverters are charged. When the power conversion is completed, the AC power supply of the cell inverter is turned on. 8. The power converter according to claim 3, wherein the single-phase multiplex converter has a multi-phase connection method in a star connection. 9. The power converter according to claim 7, wherein a neutral point of the polyphase load is connected to a neutral point in which the single-phase multiplex converter is star-connected. Power converter. 10. The power converter according to claim 7, wherein a star-connected impedance element is connected to an output side of each of said single-phase multiplex converters, and said star-connected impedance element is connected to a star-connected impedance element. A power converter wherein a neutral point is connected to a neutral point in which the single-phase multiplex converter is star-connected. 11. The power converter according to claim 3, wherein a multi-phase connection method of the single-phase multiplex converter is a loop connection. 12. The power converter according to claim 3, wherein the multi-phase connection method of the single-phase multiplex converter is a V-connection. 13. The power conversion device according to claim 3, wherein the cell inverter provided with the inrush current suppression circuit is a cell inverter having the highest DC circuit voltage of the cell inverter. 14. The power converter according to claim 1, wherein the magnitude of the charging current is controlled by varying an output voltage of a cell inverter having an inrush current suppression circuit or an initial charging power supply. A power conversion device characterized in that: 15. The cell inverter according to claim 1, wherein an output voltage of a cell inverter having an inrush current suppression circuit or an initial charging power supply is substantially constant, and the cell inverter is charged. A power converter characterized in that the magnitude of a charging current is controlled by controlling an on / off ratio of a semiconductor switch constituting the above. 16. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting DC power of the capacitor to AC power. A single-phase multiplex converter is configured by connecting a plurality of single-phase output sides in series to form a single-phase multiplex converter, and the single-phase multiplex converter is connected in a multi-phase manner to supply power to a multi-phase load. The power converter wherein at least one cell inverter of each phase is a converter capable of regenerating power. 17. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplex converter is configured by connecting a plurality of single-phase output sides in series to form a single-phase multiplex converter, and the single-phase multiplex converter is star-connected to supply power to a multi-phase load. A power conversion device comprising a polyphase inverter having a power regenerable converter instead of a cell inverter for each phase. 18. The power converter according to claim 16, wherein the polyphase load is an AC motor, and a cell having no power regeneration capability when generating a deceleration torque in the AC motor. A power converter, wherein the inverter is short-circuited, and the AC motor is controlled by a cell inverter having the power regenerable converter or a polyphase inverter having the power regenerable converter. 19. The power converter according to claim 16, wherein when a power factor of the multi-phase load is a delayed power factor, a phase of an output combined voltage fundamental wave applied to the multi-phase load is reduced. Also advances the output voltage fundamental wave phase of the cell inverter having the power regenerable converter or the multi-phase inverter having the power regenerable converter, and when the power factor of the The control is performed such that the output voltage fundamental phase of the cell inverter having the power regenerable converter or the multi-phase inverter having the power regenerable converter is delayed more than the output composite voltage fundamental wave phase applied to the load. Characteristic power converter. 20. The power converter according to claim 19, wherein the output voltage fundamental wave phase of the cell inverter having the power regenerable converter or the multi-phase inverter having the power regenerable converter is orthogonal to the load current. A control device for a power conversion device, characterized in that the control is performed as follows. 21. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplex converter is configured by connecting a plurality of single-phase output sides in series to form a single-phase multiplex converter. The power converter wherein at least one cell inverter in each phase does not have the converter. 22. A cell inverter having a converter for rectifying AC power from an AC power supply and converting it to DC power, a capacitor for smoothing the DC power, and a single-phase inverter for converting the DC power of the capacitor to AC power. A single-phase multiplex converter is configured by connecting a plurality of single-phase output sides in series to form a single-phase multiplex converter, and the single-phase multiplex converter is star-connected to supply power to a multi-phase load. A power converter comprising a multi-phase inverter without a converter instead of a cell inverter for each phase. 23. The power converter according to claim 21 or 22, wherein the output voltage fundamental wave phase of the cell inverter without the converter or the multi-phase inverter without the converter is controlled to be orthogonal to the load current. Power converter characterized by the following: 24. The power converter according to claim 19, wherein the multi-phase load is an induction motor, and when the load of the induction motor is small, the load of the induction motor is reduced. A power converter characterized by controlling a voltage lower than a rating. 25. The power converter according to claim 24, wherein the voltage of the induction motor is set according to an operation frequency of the induction motor. 26. The power converter according to claim 16, wherein the converter capable of regenerating power is constituted by a PWM converter capable of controlling an instantaneous current waveform on a power supply side. apparatus. 27. The power converter according to claim 26, wherein the PWM converter controls a power supply current of a cell inverter having the PWM converter to have a sine wave and a power regeneration of the same single-phase multiplex power converter. A power converter characterized in that control is performed so as to cancel harmonics of a power supply current of a cell inverter having no capability. 28. The power converter according to claim 27, wherein the reactive current of the power supply current of the PWM converter is controlled so that the reactive current of the combined power supply current of the single-phase multiple power converter becomes zero. Power converter.