JPH1118468A - Controller for reluctance type synchronous motor - Google Patents
Controller for reluctance type synchronous motorInfo
- Publication number
- JPH1118468A JPH1118468A JP9161429A JP16142997A JPH1118468A JP H1118468 A JPH1118468 A JP H1118468A JP 9161429 A JP9161429 A JP 9161429A JP 16142997 A JP16142997 A JP 16142997A JP H1118468 A JPH1118468 A JP H1118468A
- Authority
- JP
- Japan
- Prior art keywords
- current
- command value
- motor
- rotor
- current command
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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Links
Landscapes
- Control Of Electric Motors In General (AREA)
Abstract
Description
【0001】[0001]
【発明の属する技術分野】本発明は、工作機械や産業機
械、電気自動車等に利用される同期電動機の制御装置に
関するものであり、特にリラクタンス型同期電動機の制
御特性改良に関するものである。BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for a synchronous motor used in machine tools, industrial machines, electric vehicles and the like, and more particularly to an improvement in control characteristics of a reluctance type synchronous motor.
【0002】[0002]
【従来の技術】従来例を図11に示す。3相(U,V,
W相)での制御を例にとる。上位制御器より速度指令値
SVCが指令され減算器1により回転子速度SPDの差
DIFが演算される。DIFが入力されたPI制御器2
により電機子電流指令値STCが電流指令演算部5に出
力される。また、回転子速度SPDを参照して界磁電流
演算部101内の界磁電流パターンにより界磁電流指令
値SFCが電流指令演算部5に出力される。電流指令演
算部5は回転子位置SP(機械角)を参照し、電気角に
変換後、電機子電流指令値STCと界磁電流指令値SF
Cをベクトル演算し、電流指令値SIC(SIUC、S
IVC)得る。(なお電機子電流指令値STCと界磁電
流指令値SFCの位相差は電気角でπ/2[rad]であ
る。)その後、U相電流指令値SIUCとそれとの位相
差(2π)/3または(4π)/3[rad]であるV相電
流指令値SIVCを相分配する。3相中2相が決定され
れば、残りのW相の電流(SIWC)は決定されるので
特に図示しない。各相の電流指令値は増幅器8を介して
3相電流として電動機9に印加され電動機9の回転子を
回転させる。回転子にとりつけられた回転子位置検出手
段である検出器10により回転子位置SPが得られ、電
流指令演算部5および微分器12にフィードバックされ
る。微分器12は回転子位置SPを微分して回転子速度
SPDとして減算器1および界磁電流演算部101に出
力される。以上の動作からわかるように、電流指令値S
ICは界磁電流指令値SFCと電子電流指令値STCを
ベクトル加算して作られる(図2参照)。制御する際、
電機子電流指令値STCと電流指令値SICが成す角を
動作角α21と呼ぶことにすると、電機子電流指令値S
TCおよび界磁電流指令値SFCが変化した際、動作角
αも変化してしまう。また、界磁弱め制御を行う際、電
機子電流指令値STC>界磁電流指令値SFCとなる場
合、動作角α≒0に近づくことになる。界磁弱め領域で
ない場合、界磁電流指令値SFCは一定の値をとる。こ
の場合の電動機特性は、電機子電流:Iqをx軸にとり
出力トルク:τをy軸にとった場合、界磁電流:Id
(1倍、2倍、3倍、4倍)をパラメータとすると後に
詳述する図6(a)のようにそれぞれ線形性のある出力
トルク特性になることが実験で確認されている。しかし
界磁電流が低い場合(図6(b)特性曲線65、領域
A)や逆に大きな場合(図6(b)特性曲線65、領域
C)、図に示すように非線形な特性の部分が発生する。
図6の特性も後に詳述する。2. Description of the Related Art FIG. 11 shows a conventional example. Three phases (U, V,
The control in (W phase) will be taken as an example. The speed command value SVC is instructed by the host controller, and the subtractor 1 calculates the difference DIF between the rotor speeds SPD. PI controller 2 with DIF input
As a result, the armature current command value STC is output to the current command calculation unit 5. Further, a field current command value SFC is output to the current command calculation unit 5 based on the field current pattern in the field current calculation unit 101 with reference to the rotor speed SPD. The current command calculation unit 5 refers to the rotor position SP (mechanical angle), converts it into an electrical angle, and then converts the armature current command value STC and the field current command value SF
C is vector-operated and the current command value SIC (SIUC, SUC
IVC). (Note that the phase difference between the armature current command value STC and the field current command value SFC is π / 2 [rad] in electrical angle.) Then, the U-phase current command value SIUC and the phase difference (2π) / 3 thereof Alternatively, the V-phase current command value SIVC of (4π) / 3 [rad] is distributed in phases. If two of the three phases are determined, the current (SIWC) of the remaining W phase is determined, so that it is not shown. The current command value of each phase is applied as a three-phase current to the motor 9 via the amplifier 8 to rotate the rotor of the motor 9. The rotor position SP is obtained by a detector 10 as rotor position detecting means attached to the rotor, and is fed back to the current command calculator 5 and the differentiator 12. The differentiator 12 differentiates the rotor position SP and outputs the result to the subtractor 1 and the field current calculator 101 as the rotor speed SPD. As can be seen from the above operation, the current command value S
The IC is made by vector-adding the field current command value SFC and the electronic current command value STC (see FIG. 2). When controlling,
If the angle formed between the armature current command value STC and the current command value SIC is called an operation angle α21, the armature current command value S
When the TC and the field current command value SFC change, the operating angle α also changes. Also, when performing the field weakening control, if the armature current command value STC> the field current command value SFC, the operating angle approaches α 近 0. When the current is not in the field weakening region, the field current command value SFC takes a constant value. The motor characteristics in this case are as follows: when the armature current: Iq is set on the x-axis and the output torque: τ is set on the y-axis, the field current: Id
It has been experimentally confirmed that the output torque characteristics have linear characteristics as shown in FIG. 6A, which will be described later in detail, when (1 ×, 2 ×, 3 ×, 4 ×) is a parameter. However, when the field current is low (the characteristic curve 65 in FIG. 6B, the area A) or large (FIG. 6B, the characteristic curve 65 in the area C), the non-linear characteristic portion as shown in FIG. Occur.
The characteristics of FIG. 6 will be described later in detail.
【0003】[0003]
【発明が解決しようとする課題】従来技術の制御装置を
使用した場合、電流指令値SICは界磁電流指令値SF
Cと電機子電流指令値STCをベクトル加算して作られ
るため、電機子電流指令値STCおよび界磁電流指令値
SFCが変化した際、動作角αが不定となる。このこと
は、電流に対して得られるトルクの効率・力率が常に変
化することを意味しており、制御上の不具合(例えば、
界磁電流が低い場合にトルクを得ようとするとき、電機
子電流が増加するので合成電流(電流指令)も増加す
る。しかし大きな電流を流す割に出力されるトルクが小
さいため効率が悪い。しかも、電機子電流を流すq軸
(ベクトル制御のd−q軸理論)からd軸への漏れイン
ダクタスの影響により、界磁弱めを行っているのにも関
わらずd軸成分の磁束の増加→誘起電圧の増加→電源電
圧の飽和→印可電流の低下→制御不能といった不具合の
図式が成り立つ。)が発生する。また、上記のように電
機子電流指令値STC>界磁電流指令値SFCとなる場
合(動作角α≒0になる場合)、トルク定数が低下し制
御対象が動作をしないのでトルク指令値が増加し、なお
かつ制御的な遅れを生じてハンチング現象や最悪の場合
制御不能に陥る可能性がある。以上のことは、合成電流
が低い場合の非線形な電動機出力特性の領域にも認めら
れる。本発明は上述した事情から成されたものであり、
簡単な構成で安定して効率良い電動機の制御を実現させ
ることができるリラクタンス型同期電動機の制御装置を
提供することを目的とする。When the conventional control device is used, the current command value SIC becomes the field current command value SF.
C and the armature current command value STC are created by vector addition. Therefore, when the armature current command value STC and the field current command value SFC change, the operation angle α becomes indefinite. This means that the efficiency and the power factor of the torque obtained with respect to the current always change, and a control defect (for example,
When an attempt is made to obtain torque when the field current is low, the armature current increases, so that the combined current (current command) also increases. However, the efficiency is poor because the torque output is small for a large current. In addition, due to the influence of leakage inductance from the q-axis (dq-axis theory of vector control) through which the armature current flows to the d-axis, the flux of the d-axis component increases despite the field weakening. → Increase in induced voltage → Saturation of power supply voltage → Decrease in applied current → Inability to control, a diagram of the problem is established. ) Occurs. Also, as described above, when the armature current command value STC> the field current command value SFC (when the operation angle α ≒ 0), the torque constant decreases and the control target does not operate, so the torque command value increases. However, there is a possibility that a hunting phenomenon or in the worst case, loss of control may occur due to a control delay. The above is also observed in a non-linear motor output characteristic region when the combined current is low. The present invention has been made under the circumstances described above,
It is an object of the present invention to provide a control device for a reluctance type synchronous motor capable of stably and efficiently controlling an electric motor with a simple configuration.
【0004】[0004]
【課題を解決するための手段】本発明の上記目的は、固
定子スロット内に施された巻線に一定直流電流を流し、
軟磁性体からなる回転子表面の磁気抵抗が固定子から見
て回転方向に高低差を持つよう内部に磁気絶縁手段を備
えた回転子を回転させた場合に回転角θに対して発生す
るトルクτが正負の周期的な特性を持つようなリラクタ
ンス型同期電動の制御装置であって、前記回転子の回転
子位置を検出する手段と、該電動機に電流を流すため
に、前記回転子位置とトルク指令値より電流指令を算出
する電流指令演算部を具備する制御装置において、トル
ク指令値の極性により符号が決定され前記回転子位置に
加算される動作角α(0<α<π/2[rad];電気角)
を持ち、回転子速度を参照して変化する関数パターンA
より係数Aを演算し動作角に乗ずることで回転子速度に
より動作角が変化するような動作角演算部を持ち、電流
振幅制御部内で電流指令値に乗ずることで補正後電流指
令値とするような係数Bを、回転子速度を参照して変化
する関数パターンBより演算するような減衰値演算部を
持ち、また、制御器外部より設定の行えるモード選択器
により、電動機駆動効率もしくは出力トルク特性を優先
するように制御が行えるモードを選択できる減衰値演算
部を持ち、入力された指令値に対して出力される指令値
が線形な電動機特性を持つようにを変換し、電動機の持
つ非線形特性を線形のように扱う電動機特性/指令変換
部を持つことを具備することで達成される。本発明にあ
っては、トルク指令値の極性により符号が決定され前記
回転子位置に加算される動作角α(0<α<π/2[ra
d];電気角)を持ち、回転子速度を参照して変化する関
数パターンAより係数Aを演算し動作角に乗ずることで
回転子速度により動作角が変化するような動作角演算部
を持つことにより効率の良い制御装置が得られた。ま
た、制御器外部より設定の行えるモード選択器により、
電動機駆動効率もしくは出力トルク特性を優先するよう
に制御が行えるモードを選択でき、電流振幅制御部内で
電流指令値に乗ずることで補正後電流指令値とするよう
な係数Bを、回転子速度を参照して変化する関数パター
ンBより演算するような減衰値演算部を持つことで、制
御性の良い制御装置が得られた。また、非線形特性を持
つリラクタンス型電動機を永久磁石型同期電動機のよう
な線形性をもつ電動機のように制御を行うことが実現で
きたため、工作機械等サーボ用途に使用できるという利
点も得られた。SUMMARY OF THE INVENTION It is an object of the present invention to supply a constant direct current to a winding provided in a stator slot,
Torque generated with respect to rotation angle θ when rotating a rotor with magnetic insulation inside so that the magnetic resistance of the rotor surface made of soft magnetic material has a height difference in the direction of rotation when viewed from the stator τ is a reluctance type synchronous motor control device having a positive and negative periodic characteristics, a means for detecting the rotor position of the rotor, and the rotor position for flowing a current to the motor, In a control device including a current command calculation unit for calculating a current command from a torque command value, a sign is determined based on the polarity of the torque command value, and an operation angle α (0 <α <π / 2 [ rad]; electrical angle)
And a function pattern A that changes with reference to the rotor speed
The current angle control unit has an operation angle calculation unit that calculates the coefficient A and multiplies the operation angle to change the operation angle according to the rotor speed, and multiplies the current command value in the current amplitude control unit to obtain the corrected current command value. The motor drive efficiency or the output torque characteristic is provided by a mode selector that has a damping value calculation unit that calculates a simple coefficient B from a function pattern B that changes with reference to the rotor speed. Has a damping value calculation unit that can select a mode in which control can be performed so as to give priority to the input, and converts the command value output to the input command value so that it has linear motor characteristics, and the nonlinear characteristics of the motor This is achieved by having a motor characteristic / command conversion unit that treats as a linear. In the present invention, the operating angle α (0 <α <π / 2 [ra], whose sign is determined by the polarity of the torque command value and is added to the rotor position.
d]; electrical angle), and has an operation angle calculation unit that calculates the coefficient A from a function pattern A that changes with reference to the rotor speed and multiplies the operation angle to change the operation angle according to the rotor speed. As a result, an efficient control device was obtained. In addition, a mode selector that can be set from outside the controller allows
A mode in which control can be performed so as to give priority to motor drive efficiency or output torque characteristics can be selected, and a coefficient B that makes the corrected current command value by multiplying the current command value in the current amplitude control unit is referred to the rotor speed. Thus, a control device with good controllability was obtained by having an attenuation value calculation unit that calculates from the function pattern B that changes. In addition, since a reluctance motor having nonlinear characteristics can be controlled like a motor having linearity like a permanent magnet synchronous motor, there is an advantage that the motor can be used for servo applications such as machine tools.
【0005】[0005]
【発明の実施の形態】以下、添付図面をもとに本発明の
実施形態を説明する。なお、特にことわらない限り、同
記号、番号のものは同一の構成および機能を有するもの
である。図1は本発明の制御ブロック図例である。上位
制御器より速度指令値SVCが指令され減算器1により
回転子速度SPDとの差DIFが演算され、PI制御器
2により電機子電流指令値STCを電流指令演算部5に
出力する部分は、従来例と同様である。電機子電流指令
値STCは特性補正部4と、動作角演算部3に出力さ
れ、特性補正部4は補正電機子電流指令値STCCを演
算、電流指令演算部5を介して、電流指令値SIC(S
IUC、SIVC)とし相分配後、増幅器8により電動
機9を駆動する。電動機9の回転子位置を検出するため
の検出器10が取り付けられており、回転子位置SPを
電流指令演算部5の電流位相制御部7と微分器12に出
力する。また微分器12により回転子位置SPは回転子
速度SPDとなり、本発明の特徴である動作角演算部3
と減衰値演算部11に出力される。動作角演算部3には
電機子電流指令値STCと回転子速度SPDが入力さ
れ、補正後動作角SACを電流位相制御部7に出力す
る。電流指令演算部5内には電流振幅制御部6と、相分
配器を備えた電流位相制御部7があり、電流振幅制御部
6は減衰値演算部11からの出力である係数SK2と補
正電機子電流指令値STCCを演算し、電流指令値SI
Cとする。電流位相制御部7には、電流指令値SICが
電流振幅制御部6より入力され、また動作角演算部3よ
り補正後動作角SACが入力され、更に検出器10より
回転子位置SPが入力される。ここで電流位相制御部7
について詳しく説明すると、回転子位置SPを制御に利
用できるように機械角(回転子1回転で2πrad)に対
し、電気角(正弦波1周期で2πradになる。通常は、
回転子の1磁極対で2πradになる。)に変換後、回転
子磁極のq軸(電機子電流)に電流が流れる様に位相合
わせを行う。その後、補正後動作角SAC分だけ加算器
により位相を移行させ、U相、V相の位相操作の後(3
相の場合、2相が決定されると1相(W相)は自ずと決
定される。)、電流指令値SICと移行後の位相を乗算
器により相別に演算し、U相電流指令値SIUC、V相
電流指令値SIVCとして出力される。減衰値演算部1
1は、外部動作モード選択器からの信号MODEによ
り、駆動効率優先モードまたは定出力制御モードの選択
されたモード別に、回転子速度SPDを参照して係数S
K2を電流振幅制御部6に出力する。また動作角演算部
3も回転子速度SPDを参照し、内部的に係数Kaを演
算、動作角αに乗ずることで、補正後動作角Ka・α
(=α’)を電流位相制御部7に出力する。Embodiments of the present invention will be described below with reference to the accompanying drawings. Unless otherwise specified, components having the same symbols and numbers have the same configuration and function. FIG. 1 is an example of a control block diagram of the present invention. A portion that instructs the speed command value SVC from the host controller, calculates the difference DIF from the rotor speed SPD by the subtractor 1, and outputs the armature current command value STC to the current command calculation unit 5 by the PI controller 2 This is the same as the conventional example. The armature current command value STC is output to the characteristic correction unit 4 and the operating angle calculation unit 3. The characteristic correction unit 4 calculates the corrected armature current command value STCC, and outputs the current command value SIC via the current command calculation unit 5. (S
IUC, SIVC), and after the phase distribution, the motor 8 is driven by the amplifier 8. A detector 10 for detecting the rotor position of the electric motor 9 is attached, and outputs the rotor position SP to the current phase controller 7 and the differentiator 12 of the current command calculator 5. Further, the rotor position SP becomes the rotor speed SPD by the differentiator 12, and the operation angle calculation unit 3 which is a feature of the present invention is provided.
Is output to the attenuation value calculator 11. The armature current command value STC and the rotor speed SPD are input to the operating angle calculation unit 3, and the corrected operating angle SAC is output to the current phase control unit 7. The current command calculation unit 5 includes a current amplitude control unit 6 and a current phase control unit 7 having a phase distributor. The current amplitude control unit 6 includes a coefficient SK2 output from the attenuation value calculation unit 11 and a correction electric machine. The child current command value STCC is calculated and the current command value SI
C. The current phase controller 7 receives the current command value SIC from the current amplitude controller 6, the corrected operating angle SAC from the operating angle calculator 3, and the rotor position SP from the detector 10. You. Here, the current phase controller 7
To be more specific, the mechanical angle (2πrad for one rotation of the rotor) becomes an electrical angle (2πrad for one cycle of a sine wave) so that the rotor position SP can be used for control.
It becomes 2πrad for one magnetic pole pair of the rotor. ), The phases are adjusted so that a current flows on the q axis (armature current) of the rotor magnetic pole. Then, the phase is shifted by the adder by the corrected operating angle SAC, and after the U-phase and V-phase operations (3
In the case of a phase, when two phases are determined, one phase (W phase) is automatically determined. ), The current command value SIC and the shifted phase are calculated for each phase by a multiplier, and output as a U-phase current command value SIUC and a V-phase current command value SIVC. Attenuation value calculator 1
1 is a coefficient S by referring to the rotor speed SPD for each of the drive efficiency priority mode and the constant output control mode selected by the signal MODE from the external operation mode selector.
K2 is output to the current amplitude controller 6. The operating angle calculating unit 3 also internally refers to the rotor speed SPD, internally calculates a coefficient Ka, and multiplies the operating angle α to obtain the corrected operating angle Ka · α.
(= Α ′) is output to the current phase controller 7.
【0006】図2に電流のベクトル図を示す。電機子電
流Idと合成電流Ioの成す角を動作角αとする。図の
ベクトルを関係式に表すと[数1]および[数2]のよ
うになる。FIG. 2 shows a current vector diagram. The angle between the armature current Id and the combined current Io is defined as an operation angle α. When the vectors in the figure are represented by relational expressions, they are as shown in [Equation 1] and [Equation 2].
【数1】界磁電流成分 :Id=Io・sinα[Equation 1] Field current component: Id = Io · sinα
【数2】電機子電流成分:Iq=Io・cosαつまり、印
加電流(合成電流)Ioを従来のベクトル制御のように
界磁電流成Idと電機子電流成分Iqに分けて考えるこ
とができる。## EQU2 ## Armature current component: Iq = Io · cos α That is, the applied current (synthetic current) Io can be divided into a field current component Id and an armature current component Iq as in conventional vector control.
【0007】図3は本発明が適用される電動機の特性例
である。3相(U,V,W相)電動機において、V,W
相端子を短絡しU→V,W相に直流電流を流した場合の
回転角θに対する発生するトルクτを示した特性であ
る。特性曲線36は定格電流Io、特性曲線35は定格
電流Ioの2倍、特性曲線34は定格電流Ioの3倍、
特性曲線33は定格電流Ioの4倍、特性曲線32は定
格電流Ioの5倍を流した場合のトルク特性である。以
上のように直流電流の大きさの違いにより発生するトル
クτの大きさおよびピークトルク(P1〜P5)を示す
角度θが異なることがわかる。この図において、直流電
流をIo固定とすると角度θ=0の場合、[式1]、
[式2]よりIoに対して界磁電流成分が0%で電機子
電流成分100%であり、角度θ=π/2(電気角)の
場合、界磁電流成分が100%で電機子電流が0%であ
る。また角度θ=π/4(電気角)の場合、Ioに対し
て界磁電流成分、電機子電流成分共に約71%を流した
場合の発生トルクτだとみなすことができる。角度θ=
動作角αの場合、流した電流値がわかっているので特性
曲線より得られるトルク値もわかる上、前記[式1]お
よび[式2]を使用することで従来のベクトル制御時の
ような界磁電流成分と電機子電流成分とを分離した考え
方を適用することが可能である。FIG. 3 shows a characteristic example of a motor to which the present invention is applied. In a three-phase (U, V, W phase) motor, V, W
This is a characteristic showing the generated torque τ with respect to the rotation angle θ when a phase terminal is short-circuited and a direct current flows in the U → V, W phase. The characteristic curve 36 is the rated current Io, the characteristic curve 35 is twice the rated current Io, the characteristic curve 34 is three times the rated current Io,
The characteristic curve 33 is a torque characteristic when the rated current Io is four times the rated current, and the characteristic curve 32 is a torque characteristic when the rated current Io is five times. As described above, it can be seen that the magnitude of the torque τ generated by the difference in the magnitude of the DC current and the angle θ indicating the peak torque (P1 to P5) are different. In this figure, when the DC current is fixed to Io, when the angle θ = 0, [Equation 1]
According to [Equation 2], the field current component is 0% and the armature current component is 100% with respect to Io. When the angle θ = π / 2 (electrical angle), the armature current is 100% and the field current component is 100%. Is 0%. When the angle θ = π / 4 (electrical angle), it can be regarded as the generated torque τ when about 71% of both the field current component and the armature current component flow to Io. Angle θ =
In the case of the operating angle α, since the value of the flowing current is known, the torque value obtained from the characteristic curve can be known, and by using the above [Equation 1] and [Equation 2], the field as in the conventional vector control is obtained. It is possible to apply a concept in which the magnetic current component and the armature current component are separated.
【0008】図4に動作角演算部3内の制御ブロック図
の例を示す。また、図5に計数演算41の関数パターン
を示す。動作角演算部3は、回転子速度SPDを参照し
て係数Kaを出力する係数演算部41と初期固定値の動
作角αとを乗算器42で演算し、補正後動作角Ka・α
を出力する。その際、トルク指令値STCの極性を極性
判定部46で判定し、補正後動作角Ka・αの符号スイ
ッチ43を決定し、それぞれの条件に従い、係数44、
または係数45を乗じて補正後動作角SACとして出力
する。なお演算部41は回転子速度SPDをパラメータ
とする演算式、もしくはデータテーブルを持っており、
図5のように、基底回転数Nbaseで関数が変化し、係数
Kaを算出する。FIG. 4 shows an example of a control block diagram in the operation angle calculation unit 3. FIG. 5 shows a function pattern of the counting operation 41. The operating angle calculating unit 3 calculates a coefficient calculating unit 41 that outputs a coefficient Ka with reference to the rotor speed SPD and an operating angle α of an initial fixed value by a multiplier 42, and calculates a corrected operating angle Ka · α.
Is output. At this time, the polarity of the torque command value STC is determined by the polarity determining unit 46, the sign switch 43 of the corrected operating angle Ka · α is determined, and the coefficient 44,
Alternatively, the corrected operating angle SAC is output by multiplying by the coefficient 45. The arithmetic unit 41 has an arithmetic expression or a data table using the rotor speed SPD as a parameter.
As shown in FIG. 5, the function changes with the base rotation speed Nbase, and the coefficient Ka is calculated.
【0009】図6(a)は電機子電流Iqに対するトル
クτを示した図で、界磁電流Idを1、2、3、4倍流
した特性図であり、界磁電流Idが大きくなっても、ト
ルク特性は比例的に大きくなるわけではない。表現を変
えて、界磁電流Idと電機子電流Iqの合成電流Ioに
対する特性を図6(b)に示す。これは、合成電流Io
に対して発生しうるピークトルク(動作角α=不定))
を示している。特性曲線65からわかるように、領域A
では2次関数的に電流に対して出力トルクが増加し、領
域Bでは直線的に電流トルクが増加する。領域Cの特性
は、電動機の設計にもよるが補2次関数的に特性が変化
する。(通常の電動機の設計では領域Cは積極的に使用
しない場合が多い。) 領域Aの特性になる理由は、印加電流の界磁電流成分が
回転子と固定子間のエアギャップ部で消費されてしま
い、回転子内の界磁磁束が不足するものだと考えられ
る。領域Bの直線的な特性の部分は、エアギャップでの
エネルギー消費が飽和し充分に界磁磁束が回転子内に形
成できている場合の特性である。領域Cでは界磁磁束が
エアギャップ部および回転子内で飽和してしまうために
起こる特性である。FIG. 6A is a graph showing the torque τ with respect to the armature current Iq, and is a characteristic diagram in which the field current Id flows 1, 2, 3, and 4 times. However, the torque characteristics do not increase proportionally. In other words, FIG. 6B shows the characteristics of the field current Id and the armature current Iq with respect to the combined current Io. This is the combined current Io
Torque that can be generated with respect to (operation angle α = undefined))
Is shown. As can be seen from the characteristic curve 65, the area A
, The output torque increases quadratically with respect to the current, and the current torque increases linearly in the region B. The characteristics of the region C vary according to a quadratic function, though depending on the design of the motor. (The area C is often not actively used in a normal motor design.) The reason for the characteristic of the area A is that the field current component of the applied current is consumed in the air gap between the rotor and the stator. It is considered that the field magnetic flux in the rotor is insufficient. The linear characteristic portion in the region B is a characteristic in a case where energy consumption in the air gap is saturated and a sufficient field magnetic flux is formed in the rotor. In the region C, this is a characteristic that occurs because the field magnetic flux is saturated in the air gap portion and the rotor.
【0010】図7に特性補正部4のパターン例を示す。
入力として補正前トルク指令値STC、出力として補正
後トルク指令値STCCが得られる。補正曲線71は、
電動機のトルク特性(図6(b)参照)を補正するため
各領域A、B、Cの補関数形状をしており、補正曲線7
1を使用することで近似的に直線72のような直線的な
電動機特性が得られる。この特性補正部4は関数パター
ンを電動機特性に合わせることによりキャンセルも可能
である。FIG. 7 shows an example of a pattern of the characteristic correction section 4.
A pre-correction torque command value STC is obtained as an input, and a post-correction torque command value STCC is obtained as an output. The correction curve 71 is
In order to correct the torque characteristics of the electric motor (see FIG. 6B), each of the regions A, B, and C has a complementary function shape.
By using 1, an approximately linear motor characteristic such as a straight line 72 can be obtained. The characteristic correction unit 4 can also cancel by matching the function pattern to the motor characteristics.
【0011】図8に減衰値演算部11の例と回転子速度
SPDをパラメータとする関数パターン例を示す。減衰
値演算部11には回転子速度SPDと外部モード選択器
からの信号MODEで設定されるスイッチ83がある。
外部モード選択器により電動機の制御方法は電動機効率
優先モードと定出力優先モードとに選択できる。制御方
法は前記のものに限定するものではなく、演算部81、
82に示すような関数パターンを操作することで目的別
の制御を実現できる。(例えば、回転数が低い場合のみ
トルクを増加しようとする場合、低回転部分の係数を
1.0以上に設定することで可能。減衰値演算部とは、
関数パターンの設定次第で増幅作用も可能である。) スイッチ83に選択された回転子速度SPDをパラメー
タとする関数パターンにより、係数SK2を電流振幅制
御部6に出力する。FIG. 8 shows an example of the attenuation value calculator 11 and an example of a function pattern using the rotor speed SPD as a parameter. The attenuation value calculator 11 has a switch 83 set by the rotor speed SPD and a signal MODE from an external mode selector.
The motor control method can be selected between the motor efficiency priority mode and the constant output priority mode by the external mode selector. The control method is not limited to the one described above, and the operation unit 81,
By manipulating the function pattern as shown at 82, control for each purpose can be realized. (For example, if the torque is to be increased only when the rotational speed is low, it is possible to set the coefficient of the low rotational speed portion to 1.0 or more.
Amplification is also possible depending on the setting of the function pattern. The coefficient SK2 is output to the current amplitude control unit 6 according to a function pattern using the rotor speed SPD selected by the switch 83 as a parameter.
【0012】図9にモード選択器による図3に示すよう
な電動機特性図の動作点の違いを具体的に示す。どちら
も、界磁電流成分が同じ場合に制御モードを選択器で切
り替えた場合の動作点(動作角αおよび出力トルクτ)
の違いを示す。定出力モードの動作点をQ、効率優先モ
ードをPとする。同じ界磁電流でも動作角αが異なるた
め、電流指令値(合成電流Io)の大きさが異なってい
る。動作点Qでは、出力トルクが大きいが効率の面から
みると、特性曲線92のピークよりずれている為、電流
の絶対値に対して得られるトルク(=エネルギー変換
率)が低いこととなり、特性曲線91のピークで制御す
る動作点Pに比べて、効率が低いことになる。(動作点
Pでは特性曲線91のピークにあるため、電流に対して
得られるトルクの効率が良い。) つまり、界磁電流が同じ場合に、得られるトルクの絶対
値が大きいのは流す合成電流が大きい動作点Qの場合
で、逆に効率が良いのは動作点Pだといえる。制御モー
ドを選択するのは、動作点αを選択するのと同義であ
り、動作角演算部と減衰値演算部、および電動機特性補
正部の各関数パターンを操作することで、この動作点α
を回転子速度SPDと電流指令値SICをパラメータと
して制御することが可能である。FIG. 9 specifically shows the difference in operating point in the motor characteristic diagram as shown in FIG. 3 by the mode selector. In both cases, the operating point (operating angle α and output torque τ) when the control mode is switched by the selector when the field current components are the same.
Show the difference. The operating point in the constant output mode is Q, and the efficiency priority mode is P. Since the operation angle α is different even for the same field current, the magnitude of the current command value (combined current Io) is different. At the operating point Q, the output torque is large, but is shifted from the peak of the characteristic curve 92 from the viewpoint of efficiency, so that the torque (= energy conversion rate) obtained with respect to the absolute value of the current is low. The efficiency is lower than the operating point P controlled by the peak of the curve 91. (Because the operating point P is at the peak of the characteristic curve 91, the efficiency of the torque obtained with respect to the current is good.) In other words, when the field current is the same, the absolute value of the obtained torque is large because the combined current Can be said to be the operating point P where the efficiency is high. Selecting the control mode is synonymous with selecting the operating point α. By operating each function pattern of the operating angle calculator, the attenuation value calculator, and the motor characteristic correction unit, this operating point α is selected.
Can be controlled using the rotor speed SPD and the current command value SIC as parameters.
【0013】[0013]
【発明の効果】以上のように、本発明で示す同期電動機
の制御装置によれば、回転子速度を参照して変化する係
数を演算する動作角演算部および制御モード設定可能な
減衰値演算部を具備したため、ベクトル制御を行うのと
同等な制御(界磁弱め制御を含む)ができるようになっ
た。また、界磁電流成分は合成電流に含まれているた
め、小トルク時の界磁電流を少なくでき、省電力化と巻
線の小発熱化、および高効率な制御が実現できた。加減
速時などの大トルクが必要な場合、合成電流が大きくな
り界磁電流成分も大きくなるためトルク定数の増加がで
き、加速時間の短縮が実現できる。さらに入力トルク指
令に対し補正トルク特性を出力するような電動機特性補
正部を持たせた為、非線形なトルク特性を持つ電動機を
線形性の良い特性を持つ電動機のように制御できるよう
になった。なお、本発明は前述の第1図から第10図に
示した本発明の実施例に限定されるものではなく、その
主旨を逸脱しない範囲で下記のような変形を行なっても
よい。 (1)本実施形態では、3相電動機で説明したが2相以
上の多相電動機でも良い。 (2)本実施形態では、回転子を持つ電動機について説
明したが、可動子を持つ構造のリニア型電動機に適用し
ても良い。 (3)本実施形態では、合成電流Ioと動作角αを制御
する方法について説明したが、図10には本発明と同等
のベクトル演算(d軸、q軸電流独立で制御)に適用し
た例を示す。その場合の演算式と関数パターンの例を以
下に示す。動作点αを制御する場合に、界磁電流Id、
電機子電流Iqを独立にベクトル演算する場合の演算を
示すと[数3]のようになる。As described above, according to the control apparatus for a synchronous motor shown in the present invention, the operating angle calculating section for calculating the coefficient that changes with reference to the rotor speed and the damping value calculating section capable of setting the control mode. Therefore, control (including field weakening control) equivalent to performing vector control can be performed. Further, since the field current component is included in the combined current, the field current at the time of small torque can be reduced, and power saving, small heat generation of the windings, and highly efficient control can be realized. When a large torque is required during acceleration or deceleration, the combined current increases and the field current component also increases, so that the torque constant can be increased and the acceleration time can be reduced. Further, since a motor characteristic correction unit for outputting a correction torque characteristic in response to an input torque command is provided, it is possible to control a motor having a non-linear torque characteristic like a motor having good linearity characteristics. The present invention is not limited to the embodiment of the present invention shown in FIGS. 1 to 10 described above, and the following modifications may be made without departing from the gist of the present invention. (1) In this embodiment, a three-phase motor has been described, but a multi-phase motor having two or more phases may be used. (2) In the present embodiment, the motor having the rotor has been described, but the present invention may be applied to a linear motor having a structure having a mover. (3) In the present embodiment, the method of controlling the combined current Io and the operating angle α has been described. However, FIG. 10 shows an example in which the present invention is applied to a vector operation equivalent to that of the present invention (controlled independently of d-axis and q-axis currents). Is shown. An example of an arithmetic expression and a function pattern in that case is shown below. When controlling the operating point α, the field current Id,
The calculation when the armature current Iq is independently vector-calculated is as shown in [Equation 3].
【数3】Io=KIq・Iq+KId1・KId2・Id 図10(a)は、電動機回転数Nをパラメータとする界
磁電流Idに乗ずる係数KId1(界磁弱め係数)の関数
パターン例である。図10(b)は、電動機回転数Nを
パラメータとする電機子電流Iqに乗ずる係数KIq(ト
ルク減衰係数)の関数パターン例である。図10(c)
は、トルク指令値STCをパラメータとする界磁電流I
dに乗ずる係数KId2(トルク低減係数)の関数パター
ン例である。この係数を回転数に関係なく一定(例えば
KId2=1.0)とすることで、トルク指令値に関係な
く界磁電流一定値(ただし、回転数Nの関数である。)
で制御することも可能である。## EQU3 ## Io = KIq.Iq + KId1.Kid2.Id FIG. 10A is an example of a function pattern of a coefficient KId1 (field weakening coefficient) multiplied by a field current Id with the motor speed N as a parameter. FIG. 10B is an example of a function pattern of a coefficient KIq (torque damping coefficient) by which the armature current Iq is multiplied by the motor rotation speed N as a parameter. FIG. 10 (c)
Is the field current I with the torque command value STC as a parameter.
It is an example of a function pattern of a coefficient KId2 (torque reduction coefficient) by which d is multiplied. By making this coefficient constant irrespective of the rotation speed (for example, KId2 = 1.0), the field current constant value (however, a function of the rotation speed N) regardless of the torque command value.
It is also possible to control with.
【図1】 本発明の同期電動機の制御装置の実施形態を
示す説明図である。FIG. 1 is an explanatory diagram showing an embodiment of a control device for a synchronous motor of the present invention.
【図2】 本発明の同期電動機の制御装置に使用される
電流ベクトル図である。FIG. 2 is a current vector diagram used in the control device for a synchronous motor of the present invention.
【図3】 本発明の同期電動機の制御装置に使用される
電動機の特性例である。FIG. 3 is a characteristic example of a motor used in a control device for a synchronous motor of the present invention.
【図4】 本発明の同期電動機の制御装置に具備される
動作角演算部のブロック図の例である。FIG. 4 is an example of a block diagram of an operation angle calculation unit included in the control device for the synchronous motor of the present invention.
【図5】 本発明の同期電動機の制御装置に具備される
動作角演算部の関数パターンの例である。FIG. 5 is an example of a function pattern of an operation angle calculation unit provided in the control device for a synchronous motor of the present invention.
【図6】 本発明の同期電動機の制御装置に使用される
電動機の特性例である。FIG. 6 is a characteristic example of a motor used in a control device for a synchronous motor of the present invention.
【図7】 本発明の同期電動機の制御装置に具備される
電動機特性補正部の説明図である。FIG. 7 is an explanatory diagram of a motor characteristic correction unit provided in the synchronous motor control device of the present invention.
【図8】 本発明の同期電動機の制御装置に具備される
減衰値演算部のモード選択器の動作例である。FIG. 8 is an operation example of a mode selector of the attenuation value calculation unit provided in the synchronous motor control device of the present invention.
【図9】 本発明の同期電動機の制御装置に具備される
減衰値演算部のモード選択器による動作点の説明図であ
る。FIG. 9 is an explanatory diagram of an operation point by a mode selector of an attenuation value calculation unit provided in the control device for a synchronous motor of the present invention.
【図10】 本発明の同期電動機の制御装置の変更実施
適用例である。FIG. 10 is an application example of a modified embodiment of the synchronous motor control device of the present invention.
【図11】 従来の同期電動機の制御装置のブロック図
である。FIG. 11 is a block diagram of a conventional control device for a synchronous motor.
1 減算器、2 PI制御器、3 動作角演算部、4
特性補正部、5 電流指令演算部、6 電流振幅制御
部、7 電流位相制御部、8 増幅器、9 電動機、1
0 検出器、11 減衰値演算部、12 微分器、21
動作角:α、22 界磁電流成分:SFC(Id)、
23 電機子電流成分:STC(Iq)、24 合成電
流:SIC(Io)、31〜36 トルク特性(回転角
−トルク)、41 係数演算部、42 乗算器、43
スイッチ、44,45 係数乗算器、46 極性判定
部、51 係数Ka関数パターン、61〜64 トルク
特性(電機子電流−トルク(界磁電流))、65 トル
ク特性(合成電流−トルク)、71 特性補正曲線、7
2 補正後特性曲線、81,82 係数演算部、83モ
ードスイッチ、91 トルク特性(電流:Io)、92
トルク特性(電流:Io/2)、101 界磁電流演
算部、111〜113 係数演算部関数パターン。1 subtracter, 2 PI controller, 3 operating angle calculator, 4
Characteristic correction unit, 5 current command calculation unit, 6 current amplitude control unit, 7 current phase control unit, 8 amplifier, 9 motor, 1
0 detector, 11 attenuation value calculator, 12 differentiator, 21
Operating angle: α, 22 Field current component: SFC (Id),
Reference Signs List 23 armature current component: STC (Iq), 24 combined current: SIC (Io), 31 to 36 torque characteristics (rotation angle-torque), 41 coefficient calculation unit, 42 multiplier, 43
Switch, 44, 45 coefficient multiplier, 46 polarity judging section, 51 coefficient Ka function pattern, 61 to 64 torque characteristics (armature current-torque (field current)), 65 torque characteristics (synthetic current-torque), 71 characteristics Correction curve, 7
2 Characteristic curve after correction, 81, 82 Coefficient operation unit, 83 mode switch, 91 Torque characteristic (current: Io), 92
Torque characteristics (current: Io / 2), 101 field current calculation unit, 111 to 113 coefficient calculation unit function pattern.
Claims (4)
直流電流を流し、軟磁性体からなる回転子表面の磁気抵
抗が固定子から見て回転方向に高低差を持つよう内部に
磁気絶縁手段を備えた回転子を回転させた場合に回転角
θに対して発生するトルクτが正負の周期的な特性を持
つようなリラクタンス型電動機の制御装置であって、 前記回転子の回転子位置を検出する手段と、 該電動機に電流を流すために、前記回転子位置とトルク
指令値より電流指令を算出する電流指令演算部を具備す
る制御装置において、 トルク指令値の極性により符号が決定され前記回転子位
置に加算される動作角α(0<α<π/2[rad];電気
角)を持ち、回転子速度を参照して変化する関数パター
ンAより係数Aを演算し動作角に乗ずることで回転子速
度により動作角が変化するような動作角演算部を持つこ
とを特徴とするリラクタンス型同期電動機制御装置。A constant DC current is applied to a winding provided in a stator slot, and a magnetic resistance is formed inside the rotor such that a magnetic resistance on a surface of a rotor made of a soft magnetic material has a height difference in a rotational direction when viewed from the stator. A control device for a reluctance motor in which a torque τ generated with respect to a rotation angle θ when a rotor having insulating means is rotated has positive and negative periodic characteristics, wherein a rotor of the rotor is provided. A control device comprising: means for detecting a position; and a current command calculation unit for calculating a current command from the rotor position and a torque command value in order to cause a current to flow through the motor, wherein a sign is determined by the polarity of the torque command value. Has an operating angle α (0 <α <π / 2 [rad]; electrical angle) that is added to the rotor position, and calculates a coefficient A from a function pattern A that changes with reference to the rotor speed. Operating angle by rotor speed A reluctance type synchronous motor control device, characterized in that it has an operation angle calculation unit that changes.
ことで補正後電流指令値とするような係数Bを、回転子
速度を参照して変化する関数パターンBより演算するよ
うな減衰値演算部を持つことを特徴とするリラクタンス
型同期電動機の制御装置。2. A damping value calculation for calculating a coefficient B for obtaining a corrected current command value by multiplying the current command value in a current amplitude control section from a function pattern B changing with reference to the rotor speed. A control device for a reluctance type synchronous motor, characterized by having a section.
器により、電動機駆動効率もしくは出力トルク特性を優
先するように制御が行えるモードを選択できる請求項2
に記載の減衰値演算部を持つことを特徴とするリラクタ
ンス型同期電動機の制御装置。3. A mode in which control can be performed so as to give priority to motor drive efficiency or output torque characteristics by a mode selector that can be set from outside the controller.
A control device for a reluctance type synchronous motor, characterized by having a damping value calculation unit according to (1).
令値が線形な電動機特性を持つように変換し、電動機の
持つ非線形特性を線形のように扱う電動機特性/指令変
換部を持つことを特徴とするリラクタンス型同期電動機
の制御装置。4. A motor characteristic / command conversion unit for converting an input command value into an output command value so as to have a linear motor characteristic and treating a non-linear characteristic of the motor in a linear manner. A control device for a reluctance type synchronous motor, comprising:
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP16142997A JP3290099B2 (en) | 1997-06-18 | 1997-06-18 | Control device for reluctance type synchronous motor |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
JP16142997A JP3290099B2 (en) | 1997-06-18 | 1997-06-18 | Control device for reluctance type synchronous motor |
Publications (2)
Publication Number | Publication Date |
---|---|
JPH1118468A true JPH1118468A (en) | 1999-01-22 |
JP3290099B2 JP3290099B2 (en) | 2002-06-10 |
Family
ID=15734949
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Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
JP16142997A Expired - Fee Related JP3290099B2 (en) | 1997-06-18 | 1997-06-18 | Control device for reluctance type synchronous motor |
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JP (1) | JP3290099B2 (en) |
Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2009033875A (en) * | 2007-07-27 | 2009-02-12 | Okuma Corp | Control device of reluctance-type synchronous motor |
JP2013070621A (en) * | 2013-01-21 | 2013-04-18 | Okuma Corp | Control device for reluctance synchronous motor |
JP2013085377A (en) * | 2011-10-11 | 2013-05-09 | Mitsubishi Electric Corp | Synchronous machine controller |
Families Citing this family (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP7374620B2 (en) | 2019-06-17 | 2023-11-07 | 株式会社荏原製作所 | Pump tongue member, pump device, and tongue repair method |
-
1997
- 1997-06-18 JP JP16142997A patent/JP3290099B2/en not_active Expired - Fee Related
Cited By (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP2009033875A (en) * | 2007-07-27 | 2009-02-12 | Okuma Corp | Control device of reluctance-type synchronous motor |
JP2013085377A (en) * | 2011-10-11 | 2013-05-09 | Mitsubishi Electric Corp | Synchronous machine controller |
JP2013070621A (en) * | 2013-01-21 | 2013-04-18 | Okuma Corp | Control device for reluctance synchronous motor |
Also Published As
Publication number | Publication date |
---|---|
JP3290099B2 (en) | 2002-06-10 |
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