JPH10163811A - Filter circuit for orthogonal biaxial signal - Google Patents

Abstract

PROBLEM TO BE SOLVED: To secure a steep filter characteristic with narrow band width by extracting only a fundamental wave angular frequency component having a time change from an orthogonal biaxial voltage signal that is converted into two phases from three phases and eliminating other components to convert them into three phases. SOLUTION: Two outputs of a conversion circuit 17 is represented as a complex variable U=Ux+jUy of input signals Ux and Uy of the orthogonal (x) and y axes and then supplied to an orthogonal biaxial filter circuit 25. Then, the transfer coefficients of low and high pass filters of real coefficients are referred to as F(s), the desired center angular frequency to be transmitted (band width 2a) or blocked (band width 2b) as ω, the imaginary unit as (j), and the complex coefficient therefor function in which the Laplacian operator (s) of the transfer coefficient F(s) is replaced by (s-jω) is referred to as F(s-jω), respectively. Under such conditions, the transfer expression U.F (s-jω)=V which is the result of the variable U passed through the function F(s-jω) represented in terms of a complex variable V=Vx+jVy of the output signals Vx and Vy is multiplied by the denominator of F(s-jω) to make the real and imaginary parts equal to each other.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a filter circuit for a quadrature two-axis signal suitable for controlling an inverter for outputting a sine wave voltage of an arbitrary frequency or controlling a rotary motor.

[0002]

2. Description of the Related Art In an inverter device whose basic AC frequency is variable, the output is generally a rectangular wave produced by frequency width modulation (PWM). However, the rectangular wave has many harmonic components, so that noise is likely to be generated. A method has been proposed in which a sine wave waveform in which the corner is rounded and harmonic components are eliminated is output. At this time, a method is employed in which only harmonic components are extracted by a normal real coefficient high-pass filter and fed back. However, since the high-pass filter feeds back without removing the fundamental component of the low frequency completely, especially when the frequency of the fundamental component increases, the difference between the high and low frequencies narrows, so it becomes more and more difficult to remove. There is a problem that the performance of the device deteriorates.

In a conventional system using a filter of a real coefficient transfer function, not only two close resonance frequencies of the same sign but also resonance frequencies of the opposite signs (+,-) having the same or very close absolute values. Was difficult to separate. For example, if there are two resonance modes of opposite signs whose absolute values are very close, one of them has been given sufficient attenuation for some reason, so please do not touch that mode, but If only the other mode is unstable, only the unstable mode needs to be selected. However, in the conventional filter as described above, since only the mode cannot be selected, both modes are controlled at the same time, and a mode that does not need to be touched may be adversely affected.

As a countermeasure, as a tracking filter, a rotating coordinate system in which a stationary coordinate system is rotated at the angular velocity of the mode is assumed. When viewed from the rotating coordinate system, the angular velocity component has a frequency of zero. There is known a filter in which the DC component is separated by a high-pass or low-pass filter, and is again transformed back into a stationary coordinate system. The role of such a filter is not to separate a specific AC component from multiple AC components,
Only the DC component converted to the rotating coordinate system is separated, and separation by a high-pass or low-pass filter is extremely easy.

In the inverter control device, the fundamental component is removed by a high-pass filter in a coordinate transformation from the stationary coordinate system to the rotating coordinate system so that the fundamental component can be regarded as a direct current. A method has been proposed in which coordinates are converted back to.

FIG. 16 shows an example of a control device for a sine wave output inverter. AC power of a commercial voltage and frequency is input from a commercial three-phase AC power supply
Outputs equivalently an AC power of an arbitrary voltage and frequency in accordance with a command from the pulse width modulation circuit 15 and supplies the AC power to the load 13. The resonance circuit 14 is a pulse width modulated inverter 1
Since the output of No. 2 is a rectangular wave, this high-frequency component is removed to form a voltage waveform close to a sine wave. The output voltage waveform to the load 13 is converted into signals of two orthogonal axes by a three-phase / two-phase conversion circuit 17, and is converted from a stationary coordinate system to a rotating coordinate system by a coordinate conversion circuit 18. This conversion is performed by a matrix operation of the matrix C. Since the fundamental wave component of the signal converted into the rotating coordinate system becomes a DC component, the DC component is removed by the high-pass filter 19, and the gain K is multiplied by the arithmetic circuit 20. And
The coordinate conversion circuit 21 converts the data into a stationary coordinate system again by the matrix operation of the matrix C-1, and further converts into a three-phase signal by a two-phase three-phase conversion circuit. Then, the adder 23 subtracts this signal from the signal of the fundamental signal generation circuit 16 to feed back the signal to the input side of the inverter.

It is well known that a speed control circuit including an inverter, such as a motor or a synchronous machine, is designed and examined by converting it into orthogonal coordinates called dq coordinates or γ-δ coordinates. . However, in the present invention described below,
No consideration is given to the application of a filter circuit or a complex gain utilizing a complex coefficient transfer function as a two-input / two-output system.

[0008]

In the above-described example of the inverter control apparatus, since the fundamental wave is an alternating current, the rotating coordinate system has a constant rotation angle, that is, is not stationary, and continues to rotate. Therefore, for coordinate transformation, a sine wave and a cosine wave of the rotation angle of the rotating coordinate system are constantly generated as time functions as shown in FIG. It requires a circuit, a so-called resolver circuit. In addition, the resolver circuit also needs an inverse transformation to the original stationary coordinate system, so that two sets are required. If the resolver circuit is configured by an analog circuit, many elements are required as hardware, and a complicated circuit configuration is required. Further, the digital method has a disadvantage that the calculation time is long.

In the above-mentioned conventional method, it has been impossible to adjust the phase of the signal of the two orthogonal axes to an arbitrary value irrespective of the frequency similarly to the gain. Further, it has been conventionally impossible to sharply change the phase by ± 180 ° only during a specific angular frequency.

SUMMARY OF THE INVENTION The present invention has been made in view of the above circumstances, and provides a sharp filter characteristic with a narrow bandwidth without performing conversion / inverse conversion from a stationary coordinate system to a rotating coordinate system. It is an object of the present invention to provide a two-axis signal filter circuit having good filtering characteristics. In other words, the present invention provides a filter circuit that facilitates the separation of, for example, two modes of very close frequency or two modes with opposite signs of equal or very close absolute values in a control device of an inverter. To provide.
Further, an arbitrary gain and phase are given to a specific frequency range, and the phase is sharply adjusted only during a specific angular frequency.
An object of the present invention is to provide a filter circuit for orthogonal two-axis signals that can be changed by 180 °.

[0011]

According to the first aspect of the present invention, an input signal relating to two orthogonal axes (the Laplace transform amounts thereof are Ux and Uy, respectively) is converted into one complex variable (U = Ux + j).
Uy), the transfer function of a low-pass or high-pass filter with real coefficients is F (s), the central angular frequency to be passed or blocked is ω, the imaginary unit is j, and the Laplace operator s of the transfer function F (s) is The complex coefficient transfer function replaced with (s−jω) is F (s−jω), and the input signal of the complex variable U passes through the complex coefficient transfer function F (s−jω). In the transfer expression U · F (s−jω) = V obtained by expressing each of the Laplace transform amounts by one complex variable (V = Vx + jVy), the denominator of F (s−jω) is After multiplying both sides of the transfer expression, they are connected by transfer elements of real coefficients such that the real part and the imaginary part of the transfer expression are equal to each other. It is characterized by multiplication as an input.

The invention according to claim 2 of the present invention is characterized in that the two positive or negative central angular frequencies to be passed and blocked are ω1 and ω2 of constant or time functions, respectively, and ω1 ≠ ω2, with ω1 being the center. To pass components including the bandwidth (2a) before and after it and to block components including the bandwidth (2b) before and after ω2, and / or to change the phase around ω1 and ω2 Therefore, assuming that k is an arbitrary gain or coefficient and z is an arbitrary dimensionless number, the complex coefficient transfer function is k (s + b−jω2) / (s + a−jω
1) or k (s + b−jω2) / {(s−jω1) 2 + 2za (s−jω1) +
a2}.

According to a third aspect of the present invention, a numerator and / or a denominator of the transfer function F (s) in the first aspect includes:
k is an arbitrary real gain or coefficient, 2a is a bandwidth centered on the central angular frequency ω of passing or blocking, and a secondary system represented by k (s2 + 2zas + a2) is included, and s is replaced by (s−jω). ,
By arbitrarily selecting the dimensionless constant z, the vicinity of a break point at both ends of the bandwidth in the gain characteristic of the frequency characteristic of the filter is made closer to an ideal filter.

According to a fourth aspect of the present invention, a first aspect is provided.
Or a combination of a plurality of orthogonal two-axis signal filter circuits described in any one of (3) to (3) so as to pass or block a specific band in the entire frequency band, or change the phase with the center angular frequency as a boundary. It is characterized by doing so.

According to a fifth aspect of the present invention, a first aspect is provided.
The complex coefficient transfer function of the orthogonal two-axis signal filter circuit according to any one of (1) to (4) is multiplied by a complex gain (A + jB) in which A and B are real constants. , Wherein an arbitrary gain and phase are given to a specific frequency region.

According to a sixth aspect of the present invention, there is provided an inverter control device for outputting a sinusoidal voltage of an arbitrary frequency.
Applying a coordinate transformation to the phase output and inputting the signals of the two orthogonal axes to the filter circuit according to any one of claims 1 to 5,
The inverter device which passes or blocks a specific band in the entire frequency band, or changes the phase of the output signal with the center angular frequency as a boundary, and converts the output signal to polyphase if multi-phase. It is characterized by providing feedback to the user.

[0017]

FIG. 1 is a diagram showing a filter circuit according to a first embodiment of the present invention. This is, for example,
This is a configuration example of a circuit in a part of the inverter control circuit shown in FIG. 16 described above, in which a low-pass filter opposite to the high-pass filter of FIG. That is, a voltage signal of two orthogonal axes converted from three phases to two phases is input, and only the component of the time-varying fundamental wave angular frequency ω included in the input signal is extracted to sharply remove other angular frequency components. The signal is
It is an example of a tracking filter circuit that outputs an input to a phase-to-three-phase conversion circuit. In FIG. 16, θ is set as another path input, but in FIG. 1, the angular velocity ω which is the rate of change is set as another path input. The relationship between the two is θ = ∫ωdt
It is.

First, two outputs from the conversion circuit 17 in FIG. 16 are input with orthogonal x and y two-axis input signals U = (Ux, Uy). It is assumed that the output signal is V = (Vx, Vy). The real coefficient filter used for this filter is created based on the following transfer function F1 (s) of the first-order low-pass filter. F1 (s) ≡k / (s + a) (1)

The Laplace operator s of this transfer function F1 (s)
Is replaced by (s−jω), and the complex coefficient transfer function is represented by F (s−
jω), F1 (s−jω) = k / (s + a−jω) (2) The Laplace transform amounts of the x- and y-axis related input signals are Ux and Uy, respectively, and one complex variable U = Ux + jUy Where U represents the x- and y-axis-related output signals passing through the complex coefficient transfer function F (s-jω) (the Laplace transform amounts thereof are Vx and Vy, respectively) as one complex variable V = Vx + jVy. The transfer expression U · F (s−jω) = V (3) obtained as follows is k (Ux + jUy) / (s + a−jω) = Vx + jVy (4) When both sides of the above expression are multiplied by the denominator (s + a-jω) and expanded to correspond to the real part and the imaginary part, Ux−aVx−ωVy = sVx / k (5) Uy−aVy + ωVx = sVy / k (6)

When this is materialized, a filter circuit as shown in FIG. 1 is obtained. That is, an integrator 1 / s is provided in a signal path of each of the x and y axes, and a negative feedback 1 is given to each of the signal paths for a, which is a half of the pass bandwidth, with a being a gain.
The following low-pass filter is configured. Then, the output of each integrator is multiplied by a non-constant angular velocity signal ω from another path, and then the feedback crossing between the x and y axes, the feedback from the x axis to the y axis is positive, and the feedback from the y axis to the x axis Feedback is made negative. The angular frequency component is selectively passed by this feedback.

In this embodiment, ω = 3750 [rad /
s] and FIG. 2 shows a Bode diagram when a = k = 7.5 [rad / s]. As a result, the effect of passing the band including the central angular frequency ω with the sharp characteristic of selectivity is shown.

Incidentally, in the conventional biquad filter corresponding to the filter having the configuration shown in FIG. 1, V = {2zωs / (s2 + 2zωs + ω2)} U (7) z is a dimensionless number that gives the steepness of the filter, and corresponds to an attenuation ratio or a reciprocal of selectivity.
Equation (7) is a real number system and should be provided separately for the two axes x and y. FIG. 1 shows a specific block diagram thereof.
8 Integrators require a total of four for each two axes.
Further, when z is fixed, the slope of attenuation is constant in the Bode diagram, but when the horizontal axis is set to a linear scale, the steepness decreases in a high frequency region, and the bandwidth increases equivalently. In order to eliminate these drawbacks in this configuration, it is necessary to provide a large number of multipliers by making z a function of ω and providing z and ω in different paths. Therefore, the cost or the computation time for digital computation increases.

On the other hand, the complex filter of FIG. 1 has the advantage that the bandwidth remains unchanged at a, which is simple and simple and requires only about half the hardware, and the digital control requires only a short calculation time. Note that if ω is constant, the multiplier becomes unnecessary, but in the conventional example shown in FIG.
Since changes from moment to moment, matrix calculations using sine and cosine wave function generators and multipliers remain indispensable.

FIG. 3 shows a second embodiment of the present invention, which is a so-called notch filter which blocks only a signal having a specific angular frequency ω. The real coefficient transfer function that is the basis of this notch filter is: F2 (s) ≡k (s + b) / (s + a), a, b> 0 (8) The complex coefficient transfer function calculates s of this denominator and (s−j
ω) is replaced by F2 (s−jω). The specific connection method is omitted because it is a special case where ω1 = ω2 = ω in F3 described later. Also, b may be zero.

When this is materialized, a filter circuit as shown in FIG. 3 is obtained. That is, a direct connection path having a gain of k is provided in the signal path of each axis, and a first-order lag element 1 / (s + a) is provided in parallel with the signal path, and the input side has a 1st delay element 1 / (s + a). A path (a−b) obtained by subtracting b, which is １ ／ of the rejection bandwidth, from the breakpoint angular frequency a of the secondary delay element, multiplied as a gain, returned to the direct connection path, and subtracted and coupled; And a crossing path for crossing between the axes from the output of the delay element and feeding back to the primary delay element. In this crossing path, the angular frequency ω to be blocked is inserted as a variable from another path, and the crossing feedback signal from the output of the first-order lag element is multiplied by a multiplier. Path is addition, y
The path from the axis to the x-axis is connected in a subtractive manner.
Thereby, as a result, the signal of the angular frequency component is canceled and the passage is prevented. This filter circuit can prevent the passage of only one of the positive and negative angular frequencies, and can compensate for maintaining a stable operation by giving an appropriate phase to the remaining signal. If ω is constant, a multiplier is not required, and a block having ω as a gain or coefficient is used instead.

When b>a> 0, the filter becomes a pass filter. When k = a / b, the bottom of the bottom of the pass characteristic is 0 [dB].
Becomes Since there is only a phase change with almost no change in gain near the stop or outside the pass band (see FIG. 4), this feature can be utilized for control.

FIG. 5 shows a third embodiment of the present invention and relates to a bilinear tracking filter circuit. This is a combination of the first and second embodiments, and functions functionally equivalent to the case where the above two filter circuits are connected in series. That is, a filter circuit for passing the positive or negative angular frequency component ω1 necessary for the signal configuration for controlling the two axes of x and y, and blocking the other positive or negative angular frequency component ω2. is there. From the transfer function F3 分 k (s + b) / (s + a) (9) when both the denominator and the first order are (first order), s of the denominator and (s−jω1) and (s−jω
The complex coefficient transfer function F3≡k (s + b−jω2) / (s + a−jω1) obtained by replacing with (2) is expressed by (10). First, by reducing the dimension of the molecule, F3 = k {s + a−jω1− (ab) + j (ω1−ω2)} / (s + a−jω1) = k + k [{j (ω1−ω2) − (a− b)} / (s + a−jω1)] (11) Therefore, multiplying the input by k and passing it through,
It can be decomposed into a first-order lag element including the complex constant of the second term or a low-pass filter. Here, the configuration method of only the second term will be described.

The second term excluding k is [{ba + j (ω1−
ω2)} / (s + a−jω1)] multiplied by the input U = (Ux + jUy), the output V2 = V2x + jV2y, the real part and the imaginary part of the complex number equation after multiplying both sides by the denominator are Ux (ba) −Uy (ω1−ω2) = V2x (s + a) + V2yω1 (12) Ux (ω1−ω2) + Uy (ba) = V2y (s + a) −V2xω1 (13) From equation (12), V2x = [Ux (ba) -Uy (ω1−ω2) −V2yω1] / (s + a) (14) Similarly, from equation (13), V2y = [Ux (ω1−ω2) + Uy (ba) + V2xω1] / (s + a) (15) is obtained. Multiplying the denominator by the conjugate of the denominator in equation (10) to make the denominator a real number is not desirable because it increases the number of components, increases the order of the system, and increases the calculation time in digital control.

The connections according to the equations (14) and (15) are shown in FIG.
Is shown in the portion of the dotted line other than the pass through. That is, x,
The signal path of each axis y is provided with a direct connection path having a gain of 1 and two parallel circuits branched from each of the paths.
One of the parallel circuits gives, as a gain, an amount (ba) obtained by subtracting the corner frequency a of the low-pass filter LPF connected to the rear from half of the stop band width. Input to LPF. On the other hand, (ω1−ω2) is given as a gain and connected to the input of the low-pass filter LPF crosswise. This connection is such that the path from the first axis to the second axis is added and the path from the second axis to the first axis is subtracted.

The same low-pass filter LPF and ω1 as in FIG.
The circuit including the cross feedback constitutes a band-pass filter, which is the same as that described with reference to FIG. 1, in which ω is replaced by ω 1 in FIG. That is, an integrator is provided in the signal path of each axis, and each of them provides negative feedback with a which is の of the pass bandwidth. Feedback from each integrator crossing between the x and y axes is applied so that the feedback from the x axis to the y axis is positive, and the feedback from the y axis to the x axis is negative. If the angular frequency ω1 to be passed is a constant, it is inserted as a gain, and if it is a time function, the multiplier multiplies the output of the LPF. The output of the bandpass filter as a whole is summed and connected to a direct connection path.

This example is particularly effective when ω1 ≒ ω2. In other words, only the mode of ω1 is selected and ω2
This is the case when you do not want to touch the mode. a = b = 10 [ra
d / s], ω1 = 5000 [rad / s] and ω2 = 5100 [rad / s] are Bode diagrams in FIG. In spite of such close frequencies and considerable bandwidth, the gain difference between passing and blocking is about 40 [dB], demonstrating the effectiveness of this filter circuit. According to this circuit configuration, only two integrators are required.

FIG. 7 shows an example in which in the circuit of FIG. 5, the phase angle of only a certain angular frequency band is delayed by 180 degrees, and the phase in other angular frequency regions is not affected. ω1 = 10000 [rad / s], ω2 = 40000 [rad / s], b = 0, a =
This is the case of 1 [rad / s], and the phase is sharply reversed over the bandwidth between the two angular frequencies by setting a to be small enough not to oscillate. However, the gain characteristics are not completely flat. If ω1 and ω2 are reversed, the phase between them will be reversed by 180 degrees, and the slope of the gain will also be reversed.

Next, an example will be described in which a transfer function is a real number coefficient and a second-order low-pass filter represented by the following equation is used as a band-pass filter having a steep gain characteristic. F4≡ka2 / (s2 + 2zas + a2) (16) z of this filter is usually called an attenuation ratio.
It is common to set <1 to make the bump near the break point thinner and higher as shown in FIG. However, in such a usage method, the high-frequency side of the bump has a steep characteristic to some extent, but the low-frequency side has a disadvantage that the bottom is shallow, so it is general to multiply the numerator of equation (16) by s. The shape design including the width of the bump was complicated.

In the present invention, steep characteristics can be obtained even with the secondary filter, and not only the bandwidth but also the shape of the bump can be freely selected, and the sign of the pass band can be freely selected. If s in equation (16) is replaced with (s-jc) in order to take any center angular frequency c irrespective of positive or negative, the relationship with inputs U and V is obtained by multiplying both sides by a denominator including complex numbers. Ka2U = {(s−jω) 2 + 2za (s−jω) + a2} V = {(s2 + 2zas + a2−ω2) −j2 (ωs + zaω)} (Vx + jVy) (17) Since the real part and the imaginary part of this equation are equal. {Ka2Ux-2 (ωs + zaω) Vy} / (s2 + 2zas + a2−ω2) = Vx (18) {ka2Uy + 2 (ωs + zaω) Vx} / (s2 + 2zas + a2−ω2) = Vy (19) Is divided by a2, and a materialized circuit is a complex second-order filter shown in FIG. In the figure, k is omitted. However, ω is a constant, that is, in the case of the inverter described above, the fundamental frequency is unchanged. ω
Is variable, ω in the figure must be given from another path and calculated by a multiplier.

Both x and y axes are secondary circuits.
It has extremely advanced functions and performance by crossing between them. Numerical examples are as follows: a = 250 [rad / s], ω = 5000 [rad / s]
FIG. 9 shows three Bode diagrams in which z = 0.5, 0.707, and 1.0 in the case of [s]. However, the abscissa was a linear scale of angular frequency. The bandwidth can be given by 2a, and a proper choice of z helps to improve the corners of the bandwidth. The gain characteristic becomes steeper as the bandwidth is smaller.

FIG. 10 shows still another embodiment of the present invention, and is an example of a circuit for providing a gain and a phase for converting signals on the x and y axes into signals necessary for control. . A required angular frequency component or band is extracted from input signals from two orthogonal axes regardless of positive or negative, and a required phase angle is given to the component or band and output, and the output signal is effectively used for control and the like. It is for doing. At that time, assuming a fictional coordinate system that rotates at the angular frequency to be extracted or removed with respect to the stationary coordinates, it is only necessary to extract or remove as a DC component and return to the original stationary coordinate system. This can be realized only by the complex filter 5 on the left. Therefore, they are the passing or blocking filters described with reference to FIGS. 1 to 3 or a combination filter thereof, which respectively pass different frequencies or remove nearby disturbing frequency components. If the frequency is variable, it must be introduced from another path and calculated by a multiplier.

As shown in FIG. 11, the complex gain circuit 6 on the right side of the figure branches each of the x and y axes, giving a constant A as a gain to one path and a constant B as a gain to the other path. The shafts intersect with each other and are connected at the rear of the A. At this connection point, the path from the x axis to the y axis is added so that the path from the y axis to the x axis is subtracted.

In such a complex gain circuit,
Arbitrary phase angle tan-1 by selecting A and B arbitrarily
The output phase can be increased or decreased by [B / A] regardless of the input frequency. The gain at this time is (A2 + B2) 1/2.

FIG. 12 shows an application example of the filter circuit having the above configuration to a control device. In the figure, (A) shows the entire configuration of the inverter control device, and (B) shows the orthogonal 2 in (A).
A circuit configuration in which the axis filter circuit 25 is replaced with a real number system is shown. That is, the orthogonal two-axis filter circuit 25 of the present invention
It is inserted between two 3-phase / 2-phase conversion circuits shown in FIG. 16, and a coordinate conversion circuit is not required at all. The difference from FIG. 16 is that the control is performed by ω (t) without using the command voltage θ (v), and that the numerator is not simply s but s + b. The reason is that some delay in signal transmission is unavoidable, especially in the case of digital circuits,
The actually generated ω (t) does not always completely match the ω (t) of the command voltage v +, and may include a small error. For that reason, it is safer to give the notch characteristic a slight width. The circuit itself is the same as in FIG. 3, and since ω1 = ω2 in FIG. 5, the number of multipliers is reduced by half.

FIG. 13 shows a case where the present invention is applied to, for example, a control device for a magnetic bearing. In the case where the control object controlled from two orthogonal axes is 1 / (s2 + ω2) and there are two resonance modes of ± jω. It is a circuit example for giving attenuation. That is, an example has been shown in which a control signal is generated by multiplying the ± jω mode by an individual gain. For example, in the simplest simple vibration mode, two vibration modes of front rotation + ω and rear rotation -ω are generated on the x and y planes by the restoring force. Now suppose we want to give separate attenuation to both modes. For example, one may not want to touch because one has some attenuation from the other.
Including them, the plus mode is given attenuation of the gain of k +, and the opposite mode is given attenuation of the gain of k-. Since both vibration modes are circular motions in the opposite direction when viewed from the stationary coordinates, the damping is to apply a force in the tangential direction of the brake circle.

When the input signals of x and y are the displacement sensor signals to be controlled, and in the plus mode, a coordinate system rotating at ω with respect to the stationary coordinates is assumed, the plus mode is stationary for the coordinate system. And the minus mode rotates at -2 [omega], so that the two can be easily separated. Therefore, as described in the example of the inverter, since the DC component is selected by performing the coordinate conversion and the component is returned to the original stationary coordinates again, the complex filter circuit of FIG. 1 can be used. However, since ω is a constant, no multiplier is required, and ω enters the position of the multiplier as a gain. Since only + ω is selected as the output of the filter, the tangential direction, which is the braking direction, is a lagging direction perpendicular to the displacement direction.
k + complex gain. This corresponds to a specific example of the fifth aspect of the present invention.

Similarly, in the minus mode, ω in FIG. 1 is reversed, and the brake is also multiplied by the opposite complex gain of + jk−. In the complex gain circuit, in FIG. 11, the positive mode is set to A = 0, B = + k +, the negative mode is set to A = 0, and B = −k−. A gain of k + and a frequency of -ω receive a phase change of +90 degrees and a gain of k-. FIG. 14 shows the dynamic characteristic of the controlled object in the example of FIG. 13 as 1 / (s2 + ω
FIG. 2 is a complex expression block diagram expressed as 2). The operation of the complex gain circuit 6 can be arbitrarily determined whether the operation is advanced or delayed. Since the gain does not require dynamic characteristics, even if the phase is advanced, the gain does not increase unless both are increased while maintaining the ratio B / A. A plurality of such frequency (band) selection paths may be provided as necessary to provide a gain and a phase necessary for stabilization.

FIG. 15 shows a case where the object to be controlled has the same dynamic characteristics as that of FIG. 14, but the final stage actuator which generates the control force has a first-order delay having a large time constant T. 1 / (1 + Ts), where the + jω mode is sufficiently attenuated by some influence, but only the −jω mode is underdamped. Therefore, in this case, it is sufficient to select only the −jω mode component, give a constant complex gain k, and feed back. The Laplace transform of the motion equation of the whole system including this feedback is as follows: k is an unknown complex number, and s2 + ω2 = k / [(1 + Ts) (s + a + jω)] (20) The characteristic equation of this system is F (s) = (s2 + ω2) (1 + Ts) (s + a + jω) −k = 0 (21) When Taylor expansion is performed near s = −jω, approximately F (s) | s = −jω = −2jωa (1-jωT) (s + jω ) −k (22) and the approximate solution in the vicinity is s = −jω + k / {− 2jωa (1-jωT)} (23) Since the second term on the right side is an attenuation term, it is desirable that the second term be a negative real number. Therefore, if k = j (1−jωT) k (24), s = −jω−k / 2ωa (25) where k is a positive real number and corresponds to a normal feedback gain.

Eventually, the expression (20) becomes: s2 + ω2 = jk (1-jωT) / {(1 + Ts) (s + a + jω)} (26) Here, (1-jωT) is newly added to the numerator in order to remove the influence of the first-order lag. The specific connection of the numerator, excluding the selection filter, is given by (x + ji) jk (1-jωT) = k [-y + xωT + j (x + yωT)] (27), and the connection shown in FIG. Becomes With such a configuration, in this control circuit, ± ω
Of these, only −ω can be passed to compensate for the delay of the electromagnet to provide damping.

[0045]

As described above in detail, according to the present invention, a high-pass or a low-pass is applied to an input signal of two orthogonal axes without using a transformation / inverse transformation from a stationary coordinate system to a rotating coordinate system. Various filter characteristics such as a sharp bandpass characteristic can be obtained by using the filter. Therefore, for example, by using it in a feedback circuit of a control device of an inverter that forms a sine wave voltage waveform, the circuit configuration can be greatly simplified and good filter characteristics can be provided.

In particular, according to the two-axis signal filter circuit, two modes having extremely close frequencies and two modes having opposite signs of equal or close absolute values are present in the two-axis signal. It is possible to select only one of the modes and estimate the state quantity or apply a predetermined control rule. Therefore, with a relatively simple device and software, it is possible to create filtering and related control signals suitable for the operating conditions of the orthogonal two-axis signal, and to control the inverter, the rotation motor, and the magnetic bearing of the basic AC frequency variable. Make the functions such as control more effective.
Further, in the filter circuit or circuit system for a two-axis signal of the rotating coordinate axis according to the present invention, it is possible to simultaneously adjust the gain and the phase, which have been impossible in the conventional method, independently of the frequency. Control suitable for operating conditions such as frequency-variable inverter control, rotation motor control, or magnetic bearing control can be performed.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a first embodiment of the present invention.

FIG. 2 is a graph showing characteristics of the circuit according to the embodiment of FIG. 1;

FIG. 3 is a block diagram showing a second embodiment of the present invention.

FIG. 4 is a graph showing characteristics of the circuit according to the embodiment of FIG. 3;

FIG. 5 is a block diagram showing a third embodiment of the present invention.

FIG. 6 is a graph showing characteristics of the circuit according to the embodiment of FIG. 5;

FIG. 7 is a graph showing characteristics of another example in which constants are changed in the circuit of the embodiment of FIG. 5;

FIG. 8 is a block diagram showing a fourth embodiment of the present invention.

FIG. 9 is a graph showing characteristics of the circuit according to the embodiment of FIG. 8;

FIG. 10 is a block diagram of a control device according to an embodiment of the present invention.

FIG. 11 is a block diagram for realizing a complex gain according to another embodiment of the present invention.

12A is a block diagram of an inverter control device according to an embodiment of the present invention, and FIG. 12B is a block diagram of a quadrature two-axis filter circuit.

FIG. 13 is a block diagram of a control device using a control circuit for a rotating body according to another embodiment of the present invention.

FIG. 14 is a block diagram showing a transfer function of the control circuit of FIG.

FIG. 15 is a block diagram of a control device using a control circuit for a rotating body according to another embodiment of the present invention.

FIG. 16 is a block diagram of a conventional inverter control device.

FIG. 17 is a diagram showing a conventional filter circuit.

FIG. 18 is a graph showing characteristics of a conventional secondary filter circuit.

[Explanation of symbols]

 Reference Signs List 5 complex bandpass filter 6 complex gain 12 inverter 15 pulse width modulation circuit 16 fundamental signal generation circuit 23 adder 25 orthogonal 2-axis filter circuit

Claims (6)

[Claims] 1. An input signal relating to two orthogonal axes (the Laplace transform amounts thereof are Ux and Uy, respectively) is converted into one complex variable (U = Ux
+ JUy), the transfer function of a low-pass or high-pass filter with real coefficients is F (s), the central angular frequency to be passed or blocked is ω, the imaginary unit is j, and the Laplace operator s of the transfer function F (s) is The complex coefficient transfer function replaced with (s−jω) is F (s−jω), and the input signal of the complex variable U passes through the complex coefficient transfer function F (s−jω). In the transfer expression U · F (s−jω) = V obtained by expressing each of the Laplace transform amounts by Vx, Vy) by one complex variable (V = Vx + jVy), the denominator of F (s−jω) is After multiplying both sides of the transfer expression, they are connected by transfer elements of real coefficients such that the real part and the imaginary part of the transfer expression are equal to each other. A filter circuit for quadrature two-axis signals characterized by multiplying as an input. 2. The two positive or negative center angular frequencies to be passed and blocked are respectively expressed by a constant or a time function ω.
1 and ω2, and as ω1 ≠ ω2, passes components including the bandwidth (2a) before and after ω1 and blocks components including the bandwidth (2b) before and after ω2. And / or to change the phase with ω1 and ω2 as boundaries, where k is any gain or coefficient and z is any dimensionless number, the complex coefficient transfer function is k (s + b−jω2) / ( s + a−jω1) or k (s + b−jω2) / {(s−jω1) 2 + 2za (s−jω1) +
a2}, the filter circuit for orthogonal two-axis signals according to claim 1, wherein 3. A bandwidth around a central angular frequency ω of passing or blocking 2a, where k is an arbitrary real gain or coefficient, in the numerator and / or denominator of the transfer function F (s) according to claim 1. Including a quadratic system represented by k (s2 + 2zas + a2), replacing s with (s-jω) and arbitrarily selecting a dimensionless constant z to divide the bandwidth characteristic at both ends of the bandwidth in the gain characteristic of the frequency characteristic. 2. The filter circuit for quadrature two-axis signals according to claim 1, wherein the vicinity of the point is made closer to the ideal filter. 4. A combination of a plurality of quadrature two-axis signal filter circuits according to any one of claims 1 to 3, so as to pass or block a specific band in the entire frequency band, or to control the center. A filter circuit for orthogonal two-axis signals, wherein the phase is changed with an angular frequency as a boundary. 5. The complex coefficient transfer function of the orthogonal two-axis signal filter circuit according to claim 1 is multiplied by a complex gain (A + jB) where A and B are real constants. A quadrature two-axis signal filter circuit, wherein a circuit is configured by similar means to give an arbitrary gain and phase to a specific frequency region. 6. A control device for an inverter that outputs a sine wave voltage of an arbitrary frequency, wherein if the inverter device is a polyphase output, a coordinate transformation is performed on a two-phase output of two orthogonal axes, and a signal of the two orthogonal axes is output. An output which is input to the filter circuit according to any one of claims 1 to 5, and which passes or blocks a specific band in the entire frequency band, or changes the phase around the center angular frequency as a boundary. A control device for an inverter, wherein a signal is converted into a polyphase signal if the signal is polyphase, and is fed back to the inverter device.