JP2013085435A - Control circuit for power conversion circuit, and system interconnection inverter system and three-phase pwm converter system using the control circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control circuit that performs such processing as is same as resting coordinate transformation of harmonic compensation signals generated by predetermined control following rotating coordinate transformation and as has linear time invariance.SOLUTION: A control circuit 7 includes: a three-phase-to-two-phase conversion section 73 for converting three-phase current signals to an α axis current signal and a β axis current signal; a fifth harmonic compensation section 81 for applying signal processing by a first transfer function and phase adjustment processing to each of the α axis current signal and the β axis current signal to generate a first harmonic compensation signal and a second harmonic compensation signal; and a two-phase-to-three-phase conversion section 76 and a PWM signal generation section 77 for generating PWM signals on the basis of the first harmonic compensation signal and the second harmonic compensation signal. The first transfer function is G(s)={F(s+j*5*ω)+F(s-j*5*ω)}/2, where F(s) is a transfer function representing predetermined control processing.

Description

  The present invention relates to a control circuit for controlling the output or input of a power conversion circuit, a grid-connected inverter system and a three-phase PWM converter system using the control circuit, and more particularly to a circuit for compensating for harmonics.

  2. Description of the Related Art Conventionally, a grid-connected inverter system has been developed that converts DC power generated by a solar cell or the like into AC power and supplies it to an electric power system.

  FIG. 18 is a block diagram for explaining a conventional general grid-connected inverter system.

  The grid interconnection inverter system A100 converts the power generated by the DC power supply 1 and supplies it to the three-phase power system B. Hereinafter, the three phases are referred to as a U phase, a V phase, and a W phase.

  The inverter circuit 2 converts a DC voltage input from the DC power supply 1 into an AC voltage by switching a switching element (not shown). The filter circuit 3 removes a switching frequency component included in the AC voltage output from the inverter circuit 2. The transformer circuit 4 boosts (or steps down) the AC voltage output from the filter circuit 3 to the system voltage of the three-phase power system B. The control circuit 700 receives the current signal and the voltage signal detected by the current sensor 5 and the voltage sensor 6, generates a PWM signal based on the current signal and the voltage signal, and outputs the PWM signal to the inverter circuit 2. The inverter circuit 2 performs switching of the switching element based on the PWM signal input from the control circuit 700.

  FIG. 19 is a block diagram for explaining the internal configuration of the control circuit 700.

  The current signal of each phase input from the current sensor 5 is input to the three-phase / two-phase converter 73.

  The three-phase / two-phase conversion unit 73 converts the three input current signals Iu, Iv, and Iw into an α-axis current signal Iα and a β-axis current signal Iβ. The three-phase / two-phase conversion unit 73 performs a so-called three-phase / two-phase conversion process (αβ conversion process). The current signals Iu, Iv, Iw are respectively converted into an α-axis component and a β-axis component that are orthogonal to each other. The α-axis current signal Iα and the β-axis current signal Iβ are generated by decomposing and collecting the respective axis components.

The conversion process performed by the three-phase / two-phase conversion unit 73 is represented by a determinant represented by the following equation (1).

  The rotation coordinate conversion unit 78 converts the α-axis current signal Iα and the β-axis current signal Iβ input from the three-phase / two-phase conversion unit 73 into a d-axis current signal Id and a q-axis current signal Iq in the rotation coordinate system. Is. The rotating coordinate system is an orthogonal coordinate system having orthogonal d-axis and q-axis and rotating in the same direction at the same angular velocity as the fundamental wave of the system voltage of the three-phase power system B. As a concept opposite to the rotating coordinate system, a non-rotating coordinate system is a stationary coordinate system. The rotation coordinate conversion unit 78 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 78 is expressed by a determinant represented by the following expression (2).

  The LPF 74a and the LPF 75a are low-pass filters and pass only the DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. Through the rotation coordinate conversion process, the fundamental wave components of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. The PI control unit 74b and the PI control unit 75b perform PI control (proportional integration control) based on the deviation between the DC component of the d-axis current signal Id and the q-axis current signal Iq and the target value, respectively, and the fundamental compensation signal Xd and Xq are output. Since a DC component can be used as the target value, the PI control unit 74b and the PI control unit 75b can perform control with high accuracy.

  The stationary coordinate conversion unit 79 converts the fundamental wave compensation signals Xd and Xq input from the PI control unit 74b and the PI control unit 75b, respectively, into two fundamental wave compensation signals Xα and Xβ in the stationary coordinate system. The rotation coordinate conversion unit 78 performs a reverse conversion process. The static coordinate conversion unit 79 performs a so-called static coordinate conversion process (inverse dq conversion process), and uses the fundamental wave compensation signals Xd and Xq of the rotating coordinate system based on the phase θ to compensate the fundamental wave of the static coordinate system The signals are converted into signals Xα and Xβ.

The conversion process performed by the stationary coordinate conversion unit 79 is expressed by a determinant represented by the following expression (3).

  The two-phase / three-phase conversion unit 76 adds the harmonic compensation signals Yα and Yβ output from the harmonic compensation controller 800 described later to the fundamental wave compensation signals Xα and Xβ output from the stationary coordinate conversion unit 79. The value signals X′α, X′β are converted into three correction value signals Xu, Xv, Xw. The two-phase / three-phase conversion unit 76 performs a so-called two-phase / three-phase conversion process (reverse αβ conversion process), and performs a conversion process opposite to the three-phase / two-phase conversion unit 73.

The conversion process performed by the two-phase / three-phase conversion unit 76 is expressed by a determinant represented by the following equation (4).

  The PWM signal generation unit 77 generates and outputs a PWM signal based on the correction value signals Xu, Xv, and Xw output from the two-phase / three-phase conversion unit 76.

  The control circuit 700 has a function of suppressing harmonics input from the three-phase power system B and harmonics output from the inverter circuit 2. The harmonic compensation controller 800 extracts a harmonic component from each phase current signal input from the current sensor 5 and outputs a harmonic compensation signal for outputting a harmonic that cancels the harmonic component. The grid interconnection inverter system A100 outputs harmonics based on the harmonic compensation signal (that is, harmonics having a phase opposite to the detected harmonics) and cancels the harmonics, thereby suppressing harmonics.

  FIG. 20 is a block diagram for explaining the internal configuration of the harmonic compensation controller 800. In general, the harmonics from the three-phase power system B or the inverter circuit 2 are mostly the fifth harmonic, the seventh harmonic, and the eleventh harmonic. In order to suppress these harmonics, a fifth harmonic compensator 810 for suppressing the fifth harmonic, a seventh harmonic compensator 820 for suppressing the seventh harmonic, and an eleventh harmonic are provided. An eleventh harmonic compensation unit 830 for suppression is provided in the harmonic compensation controller 800. The fifth harmonic compensation unit 810 includes a rotation coordinate conversion unit 811, LPFs 812 and 813, I control units 814 and 815, and a stationary coordinate conversion unit 816. Since the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 have the same configuration as the fifth harmonic compensation unit 810, the description and description in FIG. 20 are omitted.

The rotation coordinate conversion unit 811 converts the α-axis current signal Iα and β-axis current signal Iβ input from the three-phase / two-phase conversion unit 73 into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotation coordinate system. To convert. This rotating coordinate system is an orthogonal coordinate system that rotates in the opposite direction at an angular velocity that is five times the angular velocity of the fundamental wave of the system voltage. The rotation coordinate conversion unit 811 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotating coordinate system.

The conversion process performed by the rotating coordinate conversion unit 811 is represented by a determinant represented by the following expression (5).
Note that the rotation coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (5).

LPF 812 and LPF 813 are low-pass filters and pass only the DC components of d-axis current signal Id ₅ and q-axis current signal Iq ₅ , respectively. By the rotational coordinate conversion process, the fifth harmonics of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively. The I control unit 814 and the I control unit 815 perform I control (integration control) based on the DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively, and the fifth harmonic compensation signals Yd ₅ , Yq ₅ is output. Since a DC component can be used as the target value, the I control unit 814 and the I control unit 815 can perform highly accurate control.

The stationary coordinate conversion unit 816 converts the fifth harmonic compensation signals Yd ₅ and Yq ₅ input from the I control unit 814 and the I control unit 815, respectively, into two fifth harmonic compensation signals Yα ₅ and Yβ in the stationary coordinate system. ₅ is converted, and the conversion process reverse to that of the rotating coordinate conversion unit 811 is performed. The static coordinate conversion unit 816 performs so-called static coordinate conversion processing (inverse dq conversion processing), and converts the fifth-order harmonic compensation signals Yd ₅ and Yq ₅ of the rotating coordinate system into a static coordinate system based on the phase θ. of the fifth harmonic compensation signal Yarufa _5, converted into Ybeta _5.

The conversion process performed by the stationary coordinate conversion unit 816 is represented by a determinant represented by the following expression (6).
Note that the phase θ ₅ is for adjusting the phase of the harmonics to be output and is set in advance. For example, in the case where the phase is delayed by 90 degrees in the object to be controlled, θ ₅ = −90 degrees and the phase is reversed by delaying the phase by 180 degrees. In addition, the rotation coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830, in the above formula (6), (−5θ−θ ₅ ) are (7θ + θ ₇ ) and (−11θ−θ), respectively. ₁₁ ) Perform the process described above. The phases θ ₇ and θ ₁₁ are for adjusting the phases of the output harmonics.

Similarly, the seventh harmonic compensator 820 7 harmonic compensation signal Yarufa _7, generates and outputs a Ybeta _7, 11 harmonic compensator 830 11 harmonic compensation signal Yarufa _11, the Ybeta ₁₁ Generate and output. 5 harmonic compensation signal Yarufa _5, 7 harmonic compensation signal Yα _7, 11 th and harmonic compensation signal Yarufa the harmonic compensation signal Yarufa ₁₁ by adding, fifth harmonic compensation signal Ybeta _5, 7 harmonic compensation The harmonic compensation signal Yβ obtained by adding the signal Yβ ₇ and the 11th-order harmonic compensation signal Yβ ₁₁ is output from the harmonic compensation controller 800 to the fundamental wave compensation signals Xα and Xβ output from the stationary coordinate converter 79, respectively. The added values are input to the two-phase / three-phase converter 76 as correction value signals X′α and X′β.

Japanese Patent No. 4421700

  However, there is a problem that a great effort is required to design a control system for each harmonic compensation. In order to design a control system for each harmonic compensation, it is necessary to optimally design the parameters of the LPFs 812 and 813 and the integral gains of the I controllers 814 and 815. However, since the rotating coordinate conversion unit 811 and the stationary coordinate conversion unit 816 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  The present invention has been conceived under the circumstances described above, and is similar to performing a predetermined control after performing a rotational coordinate transformation and performing a stationary coordinate transformation on the generated harmonic compensation signal. It is an object of the present invention to provide a control circuit that performs processing and performs processing having linearity and time invariance.

  In order to solve the above problems, the present invention takes the following technical means.

A control circuit provided by the first aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to three-phase alternating current using a PWM signal, and includes a three-phase control circuit. Three-phase two-phase conversion means for converting three signals based on output or input into a first signal and a second signal, and predetermined harmonic components included in the first signal and the second signal, respectively. Harmonic compensation means for generating a first harmonic compensation signal and a second harmonic compensation signal by performing suppression control, and the first harmonic compensation signal and the second harmonic compensation signal are divided into three Two-phase / three-phase conversion means for converting to a correction value signal; and PWM signal generation means for generating a PWM signal based on the three correction value signals, wherein the harmonic compensation means includes the first signal. By the first transfer function Signal processing and phase adjustment processing to generate the first harmonic compensation signal, signal processing of the second signal by the first transfer function, and phase adjustment processing The second harmonic compensation signal is generated, and the first transfer function is F (s) representing a predetermined control process, and the angular frequency of the fundamental wave of the three-phase alternating current is ω ₀ , When the imaginary unit is j and the nth harmonic is suppressed,
It is characterized by being.

In a preferred embodiment of the present invention, the harmonic compensation means processes the first signal by the first transfer function, and processes the second signal by the second transfer function, By adding these and performing phase adjustment processing, the first harmonic compensation signal is generated, the first signal is signal-processed by a third transfer function, and the second signal is converted to the second signal. Signal processing is performed with the transfer function of 1 and the phase adjustment process is performed by adding them, thereby generating the second harmonic compensation signal, and the second transfer function and the third transfer function are: When suppressing n-order harmonics (n = 3k + 1: k = 1, 2,...), respectively,
And when suppressing the n-th harmonic (n = 3k + 2: k = 0, 1, 2,...)
It is.

  In a preferred embodiment of the present invention, the first fundamental wave compensation signal and the second fundamental wave component are controlled by controlling the fundamental wave components included in the first signal and the second signal to follow the respective target values. Control means for generating two fundamental wave compensation signals, and adding the first fundamental wave compensation signal and the first harmonic compensation signal to produce a first correction value signal, and generating the second fundamental wave compensation signal. A correction value signal generating means for adding a wave compensation signal and the second harmonic compensation signal to generate a second correction value signal, wherein the two-phase / three-phase conversion means includes the first correction signal. The value signal and the second correction value signal are converted into the three correction value signals.

A control circuit provided by the second aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to three-phase alternating current by using a PWM signal, The first harmonic compensation signal is obtained by performing control to suppress predetermined harmonic components included in the first signal, the second signal, and the third signal, which are three signals based on the output or the input, A harmonic compensation unit that generates a second harmonic compensation signal and a third harmonic compensation signal; and a PWM signal generation unit that generates a PWM signal based on the three harmonic compensation signals. The harmonic compensation means processes the first signal with a first transfer function, processes the second signal with a second transfer function, and sends the third signal to the second transfer function. Signal by function And adding these to perform phase adjustment processing, thereby generating the first harmonic compensation signal, subjecting the first signal to signal processing by the second transfer function, The signal is processed by the first transfer function, the third signal is signal processed by the second transfer function, and these are added to perform phase adjustment processing, whereby the second harmonic A compensation signal is generated, the first signal is signal-processed by the second transfer function, the second signal is signal-processed by the second transfer function, and the third signal is processed by the first transfer function Signal processing is performed using a transfer function, and the third harmonic compensation signal is generated by performing phase adjustment processing by adding these signals,
The first transfer function and the second transfer function have a transfer function representing a predetermined control process as F (s), an angular frequency of the fundamental wave of the three-phase alternating current as ω ₀ , an imaginary unit as j, and n When suppressing the second harmonic,
It is characterized by being.

A control circuit provided by the third aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to three-phase alternating current by means of a PWM signal. The first harmonic compensation signal is obtained by performing control to suppress predetermined harmonic components included in the first signal, the second signal, and the third signal, which are three signals based on the output or the input, A harmonic compensation unit that generates a second harmonic compensation signal and a third harmonic compensation signal; and a PWM signal generation unit that generates a PWM signal based on the three harmonic compensation signals. The harmonic compensation means processes the first signal with a first transfer function, processes the second signal with a second transfer function, and converts the third signal into a third transfer function. By signal processing Then, by adding these and performing phase adjustment processing, the first harmonic compensation signal is generated, the first signal is signal-processed by the third transfer function, and the second signal is The second harmonic compensation signal is signal-processed by the first transfer function, the third signal is signal-processed by the second transfer function, and these are added to perform a phase adjustment process. And processing the first signal with the second transfer function, processing the second signal with the third transfer function, and converting the third signal into the first transfer function. The third harmonic compensation signal is generated by performing phase adjustment processing by adding the signals and performing the phase adjustment processing, and the first to third transfer functions are transfer functions representing predetermined control processing. F (s), the angular frequency of the fundamental wave of the three-phase alternating current is ω ₀ , where j is the imaginary number unit and n-th harmonics (n = 3k + 1: k = 1, 2,...) Are suppressed,
And
When suppressing the n-th harmonic (n = 3k + 2: k = 0, 1, 2,...)
It is characterized by being.

  In a preferred embodiment of the present invention, control is performed to cause the fundamental wave components included in the first signal, the second signal, and the third signal to follow the respective target values, and the first fundamental A control means for generating a wave compensation signal, a second fundamental wave compensation signal, and a third fundamental wave compensation signal, and adding the first fundamental wave compensation signal and the first harmonic compensation signal to 1 correction value signal is generated, and the second fundamental wave compensation signal and the second harmonic compensation signal are added to generate a second correction value signal, and the third fundamental wave compensation signal and Correction value signal generating means for adding the third harmonic compensation signal to generate a third correction value signal, wherein the PWM signal generating means includes the first correction value signal, the second correction value signal, A PWM signal is generated based on the correction value signal and the third correction value signal.

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _I / s (where K _I is an integral gain).

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _P + K _I / s (where K _P and K _I are proportional gain and integral gain, respectively). is there.

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _P + K _I / s + K _D · s (where K _P , K _I and K _D are proportional to each other) Gain, integral gain, differential gain).

  In a preferred embodiment of the present invention, the three signals are signals obtained by detecting an output current or an input current of each phase.

  In a preferred embodiment of the present invention, the three signals are signals obtained by detecting an output voltage or an input voltage of each phase.

  In a preferred embodiment of the present invention, the control system is designed using a robust control design.

  In a preferred embodiment of the present invention, the control system is designed using the H∞ loop shaping method.

  In a preferred embodiment of the present invention, based on the first or second harmonic compensation signal output from the harmonic compensation means, it is determined that the control for suppressing the harmonic component tends to diverge. And a divergence determining means for stopping the output of the first and second harmonic compensation signals when the divergence determining means determines that there is a divergence tendency.

  In a preferred embodiment of the present invention, based on the first or second harmonic compensation signal output from the harmonic compensation means, it is determined that the control for suppressing the harmonic component tends to diverge. A divergence determining means for performing, and a phase changing means for changing the phase of the first and second harmonic compensation signals to a phase in which the control does not diverge when it is determined by the divergence determining means that there is a tendency to diverge. I have.

  In a preferred embodiment of the present invention, the divergence determining means determines that the first or second harmonic compensation signal has a divergence tendency when the first or second harmonic compensation signal exceeds a predetermined threshold.

  A grid-connected inverter system provided by the fourth aspect of the present invention includes an inverter circuit and a control circuit provided by the first to third aspects of the present invention.

  A three-phase PWM converter system provided by the fifth aspect of the present invention includes a converter circuit and a control circuit provided by the first to third aspects of the present invention.

According to the present invention, the first harmonic signal and the second signal are signal-processed by the first transfer function G ₁ (s) and the phase adjustment process is performed, so that the first harmonic compensation signal and the second signal 2 harmonic compensation signals are generated. The signal processing by the first transfer function G ₁ (s) is the same processing as that in which the harmonic compensation signal generated by performing the predetermined control processing after performing the rotational coordinate transformation is subjected to the stationary coordinate transformation. Further, the signal processing by the first transfer function G ₁ (s) has linearity and time invariance. Therefore, a design method based on the linear control theory can be used, and the control system can be easily designed. System analysis can also be performed.

  Other features and advantages of the present invention will become more apparent from the detailed description given below with reference to the accompanying drawings.

It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process. It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process, and is represented by the matrix. It is a block diagram for demonstrating the calculation of a matrix. It is a block diagram which shows the process which performs static coordinate transformation after performing PI control after performing rotational coordinate transformation. It is a block diagram which shows the process which performs a stationary coordinate transformation after performing I control after performing a rotational coordinate transformation. Is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. It is a figure for demonstrating the signal for a positive phase, and the signal for a reverse phase. It is a block diagram for demonstrating the grid connection inverter system which concerns on 1st Embodiment. It is a figure for demonstrating the simulation result performed in 2nd Embodiment. It is a figure for demonstrating the experimental result performed in 2nd Embodiment. It is a table | surface for demonstrating the experimental result performed in 2nd Embodiment. It is a block diagram for demonstrating the control circuit which concerns on 3rd Embodiment. It is a block diagram for demonstrating the control circuit which concerns on 4th Embodiment. It is a Bode diagram which shows an example of the transfer function before and behind interconnection. It is a figure for demonstrating the harmonic compensation controller which concerns on 5th Embodiment. It is a figure for demonstrating the other Example of the harmonic compensation controller which concerns on 5th Embodiment. It is a block diagram for demonstrating the three-phase PWM converter system which concerns on 6th Embodiment. It is a block diagram for demonstrating the conventional general grid connection inverter system. It is a block diagram for demonstrating the internal structure of a control circuit. It is a block diagram for demonstrating the internal structure of a harmonic compensation controller.

  Embodiments of the present invention will be specifically described below with reference to the drawings.

  First, a method for converting processing involving rotational coordinate transformation and stationary coordinate transformation into linear time-invariant processing will be described.

  Fig.1 (a) is a figure for demonstrating the process accompanied by rotation coordinate transformation and stationary coordinate transformation. In this processing, first, the signals α and β are converted into signals d and q by rotational coordinate conversion. The signals d and q are each processed by a predetermined transfer function F (s), and signals d 'and q' are output. Next, the signals d ′ and q ′ are converted into signals α ′ and β ′ by stationary coordinate transformation. The nonlinear time-varying process shown in FIG. 1A is converted into a process using a matrix G of linear time-invariant transfer functions shown in FIG.

The rotational coordinate transformation shown in FIG. 1A is represented by a determinant of the following equation (7), and the stationary coordinate transformation is represented by a determinant of the following equation (8).

  Therefore, the process shown in FIG. 1A can be expressed as shown in FIG. 2A using a matrix. The matrix G shown in FIG. 1B can be calculated by calculating the product of the three matrices shown in FIG. 2A and making the calculated matrix a linear time-invariant matrix. At this time, the calculation is performed after converting the matrix of the stationary coordinate conversion and the rotation coordinate conversion into a matrix product.

The matrix for rotational coordinate conversion can be converted into a product of the right-hand side matrix shown in the following equation (9).
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

Also, the matrix of the static coordinate conversion can be converted into the product of the matrix on the right side shown in the following equation (10). The matrix in the middle of the matrix product is a linear time-invariant matrix.
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

When the product of the three matrices shown in FIG. 2A is calculated using the above equations (9) and (10), and the matrix G is calculated, the following equation (11) is calculated.

When attention is paid to the element in the first row and the first column of the three matrixes in the center of the above equation (11) and this is expressed in a block diagram, the block diagram shown in FIG. 3 is obtained. When calculating the input / output characteristics of the block diagram shown in FIG.
It becomes. Where F (s) is a one-input one-output transfer function having an impulse response f (t).

Here, when θ (t) = ω ₀ t, θ (t) −θ (τ) = ω ₀ t−ω ₀ τ = ω ₀ (t−τ) = θ (t−τ). The input / output characteristics of the block diagram shown in FIG. 3 are equal to those of a linear time invariant system having an impulse response f (t) exp (−jω ₀ t). When the impulse response f (t) exp (−jω ₀ t) is Laplace transformed, a transfer function F (s + jω ₀ ) is obtained. The input / output characteristics when exp (jθ (t)) and exp (−jθ (t)) in the block diagram shown in FIG. 3 are interchanged are the input / output characteristics of the transfer function F (s−jω ₀ ). become.

Therefore, if the calculation is further advanced from the above equation (11),
Is calculated.

  Thereby, the process shown in FIG. 2A can be converted into the process shown in FIG. The process shown in FIG. 2B is equivalent to the process of performing the stationary coordinate conversion after performing the process represented by the predetermined transfer function F (s) after performing the rotational coordinate conversion. This system is a linear time-invariant system.

The transfer function of the PI control (proportional integral control) controller is expressed as F (s) = K _P + K _I / s, where K _P and K _I are the proportional gain and integral gain, respectively. Therefore, the transfer function matrix G _PI indicating the process shown in FIG. 4, that is, the process equivalent to the process of performing the stationary coordinate conversion after performing the PI control after performing the rotational coordinate conversion, uses the above equation (12). Thus, the following equation (13) is calculated.

Further, the transfer function of the I control (integral control) controller, the integral gain and K _I, represented by _{F (s) = K I /} s. Therefore, the processing shown in FIG. 5, i.e., the matrix G _I of the transfer function from performing a rotational coordinate transformation showing the processing of the processing equivalent to a still coordinate transformation after the I control, using the equation (12) Is calculated as in the following equation (14).

Figure 6 is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. FIG (a) is first row and the first column elements (hereinafter, the same applies for. Other elements described as "(1,1) element".) The matrix G _I and (2,2) element of FIG. 4B shows the transfer function of the (1,2) element of the matrix G _I , and FIG. 4C shows the transfer function of the (2,1) element of the matrix G _I. Is shown. The figure shows the case where the frequency of the fundamental wave of the system voltage (hereinafter referred to as “center frequency”. Also, the angular frequency corresponding to the center frequency is referred to as “center angular frequency”) is 60 Hz (that is, angular frequency). ω ₀ = 120π), and the integral gain K _I is “0.1”, “1”, “10”, “100”.

FIG (a), the amplitude characteristic shown by (b) and (c) are all, there is a peak in the center frequency, the integral gain K _I is increased, the amplitude characteristic is increased. Further, the phase characteristic shown in FIG. 5A is 0 degree at the center frequency. In other words, the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without the signal of the center frequency (center angular frequency) changes the phase. The phase characteristic shown in FIG. 5B is 90 degrees at the center frequency. In other words, the transfer function of the (1,2) element of the matrix G _I causes the phase of the signal of the center frequency (center angular frequency) passes advancing 90 degrees. On the other hand, the phase characteristic shown in FIG. 4C is −90 degrees at the center frequency. In other words, the transfer function of (2,1) element of the matrix G _I is the phase of the signal of the center frequency (center angular frequency) passing delayed 90 degrees.

The matrix G (G _PI , G _I ) of the transfer function described above is for controlling the positive phase component of the fundamental wave component. Next, a method for controlling the reverse phase will be described.

  FIG. 7 is a diagram for explaining the signal for the positive phase and the signal for the negative phase of the fundamental wave. FIG. 4A shows the signal for the positive phase of the fundamental wave, and FIG. 4B shows the signal for the reverse phase of the fundamental wave.

In FIG. 5A, the positive phase signals of the fundamental wave components of the current signals Iu, Iv, Iw are indicated by the vectors u, v, w of broken line arrows. The vectors u, v, and w have directions different from each other by 120 degrees, and are arranged in a clockwise order and rotated counterclockwise at an angular frequency ω ₀ . The α-axis signal and β-axis signal obtained by converting the positive phase signal into three-phase / two-phase signals are indicated by vectors α and β of solid arrows. The vectors α and β are different in direction by 90 degrees in the clockwise order, and are rotated counterclockwise at the angular frequency ω ₀ .

That is, the α-axis signal is 90 degrees ahead of the β-axis signal. even if the processing shown in the transfer function of (1, 1) element of the matrix G _I to α-axis signal, the phase does not change (see FIG. 6 (a)). Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I to β-axis signal, it advances the phase by 90 degrees (see Figure 6 (b)). Therefore, since both phases become the same phase as the α-axis signal, they are strengthened by adding both. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I to α-axis signal, the phase is delayed 90 degrees (see Figure 6 (c)). Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in β axis signal, the phase does not change. Therefore, since both phases become the same phase as the β-axis signal, they are strengthened by adding both.

The reverse phase component is a component whose phase sequence is opposite to the normal phase component. In FIG. 7 (b), the antiphase component signals of the fundamental components of the current signals Iu, Iv, Iw are indicated by vectors u, v, w of broken line arrows. The vectors u, v, and w have directions different from each other by 120 degrees, and are arranged in the counterclockwise order and rotated in the counterclockwise direction at the angular frequency ω ₀ . An α-axis signal and a β-axis signal obtained by three-phase / two-phase conversion of the antiphase signal are indicated by vectors α and β of solid arrows. The vectors α and β are different in the direction of 90 degrees in the counterclockwise order, and rotate in the counterclockwise direction at the angular frequency ω ₀ .

That is, the α-axis signal is 90 degrees behind the β-axis signal. even if the processing shown in the transfer function of (1, 1) element of the matrix G _I to α-axis signal, the phase does not change. Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I to β-axis signal, it advances the phase by 90 degrees. Therefore, since the phases of both are reversed, adding them together cancels each other. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I to α-axis signal, the phase is delayed 90 degrees. Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in β axis signal, the phase does not change. Therefore, since the phases of both are reversed, adding them together cancels each other. Accordingly, the matrix G _I of the transfer function, performs control of the positive phase component, control of the reverse phase is not performed.

If interchanged and (1,2) element of the matrix G _I of the transfer function and (2,1) element, contrary to the above, so that cancel each other positive phase component, reverse-phase components constructive. Therefore, when controlling the reversed phase, the (1,2) of the matrix G _I of the transfer function element and the (2,1) element and matrix may be used with interchanged. The same applies to the transfer function matrices G and _GPI .

  Next, a method for controlling harmonic components will be described.

The matrix G of the transfer function shown in the above equation (12) is for controlling the fundamental wave component. The n-th harmonic is an angular frequency component obtained by multiplying the angular frequency of the fundamental wave by n. When three-phase / two-phase conversion is performed on the positive phase component of the n-th harmonic, there are cases where the phase of the α-axis signal is advanced or delayed. In the case of n = 3k + 1 (k = 1, 2,...), the order of the phase of the positive phase signal of the n-th harmonic matches the order of the phases of the positive phase signal of the fundamental wave. That is, if the U, V, and W phase signals for the positive phase of the fundamental wave are Vu = Vcos θ, Vv = Vcos (θ-2π / 3), and Vw = Vcos (θ-4π / 3), for example, 7 positive phase of U of harmonics, V, each signal of the W _{phase, Vu 7 = V 7 cos7θ,} Vv 7 = V 7 cos (7θ-14π / 3) = V 7 cos (7θ-2π / 3) , the _{Vw 7 = V 7 cos (7θ} -28π / 3) = V 7 cos (7θ-4π / 3). In this case, the phase order matches the phase order of the positive phase signal of the fundamental wave, and the phase of the α-axis signal advances 90 degrees from the phase of the β-axis signal, as in FIG. Therefore, the matrix of the transfer function when controlling the positive phase component of the nth harmonic (n = 3k + 1) is the transfer function shown in the following equation (15) where ω ₀ is n · ω ₀ in the above equation (12). A matrix G _n is obtained. On the other hand, in the case of n = 3k + 2 (k = 0, 1, 2,...), The order of the phases of the n-order harmonics of the positive phase signal coincides with the order of the phases of the negative-phase signal of the fundamental wave. That is, when the U, V, and W phase signals Vu, Vv, and Vw for the positive phase of the fundamental wave are as described above, for example, the U, V, and W phase signals for the positive phase of the fifth harmonic are respectively Vu ₅ = V ₅ cos 5θ, Vv ₅ = V ₅ cos (5θ-10π / 3) = V ₅ cos (5θ-4π / 3), Vw ₅ = V ₅ cos (5θ-20π / 3) = V ₅ cos ( 5θ-2π / 3). In this case, the phase order coincides with the phase order of the anti-phase signal of the fundamental wave, and the phase of the α-axis signal is delayed by 90 degrees from the phase of the β-axis signal, as in FIG. Therefore, the matrix of the transfer function in the case of controlling the positive phase component of the n-th harmonic (n = 3k + 2) is represented by the above equation (12) where ω ₀ is n · ω ₀ and the (1,2) element and (2, 1) The transfer function matrix G _n shown in the following equation (15 ′) with elements replaced is obtained.

The transfer function matrix G _In for controlling the positive phase of the nth-order harmonic (n = 3k + 1) is represented by the following equation (16), where ω ₀ is n · ω ₀ in the above equation (14). Is calculated as follows. On the other hand, the matrix G _In of the transfer function for performing I control of the positive phase component of the nth-order harmonic (n = 3k + 2) is represented by the above equation (14) where ω ₀ is n · ω ₀ and (1,2) elements And (2,1) element are interchanged, and calculation is performed as in the following equation (16 ′). The following formulas (16) and (16 ′) can also be calculated as F (s) = K _I / s in the above formulas (15) and (15 ′).

In the following, the case where the harmonic compensation controller that performs the processing represented by the transfer function matrix G _In of the above equations (16) and (16 ′) is applied to the control circuit of the grid-connected inverter system will be described. This will be described as the first embodiment.

  FIG. 8 is a block diagram for explaining the grid-connected inverter system according to the first embodiment.

  As shown in the figure, the grid-connected inverter system A includes a DC power source 1, an inverter circuit 2, a filter circuit 3, a transformer circuit 4, a current sensor 5, a voltage sensor 6, and a control circuit 7.

  The DC power source 1 is connected to the inverter circuit 2. The inverter circuit 2, the filter circuit 3, and the transformer circuit 4 are connected in series to the output lines of the U-phase, V-phase, and W-phase output voltages in this order, and connected to the three-phase AC power system B. Yes. The current sensor 5 and the voltage sensor 6 are installed on the output side of the transformer circuit 4. The control circuit 7 is connected to the inverter circuit 2. The grid interconnection inverter system A converts the DC power output from the DC power supply 1 into AC power and supplies it to the power grid B. In addition, the structure of the grid connection inverter system A is not restricted to this. For example, the current sensor 5 and the voltage sensor 6 may be provided on the input side of the transformer circuit 4, or other sensors necessary for controlling the inverter circuit 2 may be provided. Further, the transformer circuit 4 may be provided on the input side of the filter circuit 3, or a so-called transformer-less system in which the transformer circuit 4 is not provided. Further, a DC / DC converter circuit may be provided between the DC power supply 1 and the inverter circuit 2.

  The DC power source 1 outputs DC power and includes, for example, a solar battery. A solar cell generates direct-current power by converting solar energy into electrical energy. The DC power source 1 outputs the generated DC power to the inverter circuit 2. Note that the DC power source 1 is not limited to one that generates DC power from a solar cell. For example, the DC power source 1 may be a fuel cell, a storage battery, an electric double layer capacitor, a lithium ion battery, or AC power generated by a diesel engine generator, a micro gas turbine generator, a wind turbine generator, or the like. It may be a device that converts to DC power and outputs it.

  The inverter circuit 2 converts a DC voltage input from the DC power source 1 into an AC voltage and outputs the AC voltage to the filter circuit 3. The inverter circuit 2 is a three-phase inverter, and is a PWM control type inverter circuit including three sets of six switching elements (not shown). The inverter circuit 2 converts the DC voltage input from the DC power source 1 into an AC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. In addition, the inverter circuit 2 is not limited to this, For example, a multilevel inverter may be sufficient.

  The filter circuit 3 removes high frequency components due to switching from the AC voltage input from the inverter circuit 2. The filter circuit 3 includes a low pass filter including a reactor and a capacitor. The AC voltage from which the high frequency component has been removed by the filter circuit 3 is output to the transformer circuit 4. The configuration of the filter circuit 3 is not limited to this, and any known filter circuit for removing high frequency components may be used. The transformer circuit 4 boosts or lowers the AC voltage output from the filter circuit 3 to a level substantially the same as the system voltage.

  The current sensor 5 detects the alternating current of each phase output from the transformer circuit 4 (that is, the output current of the grid interconnection inverter system A). The detected current signal I (Iu, Iv, Iw) is input to the control circuit 7. The voltage sensor 6 detects the system voltage of each phase of the power system B. The detected voltage signal V (Vu, Vv, Vw) is input to the control circuit 7. Note that the output voltage output by the grid interconnection inverter system A substantially matches the grid voltage.

  The control circuit 7 controls the inverter circuit 2 and is realized by, for example, a microcomputer. The control circuit 7 generates a PWM signal based on the current signal I input from the current sensor 5 and the voltage signal V input from the voltage sensor 6 and outputs the PWM signal to the inverter circuit 2. The control circuit 7 generates a command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A based on the detection signal input from each sensor, and is generated based on the command value signal. Output a pulse signal as a PWM signal. The inverter circuit 2 outputs an alternating voltage having a waveform corresponding to the command value signal by switching each switching element on and off based on the input PWM signal. The control circuit 7 controls the output current by changing the waveform of the command value signal to change the waveform of the output voltage of the grid interconnection inverter system A. Thereby, the control circuit 7 performs various feedback controls. In addition, the control circuit 7 suppresses harmonics by causing the inverter circuit 2 to output harmonics for canceling the harmonics input from the power system B.

  In FIG. 8, only the structure for performing output current control and harmonic suppression control is described, and the structure for other controls is omitted. Actually, the control circuit 7 performs DC voltage control (feedback control performed so that the input DC voltage becomes a preset voltage target value) and reactive power control (reactive power target value with preset output reactive power) (Feedback control to be performed) is also performed. The control method performed by the control circuit 7 is not limited to this. For example, output voltage control or active power control may be performed.

  The control circuit 7 includes a system counter component generation unit 72, a three-phase / two-phase conversion unit 73, a current controller 74, a two-phase / three-phase conversion unit 76, a PWM signal generation unit 77, and a harmonic compensation controller 8. .

  The system counter-part generating unit 72 receives the voltage signal V from the voltage sensor 6 and generates and outputs system command value signals Ku, Kv, Kw. The system command value signals Ku, Kv, Kw serve as a reference for the command value signal for instructing the waveform of the output voltage output from the system interconnection inverter system A. The system command value signals Ku, Kv, Kw are described later. The command value signal is generated by correcting with the correction value signals Xu, Xv, and Xw.

  The three-phase / two-phase conversion unit 73 is the same as the three-phase / two-phase conversion unit 73 shown in FIG. 19, and the three current signals Iu, Iv, Iw input from the current sensor 5 are converted into α-axis current signals. It converts to Iα and β-axis current signal Iβ. The conversion process performed by the three-phase / two-phase conversion unit 73 is represented by the determinant shown in the above equation (1).

The current controller 74 receives the deviation between the α-axis current signal Iα and β-axis current signal Iβ output from the three-phase / two-phase converter 73 and the target value for the positive phase of each fundamental wave, and performs current control. For this purpose, fundamental wave compensation signals Xα and Xβ are generated. Current controller 74, shown in the equation (14), performs the processing expressed a positive phase of the fundamental wave in a matrix G _I of the transfer function for I control. That is, when the deviations between the α-axis current signal Iα and the β-axis current signal Iβ and the target values for the positive phase of the fundamental wave are ΔIα and ΔIβ, respectively, the processing shown in the following equation (17) is performed. The angular frequency omega ₀ is the fundamental angular frequency of the system voltage _{(e.g., ω 0 = 120π [rad /} sec] (60 [Hz])) is set in advance, the integral gain K _I is pre-designed. Further, the current controller 74 performs a process for maximizing the stability margin, and the phase adjustment for correcting the phase delay in the control loop is also performed.

  In the present embodiment, as the α-axis current target value and the β-axis current target value, those obtained by converting the d-axis current target value and the q-axis current target value into stationary coordinates are used. A correction value for DC voltage control (not shown) is used for the d-axis current target value, and a correction value for reactive power control (not shown) is used for the q-axis current target value. When a three-phase current target value is given, the target value may be converted into three-phase / two-phase to obtain an α-axis current target value and a β-axis current target value. Further, the respective deviations between the three current signals Iu, Iv, Iw and the three-phase current target value are calculated in advance, and the three deviation signals are three-phase / two-phase converted and input to the current controller 74. You may do it. When the α-axis current target value and the β-axis current target value are directly given, the target values may be used as they are.

In the present embodiment, the current controller 74, using the matrix G _I of the transfer function in the frequency weighting, designed by H∞ loop shaping method, which is one of the linear control theory. Processing performed by the current controller 74, as demonstrated by the matrix G _I of the transfer function is a process of linear time invariant. Therefore, a control system design using linear control theory can be performed.

The current controller 74 is required to have a design specification that the output current follows the sine wave target value and that the output is returned to a predetermined ratio within a predetermined time (speed response) at the time of a sag. In order for the output of the system to completely follow a certain target value, the closed-loop system must have the same pole as the target generator, and the closed-loop system must be asymptotically stable (internal model principle). Pole of the sine-wave target value is ± jωo, terms poles of 1 / included in the transfer function of each element of the matrix ^{_{G I (s 2 + ω 0}} 2) is also ± jωo. Therefore, the poles of the closed loop system and the target generator are the same. In addition, if the H∞ loop shaping method is used, a controller in which the closed loop system becomes asymptotically stable can be designed. Therefore, by designing using the H∞ loop shaping method so as to satisfy the condition of rapid response, it is possible to easily design the most stable control system that meets the design specifications.

  Note that the design method used for designing the control system is not limited to this, and other linear control theories can also be used. For example, the design may be performed using a loop shaping method, optimal control, H∞ control, a mixed sensitivity problem, or the like.

  Returning to FIG. 8, the harmonic compensation controller 8 receives the α-axis current signal Iα and the β-axis current signal Iβ output from the three-phase / two-phase conversion unit 73, and performs harmonic compensation for harmonic suppression control. The signals Yα and Yβ are generated. The harmonic compensation controller 8 suppresses the fifth harmonic compensation unit 81 for suppressing the fifth harmonic, the seventh harmonic compensation unit 82 for suppressing the seventh harmonic, and the eleventh harmonic. The 11th harmonic compensation part 83 for this is provided.

The fifth harmonic compensation unit 81 is for suppressing the positive phase component of the fifth harmonic. The fifth-order harmonic compensator 81 is represented in the transfer function matrix G _I5 for controlling the positive phase component of the fifth-order harmonic with n = 5 in the transfer function matrix G _In of the above equation (16 ′). Perform the process. That is, the fifth harmonic compensator 81 performs the process shown below (18), the fifth harmonic compensation signal Yarufa _5, and outputs the Ybeta _5. As the angular frequency ω _{0, the} angular frequency of the fundamental wave of the system voltage (for example, ω ₀ = 120π [rad / sec] (60 [Hz])) is set in advance, and the integral gain K _I5 is designed in advance. The fifth-order harmonic compensator 81 also performs processing for maximizing the stability margin. Among these, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken.

In the present embodiment, the fifth-order harmonic compensator 81 is designed by the H∞ loop shaping method, which is one of linear control theory, using a transfer function matrix _GI5 as frequency weights. The processing performed in the fifth-order harmonic compensator 81 is a linear time-invariant processing because it is represented by a transfer function matrix _GI5 . Therefore, a control system design using linear control theory can be performed.

Note that the design method used for designing the control system is not limited to this, and other linear control theories can also be used. For example, the design may be performed using a loop shaping method, optimal control, H∞ control, a mixed sensitivity problem, or the like. Alternatively, the phase θ ₅ for adjustment from the phase delay may be calculated and set in advance. For example, if the phase is delayed by 90 degrees in the controlled object, θ ₅ = −90 degrees may be set in order to delay the phase by 180 degrees. In this case, a rotation transformation matrix based on the phase θ ₅ is added to the above equation (18).

The seventh harmonic compensation unit 82 is for suppressing the positive phase component of the seventh harmonic. The seventh-order harmonic compensator 82 is represented in the transfer function matrix G _I7 for controlling the positive phase component of the seventh-order harmonic with n = 7 in the transfer function matrix G _In of the above equation (16). Process. That is, the seventh harmonic compensation unit 82 performs processing shown in the following equation (19), and outputs _seventh harmonic compensation signals Yα ₇ and Yβ ₇ . As the angular frequency ω _0, the angular frequency of the fundamental wave of the system voltage is set in advance, and the integral gain K _I7 is designed in advance. The seventh harmonic compensation unit 82 also performs processing for maximizing the stability margin, and in this process, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken. The seventh harmonic compensation unit 82 is also designed in the same manner as the fifth harmonic compensation unit 81.

The 11th harmonic compensator 83 is for suppressing the positive phase component of the 11th harmonic. The eleventh-order harmonic compensator 83 is represented in the transfer function matrix G _I11 for controlling the positive phase of the eleventh-order harmonic with n = 11 in the transfer function matrix G _In of the above equation (16 ′). Perform the process. That is, 11 harmonic compensator 83 performs the process shown below (20), 11-order harmonic compensation signal Yarufa _11, and outputs the Ybeta _11. As the angular frequency ω _0, the angular frequency of the fundamental wave of the system voltage is set in advance, and the integral gain K _I11 is designed in advance. The eleventh harmonic compensation unit 83 also performs processing for maximizing the stability margin, and in this process, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken. The eleventh harmonic compensation unit 83 is also designed in the same manner as the fifth harmonic compensation unit 81.

5 harmonic compensation unit 81 outputs the fifth harmonic compensation signal Yα _5, Yβ _5, 7-order 7 harmonic compensator 82 has output harmonic compensation signal Yα _7, Yβ _7, and, 11 harmonics The 11th-order harmonic compensation signals Yα ₁₁ and Yβ ₁₁ output from the compensation unit 83 are added and output from the harmonic compensation controller 8 as harmonic compensation signals Yα and Yβ. In the present embodiment, the case where the harmonic compensation controller 8 includes the fifth-order harmonic compensation unit 81, the seventh-order harmonic compensation unit 82, and the eleventh-order harmonic compensation unit 83 has been described. Not limited. The harmonic compensation controller 8 may be designed according to the harmonic order that needs to be suppressed. For example, when it is desired to suppress only the fifth harmonic, only the fifth harmonic compensator 81 needs to be provided. Further, when it is desired to further suppress the 13th harmonic, the 13th harmonic is subjected to the processing represented by the transfer function matrix G _I13 where n = 13 in the transfer function matrix G _In of the above equation (16). What is necessary is just to further provide a compensation part.

  Returning to FIG. 8, the two-phase / three-phase converter 76 is the same as the two-phase / three-phase converter 76 shown in FIG. 19, and the correction value signals X′α and X′β are converted into three correction values. The signals are converted into signals Xu, Xv, and Xw. The conversion process performed by the two-phase / three-phase conversion unit 76 is expressed by the determinant shown in the above equation (4). The correction value signals X′α and X′β are obtained by adding the harmonic compensation signals Yα and Yβ output from the harmonic compensation controller 8 to the fundamental wave compensation signals Xα and Xβ output from the current controller 74.

  The system command value signals Ku, Kv, Kw output from the system counter-part generating unit 72 and the correction value signals Xu, Xv, Xw output from the two-phase / three-phase conversion unit 76 are added, respectively, and the command value signal X 'u, X'v, X'w are calculated and input to the PWM signal generation unit 77.

  The PWM signal generation unit 77 performs a triangular wave comparison based on the input command value signals X′u, X′v, and X′w and a carrier signal generated as a triangular wave signal having a predetermined frequency (for example, 4 kHz). PWM signals Pu, Pv, Pw are generated by the method. In the triangular wave comparison method, the command value signals X′u, X′v, and X′w are respectively compared with the carrier signal. For example, when the command value signal X′u is larger than the carrier signal, the high level is obtained. A low-level pulse signal is generated as the PWM signal Pu. The generated PWM signals Pu, Pv, Pw are output to the inverter circuit 2.

In the present embodiment, the fifth-order harmonic compensator 81 performs control in the stationary coordinate system without performing rotational coordinate conversion and stationary coordinate conversion. The transfer function matrix _GI5 is a transfer function matrix indicating a process equivalent to the process of performing the stationary coordinate conversion after performing the I control after performing the rotational coordinate conversion. Thus, the fifth harmonic compensator 81 which performs the processing expressed by the matrix G _I5 of the transfer function, the rotation coordinate conversion unit 811 shown in FIG. 20, the stationary coordinate transformation unit 816, and the I controller 814 and 815 and the equivalent Processing is in progress. Further, the Bode diagram of each element of the matrix _GI5 is the same as the Bode diagram shown in FIG. 6 with the center frequency set to a frequency five times the fundamental frequency. That is, the fifth-order harmonic compensator 81 has a high gain only for the frequency component five times the fundamental frequency. Therefore, it is not necessary to provide the LPFs 812 and 813 shown in FIG. That is, the fifth-order harmonic compensation unit 81 performs processing equivalent to the fifth-order harmonic compensation unit 810 shown in FIG.

The processing performed by the fifth harmonic compensation unit 81 is a linear time-invariant processing because it is represented by a transfer function matrix _GI5 . Further, the control loop for the fifth harmonic compensation does not include the rotation coordinate conversion process and the stationary coordinate conversion process, which are nonlinear time-varying processes, and thus is a linear time-invariant system. Therefore, control system design and system analysis using linear control theory are possible. As described above, by using the transfer function matrix G _I5 shown in the above equation (18), nonlinear processing for performing stationary coordinate transformation after performing I control after performing rotational coordinate transformation can be performed in a linear time invariant manner. It can be reduced to the input / output system, which facilitates system analysis and control system design.

  The same applies to the seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83. The seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83 include the seventh harmonic compensation unit 820 and the seventh harmonic compensation unit 820 shown in FIG. Equivalent processing with the 11th-order harmonic compensation unit 830 is performed. In addition, the processing performed by the seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83 is also a linear time-invariant processing, which enables control system design and system analysis using linear control theory.

In the above embodiment, the case where the integral gain of each element of the matrix of the transfer function is the same has been described, but a different value may be used for each element. For example, it can be designed to provide additional characteristics such as improving the quick response of the α-axis component and increasing the stability. It is also possible to provide additional properties that control both the (1,2) element and the integral gain K _I of (2,1) element designed to "0", normal phase content, reverse phase. The case of controlling both the positive phase portion and the reverse phase portion will be described later.

  Moreover, although the said embodiment demonstrated the case where the 5th harmonic compensation part 81, the 7th harmonic compensation part 82, and the 11th harmonic compensation part 83 were each designed individually, it is not restricted to this. The fifth-order harmonic compensator 81, the seventh-order harmonic compensator 82, and the eleventh-order harmonic compensator 83 may be designed at a time so that the integral gain is shared.

In the said embodiment, although the case where the positive phase part of each harmonic was controlled was demonstrated, it is not restricted to this. You may make it control the antiphase part of each harmonic. In this case, a matrix obtained by exchanging the (1, 2) element and the (2, 1) element of the transfer function matrix G _In used for controlling the positive phase component may be used. Moreover, you may make it perform control of both the part for a normal phase and a reverse phase. Hereinafter, a case where both the normal phase and the reverse phase are controlled will be described as a second embodiment.

Matrix processing shown in the transfer function of (1, 1) element and (2,2) element of G _I is passed through without changing the positive phase component and negative phase of the phase (see FIG. 6 (a)). Therefore, performing the use of the matrix (1,2) element and (2,1) element of the matrix G _I shown in the equation (14) to "0", the positive phase component, control of both the reverse phase be able to. The same applies to the matrices G _n and G _{In. In} the above equations (15) and (16), if a matrix in which the (1,2) element and the (2,1) element are set to “0” is used, Both of the reverse phase can be controlled.

The control circuit according to the second embodiment includes transfer functions used in the fifth harmonic compensation unit 81, the seventh harmonic compensation unit 82, and the eleventh harmonic compensation unit 83 in the control circuit 7 shown in FIG. The (1,2) and (2,1) elements of the matrices G _I5 , G _I7 , G _I11 are set to “0”. In the second embodiment, it is possible to control both the positive phase component and the reverse phase component of the harmonics of each order. Also in the second embodiment, as in the first embodiment, there is an effect that control system design and system analysis using linear control theory can be performed.

  FIG. 9 is a diagram for explaining a simulation result performed in the second embodiment.

  A simulation was performed in the case where the target current was set to 20 [A] by adding unbalanced disturbance and harmonic disturbance to the current of each phase of the grid-connected inverter system A (see FIG. 8). FIG. 9 shows waveforms of current signals Iu, Iv, and Iw obtained by detecting the output current of each phase by the current sensor 5. FIG. 6A shows the state immediately after the start of the simulation, and FIG. 10B shows the state after 50 seconds from the start of the simulation. As shown in FIG. 5B, each waveform of the current signals Iu, Iv, and Iw is a smooth waveform in which each harmonic component is suppressed.

  10 and 11 are diagrams for explaining the results of the experiment performed in the second embodiment.

  When the harmonic compensation controller 8 is provided in the grid interconnection inverter system A (see FIG. 8) connected to the electric power system B including the unbalanced disturbance and each harmonic disturbance (fifth harmonic compensation units 81 and 7). Experiments were performed using only the second harmonic compensation unit 82) and the case where the second harmonic compensation unit 82 was not provided. FIG. 10 shows the waveform of the U-phase current signal Iu when the steady state is reached. FIG. 4A shows the case where the harmonic compensation controller 8 is not provided, and FIG. 4B shows the case where the harmonic compensation controller 8 is provided. Compared to FIG. 6A, the waveform of FIG. 5B is a smoother waveform in which the fifth and seventh harmonic components are suppressed. FIG. 11 is a table showing the ratio of each harmonic component included in the current signals Iu, Iv, and Iw when the steady state is reached, and the ratio of each harmonic component when the ratio of the fundamental wave component is 100. Is shown. FIG. 4A shows the case where the harmonic compensation controller 8 is not provided, and FIG. 4B shows the case where the harmonic compensation controller 8 is provided. Compared with the table of FIG. 10A, the table of FIG. 10B suppresses the fifth-order and seventh-order harmonic components.

  In the first and second embodiments, the case where the three current signals Iu, Iv, and Iw are converted into the α-axis current signal Iα and the β-axis current signal Iβ is described, but the present invention is not limited to this. For example, direct control may be performed using three current signals Iu, Iv, and Iw. Hereinafter, an embodiment in this case will be described as a third embodiment.

  FIG. 12 is a block diagram for explaining a control circuit according to the third embodiment. In this figure, the same or similar elements as those in the control circuit 7 shown in FIG.

  The control circuit 7 ′ illustrated in FIG. 12 does not include the three-phase / two-phase conversion unit 73 and the two-phase / three-phase conversion unit 76, and includes a current controller 74 ′, a fifth harmonic compensation unit 81 ′, and a seventh harmonic. The control circuit 7 according to the first embodiment (see FIG. 8) is different in that the wave compensation unit 82 ′ and the 11th harmonic compensation unit 83 ′ directly control using the three current signals Iu, Iv, Iw. Different.

Since the three-phase / two-phase conversion and the two-phase / three-phase conversion are expressed by the above equations (1) and (4), the three-phase / two-phase conversion is performed and then the transfer function matrix G is expressed. The process of performing the two-phase / three-phase conversion after the process is represented by a transfer function matrix G ′ shown in the following equation (21).

Therefore, the transfer function matrix G ′ _I representing the processing performed by the current controller 74 ′ is expressed by the following equation (22).

The current controller 74 ′ receives deviations between the three current signals Iu, Iv, Iw output from the current sensor 5 and the target values for the positive phase of each fundamental wave, and a fundamental wave compensation signal for current control. X ″ u, X ″ v, X ″ w are generated. The current controller 74 ′ performs processing represented by the transfer function matrix G ′ _I in the above equation (22), that is, the current signal Iu. , Iv, Iw and the deviation of each ΔIu between the target value of the positive phase of each of the fundamental wave,? Iv, When? Iw, the following is performed the processing shown in equation (23). angular frequency omega ₀ is the system voltage The angular frequency of the fundamental wave (for example, ω ₀ = 120π [rad / sec] (60 [Hz])) is set in advance, and the integral gain K _I is designed in advance. Processing to maximize the stability margin is performed. In, it has been made the phase adjustment for correcting the delay of the phase of the control loop.

  In the present embodiment, the target value for the positive phase of the fundamental wave of the current signals Iu, Iv, and Iw is further subjected to two-phase / three-phase conversion by converting the d-axis current target value and the q-axis current target value to stationary coordinates. Things are used. When a three-phase current target value is directly given, the target value may be used as it is. In addition, when the α-axis current target value and the β-axis current target value are given, those obtained by two-phase / three-phase conversion may be used.

The process of performing the two-phase / three-phase conversion after performing the process represented by the transfer function matrix G _n after performing the three-phase / two-phase conversion is the transfer function matrix G shown in the following equation (24). ' _Represented by _n .

Therefore, the process of performing the two-phase / three-phase conversion after performing the process represented by the transfer function matrix G _In after performing the three-phase / two-phase conversion is the transfer function matrix G shown in the following equation (25). ' _{In In} .

The fifth harmonic compensation unit 81 ′ receives the three current signals Iu, Iv, Iw output from the current sensor 5, and a fifth harmonic compensation signal Yu for suppressing the positive phase component of the fifth harmonic. ₅ , Yv ₅ and Yw ₅ are generated, and the processing shown in the following equation (26) is performed. The transfer function matrix G ′ _I5 is set to n = 5 in the transfer function matrix G ′ _In of the above equation (25). The fifth-order harmonic compensator 81 ′ performs a process for maximizing the stability margin. Among these, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. Has been done.

In the present embodiment, the fifth-order harmonic compensator 81 ′ is designed by the H∞ loop shaping method, which is one of the linear control theories, using a transfer function matrix G ′ _I5 as the frequency weight. The processing performed by the fifth-order harmonic compensation unit 81 ′ is a linear time-invariant processing because it is represented by a transfer function matrix G ′ _I5 . Therefore, a control system design using linear control theory can be performed. The fifth-order harmonic compensator 81 ′ is designed in the same manner as the fifth-order harmonic compensator 81 according to the first embodiment. In addition, you may design using the other linear control theory.

The seventh harmonic compensation unit 82 'receives the three current signals Iu, Iv, and Iw output from the current sensor 5, and the seventh harmonic compensation signal Yu for suppressing the positive phase component of the seventh harmonic. ₇ , Yv ₇ and Yw ₇ are generated, and the processing shown in the following equation (27) is performed. The transfer function matrix G ′ _I7 is set to n = 7 in the transfer function matrix G ′ _In of the above equation (25). In addition, the seventh harmonic compensation unit 82 ′ performs a process for maximizing the stability margin, and among these, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. Has been done. The seventh harmonic compensation unit 82 ′ is also designed in the same manner as the fifth harmonic compensation unit 81 ′.

The eleventh harmonic compensation unit 83 ′ receives the three current signals Iu, Iv, and Iw output from the current sensor 5, and the eleventh harmonic compensation signal Yu for suppressing the positive phase component of the eleventh harmonic. ₁₁ , Yv ₁₁ , Yw ₁₁ are generated, and the processing shown in the following equation (28) is performed. The transfer function matrix G ′ _I11 is set to n = 11 in the transfer function matrix G ′ _In of the above equation (25). In addition, the 11th-order harmonic compensator 83 ′ performs a process for maximizing the stability margin, and among these, the phase adjustment for correcting the phase delay in the control loop to make it an opposite phase is also performed. Has been done. The eleventh harmonic compensator 83 ′ is also designed in the same manner as the fifth harmonic compensator 81 ′.

The fifth harmonic compensation signals Yu ₅ , Yv ₅ , Yw ₅ output from the fifth harmonic compensation unit 81 ′, and the seventh harmonic compensation signals Yu ₇ , Yv ₇ , Yw output from the seventh harmonic compensation unit 82 ′. ₇ and 11th harmonic compensation signals Yu ₁₁ , Yv ₁₁ , Yw ₁₁ output from the 11th harmonic compensation unit 83 ′ are added, and the harmonic compensation controller 8 is used as the harmonic compensation signals Yu, Yv, Yw. Are added to the fundamental wave compensation signals X ″ u, X ″ v, and X ″ w output from the current controller 74 ′ to become correction value signals Xu, Xv, and Xw. 8 ′ may be designed according to the order of the harmonics that need to be suppressed, for example, if only the 5th harmonics are to be suppressed, only the 5th harmonic compensation unit 81 ′ may be provided. If you want to suppress the 13th harmonic, 25) 'in the _In, the matrix G of the transfer function which is the n = 13' matrix G of the transfer function of the equation may be so further comprises a 13-order harmonic compensation unit for performing a process represented in _I13.

In the present embodiment, the fifth-order harmonic compensator 81 ′ that performs the processing represented by the transfer function matrix G ′ _I5 includes the three-phase / two-phase converter 73 and the two-phase / three-phase converter 76 shown in FIG. , A process equivalent to the rotation coordinate conversion unit 811, the stationary coordinate conversion unit 816, and the I control units 814 and 815 shown in FIG. 20 is performed. Further, since the fifth-order harmonic compensator 81 ′ has a high gain only for the frequency component that is five times the fundamental frequency, it is not necessary to provide the LPFs 812 and 813 shown in FIG. The processing performed by the fifth-order harmonic compensation unit 81 ′ is a linear time-invariant processing because it is represented by a transfer function matrix G ′ _I5 . Further, the control loop for the fifth harmonic compensation does not include the rotation coordinate conversion process and the stationary coordinate conversion process, which are nonlinear time-varying processes, and thus is a linear time-invariant system. Therefore, control system design and system analysis using linear control theory are possible. The same applies to the seventh harmonic compensation unit 82 'and the eleventh harmonic compensation unit 83'.

In the third embodiment, when controlling the _antiphase component of each harmonic, G _In12 (s), G _In23 among the elements of the matrix G ′ _In of the transfer function used for controlling the positive phase component. A matrix obtained by replacing (s) and G _In31 (s) with G _In13 (s), G _In21 (s), and G _In32 (s) (that is, a transposed matrix of the matrix G ′ _In ) may be used. Moreover, you may make it perform control of both the part for a normal phase and a reverse phase. Hereinafter, in the third embodiment, the fifth harmonic compensator 81 ′, the seventh harmonic compensator 82 ′, and the eleventh harmonic compensator 83 ′ control both the positive phase component and the negative phase component. The case where it performs is demonstrated.

Considering the case _where the (1,2) and (2,1) elements of the matrix G _n are set to “0” in the equation (24), the transfer function matrix G ″ _n shown in the following equation (29) Can be calculated.

Accordingly, when the fifth-order harmonic compensator 81 ′ controls both the positive phase component and the negative phase component, the matrix G ″ with n = 5 in the matrix G ″ _In of the transfer function of the following equation (30). _I5 should be used. Similarly, when the seventh-order harmonic compensator 82 ′ controls both the positive phase component and the negative phase component, the matrix G ″ _In of the following equation (30) is set to a matrix G with n = 7. " _I7 may be used, and when the 11th harmonic compensator 83 'controls both the positive phase component and the negative phase component, in the matrix G" _In of the transfer function of the following equation (30), A matrix G ′ _I11 with n = 11 may be used.

In the first to third embodiments, the fifth-order harmonic compensator 81 (81 ′), the seventh-order harmonic compensator 82 (82 ′), and the eleventh-order harmonic compensator 83 (83 ′) are controlled by I. Although the case of performing control in place of is described, the present invention is not limited to this. For example, control in place of PI control may be performed. In the first embodiment, when the fifth harmonic compensator 81, the seventh harmonic compensator 82, and the eleventh harmonic compensator 83 perform control instead of PI control, n = 3k + 1 (k = In the case of (1, 2,...), Ω ₀ is n · ω ₀ in the above equation (13), and a transfer function matrix G _PIn calculated as in the following equation (31) may be used. When n = 3k + 2 (k = 0, 1, 2,...), Ω _{0 is set} to n · ω ₀ in the above equation (13), and the (1, 2) element and the (2, 1) element are interchanged. Then, a transfer function matrix G _PIn calculated as shown in the following equation (31 ′) may be used. The following formulas (31) and (31 ′) can also be calculated as F (s) = K _P + K _I / s in the above formulas (15) and (15 ′).

When the fifth harmonic compensator 81 performs control in place of PI control, the matrix G _{PI5 in} which n = 5 in the above equation (31 ′) may be used, and the seventh harmonic compensator 82 performs PI control. When performing control _instead of PI, the matrix G _PI7 with n = 7 in the above equation (31) may be used. When the 11th harmonic compensator 83 performs control _instead of PI control, The matrix G _PI11 with n = 11 in the equation (31 ′) may be used.

In the second embodiment, when the fifth-order harmonic compensator 81, the seventh-order harmonic compensator 82, and the eleventh-order harmonic compensator 83 perform control in place of PI control, the above equation (31) and A matrix in _which the (1,2) element and the (2,1) element of the matrix G _PIn of the transfer function shown in the equation (31 ′) are “0” may be used.

In the third embodiment, when the fifth harmonic compensator 81 ′, the seventh harmonic compensator 82 ′, and the eleventh harmonic compensator 83 ′ perform control in place of PI control, the following (32 The transfer function matrix G ′ _PIn shown in the equation may be used.

In the case where the anti-phase component of each harmonic is controlled in the third embodiment, the fifth-order harmonic compensator 81 ′, the seventh-order harmonic compensator 82 ′, and the eleventh-harmonic compensator 83 ′ replace PI control. In the case of performing control, _among the elements of the transfer function matrix G ′ _PIn shown in the above equation (32), G _PIn12 (s), G _PIn23 (s), G _PIn31 (s), and G _PIn13 ( s), G _PIn21 (s) and G _PIn32 (s) may be replaced by a matrix (that is, a transposed matrix of the matrix G ′ _PIn ). In the third embodiment, when both the positive phase component and the reverse phase component are controlled, the fifth harmonic compensation unit 81 ′, the seventh harmonic compensation unit 82 ′, and the eleventh harmonic compensation unit 83 ′. When performing the control _instead of the PI control, the transfer function matrix G ″ _PIn shown in the following equation (33) may be used.

When performing control in place of PI control, there is a merit that a damping effect at the time of transition can be added by adjusting the proportional gain K _P , but there is a demerit that it is easily affected by modeling errors. Conversely, when performing control in place of I control, there is a demerit that a damping effect during transition cannot be added, but there is a merit that it is less susceptible to modeling errors.

The fifth harmonic compensation unit 81 (81 ′), the seventh harmonic compensation unit 82 (82 ′), and the eleventh harmonic compensation unit 83 (83 ′) are controlled in place of controls other than the I control and the PI control. May be performed. In the above equation (15), by using the transfer function F (s) as the transfer function of each control, a transfer showing a process equivalent to the process of performing the stationary coordinate conversion after performing the control after performing the rotation coordinate conversion. A matrix of functions can be calculated. Therefore, PID control (the transfer function is expressed as F (s) = K _P + K _I / s + K _D · s where K _P is a proportional gain, K _{I is} an integral gain, and K _D is a differential gain). it can be made to perform control to replace, D control (differential control:. transfer function, the differential gain and K _D, represented by _{F (s) = K D ·} s), P control (proportional Control: The transfer function is expressed by F (s) = K _P, where the proportional gain is K _P ), and can be controlled in place of PD control, ID control, and the like.

  In the first to third embodiments, the case where the output current is controlled has been described. However, the present invention is not limited to this. For example, the output voltage may be controlled. The case where the output voltage is controlled will be described below as a fourth embodiment.

  FIG. 13 is a block diagram for explaining a control circuit according to the fourth embodiment. In the figure, the same or similar elements as those in the grid interconnection inverter system A shown in FIG.

  An inverter system A ′ shown in FIG. 13 is different from the grid-connected inverter system A according to the first embodiment (see FIG. 8) in that power is supplied to the load L instead of the power system B. Since it is necessary to control the voltage supplied to the load L, the control circuit 9 controls the output voltage, not the output current. The control circuit 9 is different from the control circuit 7 according to the first embodiment (see FIG. 8) in that a PWM signal is generated based on the voltage signal V input from the voltage sensor 6. The inverter system A ′ supplies power to the load L while controlling the output voltage to a target value by feedback control.

The three-phase / two-phase converter 93 converts the three voltage signals Vu, Vv, Vw input from the voltage sensor 6 into an α-axis voltage signal Vα and a β-axis voltage signal Vβ. The conversion process performed by the three-phase / two-phase conversion unit 93 is represented by a determinant represented by the following equation (34).
The voltage signals Vu, Vv, and Vw are phase voltage signals for each phase, but a line voltage signal may be detected and used. In this case, after converting the line voltage signal into the phase voltage signal, the determinant shown in the above equation (34) is used, or instead of the matrix shown in the above equation (34), the line voltage signal is converted to the α axis. The matrix may be converted into the voltage signal Vα and the β-axis voltage signal Vβ.

The voltage controller 94 receives the deviation between the α-axis voltage signal Vα and β-axis voltage signal Vβ output from the three-phase / two-phase converter 93 and the target value of the positive phase of each fundamental wave, and performs voltage control. For this purpose, fundamental wave compensation signals Xα and Xβ are generated. Voltage controller 94 performs processing represented by the matrix G _I of the transfer function of the equation (14).

Also in this embodiment, the harmonic compensation controller 8 is provided. The fifth-order harmonic compensator 81 receives the α-axis voltage signal Vα and the β-axis voltage signal Vβ output from the three-phase / two-phase converter 93, and the process represented by the transfer function matrix G _I5 ; performs processing for adjusting the phase to the harmonics to be output to the opposite phase, the fifth harmonic compensation signal Yarufa _5, and outputs the Ybeta _5. Similarly, the seventh-order harmonic compensator 82 performs the process represented by the transfer function matrix _GI7 and the process of adjusting the phase so that the output harmonics have the opposite phase, and the seventh-order harmonics. compensation signal Yarufa _7, outputs Yβ _7, 11 harmonic compensator 83, the process represented in the matrix G _I11 of the transfer function, and adjusts the phase to the harmonics to be output to the opposite phase processing the go, 11-order harmonic compensation signal Yarufa _11, and outputs the Ybeta _11.

  In this embodiment, the same effect as that of the first embodiment can be obtained. Even when the output voltage is controlled, the control methods described in the first to third embodiments can be used. For example, control for both the positive phase and the reverse phase may be performed, control may be performed using three voltage signals Vu, Vv, and Vw directly, or control in place of PI control may be performed. You may make it perform. Further, the control of the output voltage and the control of the output current may be switched by combining the first embodiment and the fourth embodiment. In other words, current control is performed to supply power to the power system B when connected to the power system B, and voltage control is performed to supply power to the load L when not connected to the power system B. May be.

  Next, a method for preventing the harmonic suppression control from diverging will be described.

  In the first embodiment (see FIG. 8), the fifth-order harmonic compensator 81 adjusts the phase to correct the phase delay in the control loop so as to make the phase opposite. The phase to be adjusted is set based on the impedance of the grid-connected inverter system A before being linked to the power grid B (mainly due to the inductance of the reactor of the filter circuit 3 and the capacitance of the capacitor). When the grid-connected inverter system A is linked to the power grid B, if the load condition of the power grid B is different from the assumption, the resonance point of the filter circuit 3 may be shifted or increased. In this case, the phase to be adjusted is not appropriate, and control may diverge.

  FIG. 14 shows a Bode diagram of a transfer function from the output voltage of the inverter circuit 2 to the output current of the grid-connected inverter system A, and transmission before the grid-connected inverter system A is linked to the power system B. An example of the transfer function after being linked to the function is shown.

When the angular frequency ω ₀ of the fundamental wave of the system voltage is 120π [rad / sec] (60 [Hz]), the angular frequency of the fifth harmonic is 600π (≈1885) [rad / sec] (300 [Hz]). become. According to the figure, the phase of the fifth harmonic is delayed about 90 degrees before the interconnection, but is delayed about 270 degrees after the interconnection. That is, even if the phase is adjusted so that the control is the negative feedback control before the connection, the control becomes the positive feedback control after the connection and the control is diverged. Below, the case where the structure for preventing the divergence of such control is provided is demonstrated as 5th Embodiment.

  FIG. 15 is a diagram for explaining a harmonic compensation controller according to the fifth embodiment, and shows a fifth-order harmonic compensation unit 81 and a divergence prevention unit provided in the subsequent stage.

The divergence prevention unit 84 is for preventing divergence in the suppression control of the fifth harmonic. When the divergence prevention unit 84 determines that the control tends to diverge, the divergence prevention unit 84 changes the phases of the _fifth harmonic compensation signals Yα ₅ and Yβ ₅ input from the fifth harmonic compensation unit 81 and outputs the changed signals. The divergence prevention unit 84 includes a divergence determination unit 841 and a phase change unit 842.

The divergence determination unit 841 determines that the control tends to diverge. Divergence determination unit 841, the fifth harmonic compensation signal is input from the fifth harmonic compensator 81 Yα ₅ (or, the fifth harmonic compensation signal Ybeta ₅₎ a is compared with a predetermined threshold value, the fifth harmonic When the compensation signal Yα ₅ is larger than the predetermined threshold, it is determined that the control tends to diverge. The divergence determining unit 841 outputs a determination signal to the phase changing unit 842 when it is determined that the control tends to diverge. Note that the method of determining the divergence tendency by the divergence determining unit 841 is not limited to this. For example, when the fifth harmonic compensation signal Yα ₅ is larger than a predetermined threshold for a predetermined time, it may be determined that the control tends to diverge. Alternatively, the maximum value of the fifth-order harmonic compensation signal Yα ₅ may be always stored, and it may be determined that there is a divergence tendency based on the gradient of the maximum value.

Phase changer 842, the fifth harmonic compensation signal Yarufa ₅ inputted from the fifth harmonic compensator 81, and outputs by changing the phase of Ybeta _5. The phase changing unit 842 performs processing shown in the following equation (35) and outputs _fifth harmonic compensation signals Y′α ₅ and Y′β ₅ .

The [Delta] [theta] ₅ is set with an initial value "0", from a divergent determination unit 841 until the determination signal is input, the fifth harmonic compensation signal Yarufa _5, and outputs it without changing the phase of Ybeta ₅ . When a determination signal is input from the divergence determination unit 841, the phase change unit 842 changes Δθ ₅ and outputs fifth-order harmonic compensation signals Y′α ₅ and Y′β ₅ whose phases have been changed. When the determination signal is no longer input from the divergence determination unit 841, Δθ ₅ is fixed and the fifth-order harmonic compensation signals Y′α ₅ and Y′β ₅ whose phases are changed are output. If the fifth harmonic compensation signal Yα ₅ becomes larger when Δθ ₅ is changed (for example, increased), the phase changing unit 842 changes (for example, decreases) Δθ ₅ in the opposite direction, so that the control is performed. Search for Δθ ₅ for convergence.

If the control is in the divergent varies the [Delta] [theta] ₅ searches the appropriate [Delta] [theta] _5, if the control is not in the diverging trend, the fifth harmonic compensation properly phased to fix the [Delta] [theta] ₅ is changed Signals Y′α ₅ and Y′β ₅ are output. Thereby, it can prevent that control diverges.

  The configuration for preventing control divergence is not limited to the above. Control divergence may be prevented by other methods.

  FIG. 16 is a diagram for explaining a harmonic compensation controller according to another example of the fifth embodiment, and shows a fifth-order harmonic compensation unit 81 and a divergence prevention unit provided in the subsequent stage.

The divergence prevention unit 84 ′ is for preventing divergence in the suppression control of the fifth harmonic. When it is determined that the control tends to diverge, the divergence prevention unit 84 ′ stops outputting the _fifth harmonic compensation signals Yα ₅ and Yβ ₅ input from the fifth harmonic compensation unit 81. The divergence prevention unit 84 ′ is different from the divergence prevention unit 84 shown in FIG. 15 in that an output stop unit 843 is provided instead of the phase change unit 842.

Output stop unit 843, while not input a determination signal from a divergent decision unit 841 fifth harmonic compensation signal Yarufa _5, and outputs the Ybeta _5, when input a determination signal from a divergent decision unit 841 fifth harmonic The output of the wave compensation signals Yα ₅ and Yβ ₅ is stopped.

If the control is not in the diverging trend, the fifth harmonic compensation signal Yarufa ₅ inputted from the fifth harmonic compensator 81, Ybeta ₅ is outputted as it is, when the control is in the divergence tendency, the fifth harmonic compensation Output of the signals Yα ₅ and Yβ ₅ is stopped. When the fifth-order harmonic compensation signals Yα ₅ and Yβ ₅ are not output, the fifth-order harmonic suppression control is not performed, so that control divergence can be prevented. Even if the suppression control of the fifth harmonic is not performed, other controls are not affected.

A phase change unit 842 shown in FIG. 15 is provided before the output stop unit 843, and Δθ ₅ is searched while the fifth-order harmonic suppression control is stopped, and output after determining Δθ _5. You may make it cancel the stop of the stop part 843. FIG.

  In addition, the 7th harmonic compensation part 82 and the 11th harmonic compensation part 83 can also be provided with the same structure, and can prevent the divergence of each harmonic suppression control. Also in the second to fourth embodiments, the divergence of each harmonic suppression control can be similarly prevented.

  In the first to fifth embodiments, the case where the control circuit according to the present invention is used in a grid-connected inverter system (inverter system) has been described. However, the present invention is not limited to this. The present invention is, for example, an inverter that performs harmonic compensation used in a harmonic compensation device, a power active filter, an unbalance compensation device, a static reactive power compensation device (SVC, SVG), an uninterruptible power supply device (UPS), and the like. The present invention can also be applied to a control circuit that controls a circuit. For example, the harmonic compensator eliminates the current controller 74 (74 ′) or the voltage controller 94 in FIG. 8, FIG. 12, or FIG. 13, and specializes the function for suppressing harmonics by the harmonic compensation controller 8. . Moreover, the present invention is not limited to controlling an inverter circuit that converts direct current to three-phase alternating current. For example, control circuits such as a converter circuit that converts three-phase alternating current to direct current and a cycloconverter that converts the frequency of three-phase alternating current Can be applied. Hereinafter, a case where the present invention is applied to a control circuit of a converter circuit will be described as a sixth embodiment.

  FIG. 17 is a block diagram for explaining a three-phase PWM converter system according to the sixth embodiment. In the figure, the same or similar elements as those in the grid interconnection inverter system A shown in FIG.

  A three-phase PWM converter system C shown in FIG. 17 converts AC power supplied from the power system B into DC power and supplies it to a load L ′. The load L ′ is a DC load. The three-phase PWM converter system C includes a transformer circuit 4, a filter circuit 3, a current sensor 5, a voltage sensor 6, a converter circuit 10, and a control circuit 7.

  The transformer circuit 4 boosts or steps down the AC voltage input from the power system B to a predetermined level. The filter circuit 3 removes a high frequency component from the AC voltage input from the transformer circuit 4 and outputs it to the converter circuit 10. The current sensor 5 detects an alternating current of each phase input to the converter circuit 10. The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects an AC voltage of each phase input to the converter circuit 10. The detected voltage signal V is input to the control circuit 7. The converter circuit 10 converts the input AC voltage into a DC voltage and outputs it to the load L ′. The converter circuit 10 is a three-phase PWM converter, and is a voltage type converter circuit including three sets and six switching elements (not shown). The converter circuit 10 converts the input AC voltage into a DC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. The converter circuit 10 is not limited to this, and may be a current type converter circuit.

  The control circuit 7 controls the converter circuit 10. The control circuit 7 generates a PWM signal and outputs it to the converter circuit 10 in the same manner as the control circuit 7 of the first embodiment. In FIG. 17, only the configuration for performing input current control and harmonic compensation is described, and the configuration for other control is omitted. Although not shown, the control circuit 7 also includes a DC voltage controller and a reactive power controller, and also controls the output voltage and the input reactive power. The control method performed by the control circuit 7 is not limited to this. For example, when the converter circuit 10 is a current-type converter circuit, output current control may be performed instead of output voltage control.

  Also in this embodiment, the same effect as in the case of the first embodiment can be obtained. In the three-phase PWM converter system C, harmonic compensation is required to suppress harmonics of the input current. In this embodiment, a control system design for harmonic compensation is easily performed using linear control theory. It can be performed.

  The configuration of the three-phase PWM converter system C is not limited to the above. For example, instead of the control circuit 7, control circuits 7 'and 9 may be used. Alternatively, an inverter circuit may be provided on the output side of the converter circuit 10 to convert the DC power into AC power and supply the AC load to the AC load.

  The control circuit according to the present invention, the grid interconnection inverter system using the control circuit, and the three-phase PWM converter system are not limited to the above-described embodiments. The specific configuration of each part of the control circuit according to the present invention, the grid-connected inverter system using the control circuit, and the three-phase PWM converter system can be variously modified.

A Grid-connected inverter system A 'Inverter system 1 DC power supply 2 Inverter circuit (power conversion circuit)
DESCRIPTION OF SYMBOLS 3 Filter circuit 4 Transformer circuit 5 Current sensor 6 Voltage sensor 7, 7 ', 9 Control circuit 72 System opposition component generation part 73, 93 Three-phase / two-phase conversion part 74, 74' Current controller (control means)
76 Two-phase / three-phase converter 77 PWM signal generator 8, 8 'harmonic compensation controller 81, 81' fifth harmonic compensator (harmonic compensation means)
82,82 '7th harmonic compensation part (harmonic compensation means)
83, 83 '11th harmonic compensation part (harmonic compensation means)
84, 84 'divergence prevention unit 841 divergence determination unit 842 phase change unit 843 output stop unit 94 voltage controller (control means)
10 Converter circuit (Power conversion circuit)
B Power system C Three-phase PWM converter system L, L 'Load

Claims (18)

A control circuit that controls driving of a plurality of switching means in a power conversion circuit relating to three-phase alternating current using a PWM signal,
Three-phase two-phase conversion means for converting three signals based on the three-phase output or input of the power conversion circuit into a first signal and a second signal;
Harmonic compensation means for generating a first harmonic compensation signal and a second harmonic compensation signal by performing control to suppress predetermined harmonic components contained in the first signal and the second signal, respectively. When,
Two-phase three-phase conversion means for converting the first harmonic compensation signal and the second harmonic compensation signal into three correction value signals;
PWM signal generating means for generating a PWM signal based on the three correction value signals;
With
The harmonic compensation means includes
The first signal is processed by a first transfer function and a phase adjustment process is performed to generate the first harmonic compensation signal,
The second signal is signal-processed by the first transfer function, and a phase adjustment process is performed to generate the second harmonic compensation signal,
The first transfer function is a case where the transfer function representing a predetermined control process is F (s), the angular frequency of the fundamental wave of the three-phase alternating current is ω ₀ , the imaginary unit is j, and the nth harmonic is suppressed. ,
Is,
A control circuit characterized by that. The harmonic compensation means includes
The first signal is signal-processed by the first transfer function, the second signal is signal-processed by the second transfer function, and these are added to perform a phase adjustment process, whereby the first signal is processed. Generate a harmonic compensation signal of
The first signal is signal-processed by a third transfer function, the second signal is signal-processed by the first transfer function, and these are added to perform phase adjustment processing, whereby the second signal is processed. Generate a harmonic compensation signal of
The second transfer function and the third transfer function are:
When suppressing n-order harmonics (n = 3k + 1: k = 1, 2,...), respectively,
And
When suppressing the n-th harmonic (n = 3k + 2: k = 0, 1, 2,...)
Is,
The control circuit according to claim 1. Control for generating the first fundamental wave compensation signal and the second fundamental wave compensation signal by performing control for causing the fundamental wave components included in the first signal and the second signal to follow the respective target values. Means,
The first fundamental wave compensation signal and the first harmonic compensation signal are added to generate a first correction value signal, and the second fundamental wave compensation signal, the second harmonic compensation signal, Correction value signal generation means for generating a second correction value signal by adding
Further comprising
The two-phase / three-phase conversion means converts the first correction value signal and the second correction value signal into the three correction value signals.
The control circuit according to claim 1 or 2. A control circuit that controls driving of a plurality of switching means in a power conversion circuit relating to three-phase alternating current using a PWM signal,
Performing control to suppress predetermined harmonic components contained in the first signal, the second signal, and the third signal, respectively, which are three signals based on the three-phase output or input of the power conversion circuit; Harmonic compensation means for generating a first harmonic compensation signal, a second harmonic compensation signal, and a third harmonic compensation signal;
PWM signal generating means for generating a PWM signal based on the three harmonic compensation signals;
With
The harmonic compensation means includes
The first signal is signal-processed by a first transfer function, the second signal is signal-processed by a second transfer function, the third signal is signal-processed by the second transfer function, and these To adjust the phase to generate the first harmonic compensation signal,
The first signal is signal-processed by the second transfer function, the second signal is signal-processed by the first transfer function, and the third signal is signal-processed by the second transfer function. Then, by adding these and performing phase adjustment processing, the second harmonic compensation signal is generated,
The first signal is signal-processed by the second transfer function, the second signal is signal-processed by the second transfer function, and the third signal is signal-processed by the first transfer function. Then, by adding these and performing phase adjustment processing, the third harmonic compensation signal is generated,
The first transfer function and the second transfer function have a transfer function representing a predetermined control process as F (s), an angular frequency of the fundamental wave of the three-phase alternating current as ω ₀ , an imaginary unit as j, and n When suppressing the second harmonic,
Is,
A control circuit characterized by that. A control circuit that controls driving of a plurality of switching means in a power conversion circuit relating to three-phase alternating current using a PWM signal,
Performing control to suppress predetermined harmonic components contained in the first signal, the second signal, and the third signal, respectively, which are three signals based on the three-phase output or input of the power conversion circuit; Harmonic compensation means for generating a first harmonic compensation signal, a second harmonic compensation signal, and a third harmonic compensation signal;
PWM signal generating means for generating a PWM signal based on the three harmonic compensation signals;
With
The harmonic compensation means includes
Processing the first signal with a first transfer function, processing the second signal with a second transfer function, processing the third signal with a third transfer function, and By adding and performing phase adjustment processing, the first harmonic compensation signal is generated,
The first signal is signal-processed by the third transfer function, the second signal is signal-processed by the first transfer function, and the third signal is signal-processed by the second transfer function. Then, by adding these and performing phase adjustment processing, the second harmonic compensation signal is generated,
The first signal is signal-processed by the second transfer function, the second signal is signal-processed by the third transfer function, and the third signal is signal-processed by the first transfer function. Then, by adding these and performing phase adjustment processing, the third harmonic compensation signal is generated,
The first to third transfer functions are F (s) as a transfer function representing a predetermined control process, ω _{0 as} an angular frequency of the fundamental wave of the three-phase alternating current, and j as an imaginary unit,
When suppressing n-order harmonics (n = 3k + 1: k = 1, 2,...), respectively,
And
When suppressing the n-th harmonic (n = 3k + 2: k = 0, 1, 2,...)
Is,
A control circuit characterized by that. The first fundamental wave compensation signal and the second fundamental wave compensation are performed by controlling the fundamental wave components included in the first signal, the second signal, and the third signal to follow the respective target values. Control means for generating a signal and a third fundamental compensation signal;
The first fundamental wave compensation signal and the first harmonic compensation signal are added to generate a first correction value signal, and the second fundamental wave compensation signal, the second harmonic compensation signal, Are added to generate a second correction value signal, and the third fundamental value compensation signal and the third harmonic compensation signal are added to generate a third correction value signal. When,
Further comprising
The PWM signal generating means generates a PWM signal based on the first correction value signal, the second correction value signal, and the third correction value signal;
The control circuit according to claim 4 or 5. The control circuit according to claim 1, wherein a transfer function representing the predetermined control process is F (s) = K _I / s (where K _I is an integral gain). 7. The transfer function representing the predetermined control process is F (s) = K _P + K _I / s (where K _P and K _I are a proportional gain and an integral gain, respectively). Control circuit according to. The transfer function representing the predetermined control processing is F (s) = K _P + K _I / s + K _D · s (where K _P , K _I and K _D are proportional gain, integral gain and differential gain, respectively). A control circuit according to claim 1.   The control circuit according to claim 1, wherein the three signals are signals obtained by detecting an output current or an input current of each phase.   The control circuit according to claim 1, wherein the three signals are signals obtained by detecting an output voltage or an input voltage of each phase.   The control circuit according to claim 1, wherein the control system is designed using a robust control design.   The control circuit according to claim 12, wherein the control system is designed using the H∞ loop shaping method. Divergence determining means for determining that the control for suppressing the harmonic component tends to diverge based on the first or second harmonic compensation signal output from the harmonic compensation means;
Stop means for stopping the output of the first and second harmonic compensation signals when it is determined by the divergence determining means that there is a divergence tendency;
Further equipped with,
The control circuit according to claim 1. Divergence determining means for determining that the control for suppressing the harmonic component tends to diverge based on the first or second harmonic compensation signal output from the harmonic compensation means;
A phase changing means for changing the phase of the first and second harmonic compensation signals to a phase in which the control does not diverge when it is determined by the divergence determining means that there is a divergence tendency;
Further equipped with,
The control circuit according to claim 1. The divergence determining means determines that the first or second harmonic compensation signal is in a divergence tendency because it exceeds a predetermined threshold.
The control circuit according to claim 14 or 15.   A grid-connected inverter system comprising an inverter circuit and the control circuit according to any one of claims 1 to 16.   A three-phase PWM converter system comprising a converter circuit and the control circuit according to any one of claims 1 to 16.