JP5944670B2 - Control circuit for power conversion circuit, grid-connected inverter system using this control circuit, and single-phase PWM converter system - Google Patents

Description

  The present invention relates to a control circuit for controlling the output or input of a power conversion circuit, a grid-connected inverter system and a single-phase PWM converter system using the control circuit, and more particularly to a circuit for compensating for harmonics.

  2. Description of the Related Art Conventionally, a grid-connected inverter system has been developed that converts DC power generated by a solar cell or the like into AC power and supplies it to an electric power system.

  FIG. 12 is a block diagram for explaining a conventional grid-connected inverter system of a general single-phase power system. The grid interconnection inverter system A100 converts the power generated by the DC power supply 1 and supplies it to a single-phase power system B (hereinafter referred to as “power system B”).

  The inverter circuit 2 converts a DC voltage input from the DC power supply 1 into an AC voltage by switching a switching element (not shown). The filter circuit 3 removes a switching frequency component included in the AC voltage output from the inverter circuit 2. The transformer circuit 4 boosts (or steps down) the AC voltage output from the filter circuit 3 to the system voltage of the power system B. The control circuit 700 receives the current signal and the voltage signal detected by the current sensor 5 and the voltage sensor 6, generates a PWM signal based on the current signal and the voltage signal, and outputs the PWM signal to the inverter circuit 2. The inverter circuit 2 performs switching of the switching element based on the PWM signal input from the control circuit 700.

  FIG. 13 is a block diagram for explaining the internal configuration of the control circuit 700.

  The current signal input from the current sensor 5 is input to the Hilbert conversion unit 73 and the rotation coordinate conversion unit 78.

The Hilbert transform unit 73 delays the phase of the input current signal by π / 2 (90 degrees) by the Hilbert transform. An ideal Hilbert transform is represented by a transfer function H (ω) shown in the following equation (1). Where ω _S is the sampling angular frequency and j is the imaginary unit. That is, the Hilbert transform is a filter process in which the amplitude characteristic is constant regardless of the frequency and the phase characteristic is delayed by π / 2 (90 degrees) in a positive / negative frequency region. Since ideal Hilbert transform cannot be realized, it is approximately realized as, for example, a FIR (Finite impulse response) filter.

  The Hilbert transformed signal is obtained by delaying the phase of the original signal by π / 2. Therefore, since the so-called rotational coordinate conversion process (dq conversion process) can be performed by using the signal after the Hilbert transform and the original signal, the rotation can be performed even in the case of single-phase alternating current as in the case of three-phase alternating current. Control in the coordinate system can be performed.

  The rotation coordinate conversion unit 78 includes a current signal input from the current sensor 5 (hereinafter referred to as “α-axis current signal Iα”) and a current signal input from the Hilbert conversion unit 73 (hereinafter referred to as “β-axis current signal Iβ”). Is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system. The rotating coordinate system is an orthogonal coordinate system having orthogonal d-axis and q-axis and rotating in the same rotational direction at the same angular velocity as the fundamental wave of the system voltage of the power system B. As a concept opposite to the rotating coordinate system, a non-rotating coordinate system is a stationary coordinate system. The rotation coordinate conversion unit 78 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 78 is expressed by a determinant represented by the following expression (2).

  The LPF 74a and the LPF 75a are low-pass filters and pass only the DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. Through the rotation coordinate conversion process, the fundamental wave components of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. The PI control unit 74b and the PI control unit 75b perform PI control (proportional integration control) based on the deviation between the DC component of the d-axis current signal Id and the q-axis current signal Iq and the target value, respectively, and the fundamental compensation signal Xd and Xq are output. Since a DC component can be used as the target value, the PI control unit 74b and the PI control unit 75b can perform control with high accuracy.

  The stationary coordinate conversion unit 79 converts the fundamental wave compensation signals Xd and Xq input from the PI control unit 74b and the PI control unit 75b, respectively, into two fundamental wave compensation signals Xα and Xβ in the stationary coordinate system. The rotation coordinate conversion unit 78 performs a reverse conversion process. The stationary coordinate conversion unit 79 performs a so-called stationary coordinate conversion process (inverse dq conversion process), and the fundamental wave compensation signals Xd and Xq of the rotating coordinate system are converted to the fundamental wave compensation of the stationary coordinate system based on the phase θ. The signals are converted into signals Xα and Xβ.

The conversion process performed by the stationary coordinate conversion unit 79 is expressed by a determinant represented by the following expression (3).

  The control circuit 700 has a function of suppressing harmonics input from the power system B and harmonics output from the inverter circuit 2. The harmonic compensation controller 800 extracts a harmonic component from the current signal input from the current sensor 5 and outputs a harmonic compensation signal for outputting a harmonic that cancels the harmonic component. The grid interconnection inverter system A100 outputs harmonics based on the harmonic compensation signal (that is, harmonics having a phase opposite to the detected harmonics) and cancels the harmonics, thereby suppressing harmonics.

  FIG. 14 is a block diagram for explaining the internal configuration of the harmonic compensation controller 800. In general, the harmonics from the power system B or the inverter circuit 2 are mostly the fifth harmonic, the seventh harmonic, and the eleventh harmonic. In order to suppress these harmonics, a fifth harmonic compensator 810 for suppressing the fifth harmonic, a seventh harmonic compensator 820 for suppressing the seventh harmonic, and an eleventh harmonic are provided. An eleventh harmonic compensation unit 830 for suppression is provided in the harmonic compensation controller 800. The fifth harmonic compensation unit 810 includes a rotation coordinate conversion unit 811, LPFs 812 and 813, I control units 814 and 815, and a stationary coordinate conversion unit 816. Since the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 have the same configuration as the fifth harmonic compensation unit 810, the description and description in FIG. 14 are omitted.

The rotation coordinate conversion unit 811 converts the α-axis current signal Iα input from the current sensor 5 and the β-axis current signal Iβ input from the Hilbert conversion unit 73 into a d-axis current signal Id ₅ and a q-axis current signal in the rotation coordinate system. Iq ₅ is converted. This rotating coordinate system is an orthogonal coordinate system that rotates in the opposite direction at an angular velocity that is five times the angular velocity of the fundamental wave of the system voltage. The rotation coordinate conversion unit 811 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 811 is represented by a determinant represented by the following expression (4).
Note that the rotational coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (4).

LPF 812 and LPF 813 are low-pass filters and pass only the DC components of d-axis current signal Id ₅ and q-axis current signal Iq ₅ , respectively. By the rotational coordinate conversion process, the fifth harmonics of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively. The I control unit 814 and the I control unit 815 perform I control (integration control) based on the DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively, and the fifth harmonic compensation signals Yd ₅ , Yq ₅ is output. Since a DC component can be used as the target value, the I control unit 814 and the I control unit 815 can perform highly accurate control.

The stationary coordinate conversion unit 816 converts the fifth harmonic compensation signals Yd ₅ and Yq ₅ input from the I control unit 814 and the I control unit 815, respectively, into two fifth harmonic compensation signals Yα ₅ and Yβ in the stationary coordinate system. ₅ is converted, and the conversion process reverse to that of the rotating coordinate conversion unit 811 is performed. The static coordinate conversion unit 816 performs so-called static coordinate conversion processing (inverse dq conversion processing), and converts the fifth-order harmonic compensation signals Yd ₅ and Yq ₅ of the rotating coordinate system into a static coordinate system based on the phase θ. of the fifth harmonic compensation signal Yarufa _5, converted into Ybeta _5.

The conversion process performed by the stationary coordinate conversion unit 816 is expressed by a determinant represented by the following expression (5).
Note that the stationary coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (5).

Similarly, the seventh harmonic compensation unit 820 generates _seventh harmonic compensation signals Yα ₇ and Yβ ₇ , and the eleventh harmonic compensation unit 830 generates _eleventh harmonic compensation signals Yα ₁₁ and Yβ ₁₁ . A harmonic compensation signal Yα obtained by adding the fifth harmonic compensation signal Yα ₅ , the seventh harmonic compensation signal Yα ₇ , and the eleventh harmonic compensation signal Yα ₁₁ is output from the harmonic compensation controller 800.

  The PWM signal generation unit 77 is based on a correction value signal X′α obtained by adding the fundamental wave compensation signal Xα output from the stationary coordinate conversion unit 79 and the harmonic compensation signal Yα output from the harmonic compensation controller 800. Generate and output a PWM signal.

Japanese Patent Laid-Open No. 2003-143860 Japanese Patent No. 4421700

"Control method of single-phase grid-connected inverter using Hilbert transform" Electron Theory D, Vol. 121, No. 10, 2001

  Since the Hilbert transform unit 73 cannot realize an ideal Hilbert transform, the Hilbert transform unit 73 is configured by, for example, an FIR filter to realize an approximate Hilbert transform. Since a group delay occurs in the FIR filter, the β-axis current signal Iβ output from the Hilbert transform unit 73 is delayed with respect to the α-axis current signal Iα. For this reason, although not shown in FIG. 12, in actual processing, it is necessary to adjust the phase by providing a delay circuit between the current sensor 5 and the rotation coordinate conversion unit 78. The FIR filter constituting the Hilbert transform unit 73 has a characteristic that if the order n is increased to widen the conversion band, the ripple in the frequency characteristics can be reduced, but the group delay is increased and the detection is delayed. On the contrary, if the order n is reduced to narrow the conversion band, the group delay can be suppressed, but the ripple in the frequency characteristic becomes large and the detection accuracy is lowered.

  Therefore, when the Hilbert transform unit 73 is used, there is an inconvenience that the order n of the FIR filter to be configured must be designed in consideration of the tradeoff between the detection speed and the detection accuracy. In addition, in order to obtain a certain degree of detection accuracy, it is necessary to increase the order n of the FIR filter. As a result, the configuration of the Hilbert transform unit 73 becomes large, which causes a problem that the control circuit 700 is complicated.

  There is also a problem that designing a control system for current control requires a great deal of effort. Recent grid-connected inverter systems are required to have high-speed responsiveness in control, such as returning the output within a predetermined time with respect to the instantaneous drop. In order to design the control system so as to satisfy such requirements, it is necessary to optimally design the parameters of the LPF 74a and the LPF 75a and the proportional gain and integral gain of the PI control unit 74b and the PI control unit 75b. However, since the rotating coordinate conversion unit 78 and the stationary coordinate conversion unit 79 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  Furthermore, in order to design a control system for each harmonic compensation, it is necessary to optimally design the parameters of the LPFs 812 and 813 and the integral gains of the I controllers 814 and 815. However, since the rotating coordinate conversion unit 811 and the stationary coordinate conversion unit 816 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  The present invention has been conceived under the circumstances described above, and can perform high-speed and high-accuracy current control with a simple configuration, and also perform harmonic compensation in the current control control system. An object of the present invention is to provide a control circuit in which the control system has linearity and time invariance.

  In order to solve the above problems, the present invention takes the following technical means.

A control circuit provided by the first aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to single-phase alternating current using a PWM signal, and outputs or inputs of the power conversion circuit Deviation signal generating means for generating a deviation signal which is a deviation between the signal based on the fundamental wave component and the target value of the fundamental wave component, and the fundamental wave component included in the deviation signal is controlled to be zero and included in the deviation signal A control unit that generates a correction value signal for suppressing and controlling a predetermined harmonic component; and a PWM signal generation unit that generates a PWM signal based on the correction value signal. The correction signal is generated by processing the deviation signal by a transfer function designed by a robust control design method using W, and the angular frequency of the fundamental wave of the single-phase AC and .omega.0, To suppress n-th harmonic, the denominator of the frequency weight W are multiplied (s ² + ω ₀ ²⁾ and corresponding to each ^{n {s 2 + (n ·} ω 0) 2} The numerator is characterized in that (s ² + ω ₀ ² ) and {s ² + (n · ω ₀ ) ² } corresponding to each n are not multiplied .

In a preferred embodiment of the present invention, when the frequency weight W is a predetermined real number k,
It is.

  In a preferred embodiment of the present invention, the signal based on the output current or the input current is a signal detected.

  In a preferred embodiment of the present invention, the signal based on the output voltage or the input voltage is detected.

  In a preferred embodiment of the present invention, the transfer function is designed by the H∞ loop shaping method using the frequency weight W.

  The grid interconnection inverter system provided by the second aspect of the present invention includes an inverter circuit and a control circuit provided by the first aspect of the present invention.

  The single-phase PWM converter system provided by the third aspect of the present invention includes a converter circuit and a control circuit provided by the first aspect of the present invention.

According to the present invention, since it is not necessary to generate a signal whose phase is delayed by 90 degrees from a signal based on an output or an input, there is no inconvenience of designing an FIR filter for Hilbert transform processing. Further, the configuration can be simplified. In addition, according to the present invention, the correction value signal is generated by performing signal processing on the deviation signal using a transfer function. The transfer function is designed using a frequency weight W including (s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } in the denominator. Are controlled simultaneously. Therefore, high-speed and high-accuracy current control can be performed with a simple configuration, and harmonic compensation can also be performed in the current control control system. Further, since the control system has linearity and time invariance, the design of the control system can be facilitated using a design method based on the linear control theory.

  Other features and advantages of the present invention will become more apparent from the detailed description given below with reference to the accompanying drawings.

It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process. It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process, and is represented by the matrix. It is a block diagram for demonstrating the calculation of a matrix. It is a block diagram which shows the process which performs static coordinate transformation after performing PI control after performing rotational coordinate transformation. It is a block diagram which shows the process which performs a stationary coordinate transformation after performing I control after performing a rotational coordinate transformation. Is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. It is a Bode diagram for analyzing the frequency weight W. It is a Bode diagram for analyzing a transfer function K. It is a block diagram for demonstrating the grid connection inverter system which concerns on 1st Embodiment. It is a block diagram for demonstrating the control circuit which concerns on 2nd Embodiment. It is a block diagram for demonstrating the single phase PWM converter system which concerns on 3rd Embodiment. It is a block diagram for demonstrating the grid connection inverter system of the conventional common single phase electric power system. It is a block diagram for demonstrating the internal structure of a control circuit. It is a block diagram for demonstrating the internal structure of a harmonic compensation controller.

  Embodiments of the present invention will be specifically described below with reference to the drawings.

  First, a method for converting processing involving rotational coordinate transformation and stationary coordinate transformation into linear time-invariant processing will be described.

  Fig.1 (a) is a figure for demonstrating the process accompanied by rotation coordinate transformation and stationary coordinate transformation. In this processing, first, the signals α and β are converted into signals d and q by rotational coordinate conversion. The signals d and q are each processed by a predetermined transfer function F (s), and signals d 'and q' are output. Next, the signals d ′ and q ′ are converted into signals α ′ and β ′ by stationary coordinate transformation. The nonlinear time-varying process shown in FIG. 1A is converted into a process using a matrix G of linear time-invariant transfer functions shown in FIG.

The rotational coordinate transformation shown in FIG. 1A is represented by a determinant of the following equation (6), and the stationary coordinate transformation is represented by a determinant of the following equation (7).

  Therefore, the process shown in FIG. 1A can be expressed as shown in FIG. 2A using a matrix. The matrix G shown in FIG. 1B can be calculated by calculating the product of the three matrices shown in FIG. 2A and making the calculated matrix a linear time-invariant matrix. At this time, the calculation is performed after converting the matrix of the stationary coordinate conversion and the rotation coordinate conversion into a matrix product.

The matrix of rotational coordinate conversion can be converted into the product of the right-hand side matrix shown in the following equation (8).
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

Moreover, the matrix of a static coordinate transformation | transformation can be converted into the product of the matrix of the right side shown in following (9) Formula. The matrix in the middle of the matrix product is a linear time-invariant matrix.
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

When the product of the three matrices shown in FIG. 2A is calculated using the above equations (8) and (9) and the matrix G is calculated, the following equation (10) is obtained.

If attention is paid to the element in the first row and the first column of the three matrixes in the center of the above equation (10) and this is represented by a block diagram, the block diagram shown in FIG. When calculating the input / output characteristics of the block diagram shown in FIG.
It becomes. Where F (s) is a one-input one-output transfer function having an impulse response f (t).

Here, when θ (t) = ω ₀ t, θ (t) −θ (τ) = ω ₀ t−ω ₀ τ = ω ₀ (t−τ) = θ (t−τ). The input / output characteristics of the block diagram shown in FIG. 3 are equal to those of a linear time invariant system having an impulse response f (t) exp (−jω ₀ t). When the impulse response f (t) exp (−jω ₀ t) is Laplace transformed, a transfer function F (s + jω ₀ ) is obtained. The input / output characteristics when exp (jθ (t)) and exp (−jθ (t)) in the block diagram shown in FIG. 3 are interchanged are the input / output characteristics of the transfer function F (s−jω ₀ ). become.

Therefore, if the calculation is further advanced from the above equation (10),
Is calculated.

  Thereby, the process shown in FIG. 2A can be converted into the process shown in FIG. The process shown in FIG. 2B is equivalent to the process of performing the stationary coordinate conversion after performing the process represented by the predetermined transfer function F (s) after performing the rotational coordinate conversion. This system is a linear time-invariant system.

The transfer function of the PI control (proportional integral control) controller is expressed as F (s) = K _P + K _I / s, where K _P and K _I are the proportional gain and integral gain, respectively. Therefore, the transfer function matrix G _PI showing the process shown in FIG. 4, that is, the process equivalent to the process of performing the static coordinate conversion after performing the PI control after performing the rotational coordinate conversion, uses the above equation (11). Then, it is calculated as in the following equation (12).

Further, the transfer function of the I control (integral control) controller, the integral gain and K _I, represented by _{F (s) = K I /} s. Therefore, the processing shown in FIG. 5, i.e., the matrix G _I of the transfer function from performing a rotational coordinate transformation showing the processing of the processing equivalent to a still coordinate transformation after the I control, using the above equation (11) Thus, the following equation (13) is calculated.

Figure 6 is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. FIG (a) is first row and the first column elements (hereinafter, the same applies for. Other elements described as "(1,1) element".) The matrix G _I and (2,2) element of FIG. 4B shows the transfer function of the (1,2) element of the matrix G _I , and FIG. 4C shows the transfer function of the (2,1) element of the matrix G _I. Is shown. The figure shows the case where the frequency of the fundamental wave of the system voltage (hereinafter referred to as “center frequency”. Also, the angular frequency corresponding to the center frequency is referred to as “center angular frequency”) is 60 Hz (that is, angular frequency). ω ₀ = 120π), and the integral gain K _I is “0.1”, “1”, “10”, “100”.

FIG (a), the amplitude characteristic shown by (b) and (c) are all, there is a peak in the center frequency, the integral gain K _I is increased, the amplitude characteristic is increased. Further, the phase characteristic shown in FIG. 5A is 0 degree at the center frequency. In other words, the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without the signal of the center frequency (center angular frequency) changes the phase. The phase characteristic shown in FIG. 5B is 90 degrees at the center frequency. In other words, the transfer function of the (1,2) element of the matrix G _I causes the phase of the signal of the center frequency (center angular frequency) passes advancing 90 degrees. On the other hand, the phase characteristic shown in FIG. 4C is −90 degrees at the center frequency. In other words, the transfer function of (2,1) element of the matrix G _I is the phase of the signal of the center frequency (center angular frequency) passing delayed 90 degrees.

Consider a case where the phases are treated with 90-degree different signals α and the signal beta (the phase of the signal α is advanced 90 degrees from the signal beta.) The matrix G _I. Even if the processing shown in the transfer function of (1, 1) element of the matrix G _I in signal alpha, phase does not change (see FIG. 6 (a)). Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I in signal beta, advances the phase by 90 degrees (see Figure 6 (b)). Therefore, since the phase of both is the same as that of the signal α, the two are added together to strengthen each other. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I in signal alpha, phase is delayed 90 degrees (see Figure 6 (c)). Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in signal beta, phase does not change. Therefore, since both phases are the same as the signal β, they are strengthened by adding both.

The process shown in the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without changing the phase (see FIG. 6 (a)). Therefore, even if the matrix G ′ _I of the transfer function shown in the following equation (14) in which the (1,2) and (2,1) elements of the matrix G _I shown in the above equation (13) are set to “0” is used. A similar process can be performed. In the case of the following equation (14), since the (13) minutes is not constructive as compared with the case of expression, it is necessary to design the integral gain K _I in correspondingly large value.

That is, if the signal α and the signal β are processed by the transfer function G _I (s) shown in the following equation (15), the stationary coordinate conversion is performed after performing the I control after performing the rotational coordinate conversion. The same processing is performed. In the case of a single-phase system, it is only necessary to perform processing on the signal α, so that it is not necessary to generate the signal β whose phase is delayed by 90 degrees.

  Next, a method for controlling harmonic components will be described.

The transfer function G _I (s) shown in the above equation (15) is for controlling the fundamental wave component. The n-th harmonic is an angular frequency component obtained by multiplying the angular frequency of the fundamental wave by n. Therefore, the transfer function in the case of controlling the nth harmonic is the transfer function G _In (s) shown in the following equation (16) where ω ₀ is n · ω ₀ in the above equation (15).

As shown in the above equation (15), when controlling the fundamental wave, the controller can be designed by the H∞ loop shaping method using the frequency weight W ₁ shown in the following equation (17). By using the H∞ loop shaping method, the most stable controller that satisfies the design specifications can be designed.

Further, as shown in Equation (16), when performing control of the n-th harmonic, it is possible to design a controller with H∞ loop shaping method using the frequency weight W _n shown in the following equation (18).

When controlling the fundamental wave and controlling the nth harmonic, the controller may be designed by the H∞ loop shaping method using the frequency weight W shown in the following equation (19). Note that k is a real number corresponding to the response speed.

When a plurality of harmonics are controlled, a corresponding order term may be added to the denominator of the equation (19). For example, when controlling the fifth, seventh, and eleventh harmonics, the frequency weight W shown in the following equation (20) may be used.

FIG. 7 is a Bode diagram for analyzing the frequency weight W shown in the above equation (20). This figure shows a case where the angular frequency ω ₀ = 120π and k = 10 ²⁴ . As shown in the figure, the frequency weights W are ω ₀ (= 120π≈377 [rad / sec]), 5ω ₀ (= 600π≈1884 [rad / sec]), 7ω ₀ (= 840π≈2638 [rad / sec]. sec]), 11ω ₀ (= 1320π≈4145 [rad / sec]).

When the controller is designed by the H∞ loop shaping method using this frequency weight W, a transfer function K shown in the following equation (21) is calculated. 9 is calculated when the inductance of the reactor of the LC filter included in the filter circuit 3 of FIG. 9 described later is L = 1000 μH, the capacitance of the capacitor is C = 20 μF, and the leakage inductance of the transformer circuit 4 is LT = 500 μH. ing.

FIG. 8 is a Bode diagram for analyzing the transfer function K shown in the equation (21). As shown in the figure, the transfer function K inherits the characteristic that peaks at ω ₀ , 5ω ₀ , 7ω ₀ , and 11ω ₀ .

  Hereinafter, a case where a controller designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (20) is applied to a control circuit of a grid-connected inverter system will be described as a first embodiment of the present invention. .

  FIG. 9 is a block diagram for explaining the grid-connected inverter system according to the first embodiment.

  As shown in the figure, the grid-connected inverter system A includes a DC power source 1, an inverter circuit 2, a filter circuit 3, a transformer circuit 4, a current sensor 5, a voltage sensor 6, and a control circuit 7.

  The DC power source 1 is connected to the inverter circuit 2. The inverter circuit 2, the filter circuit 3, and the transformer circuit 4 are connected in series to the output line in this order, and are connected to the single-phase AC power system B. The current sensor 5 and the voltage sensor 6 are installed on the output side of the transformer circuit 4. The control circuit 7 is connected to the inverter circuit 2. The grid interconnection inverter system A converts the DC power output from the DC power supply 1 into AC power and supplies it to the power grid B. In addition, the structure of the grid connection inverter system A is not restricted to this. For example, the current sensor 5 and the voltage sensor 6 may be provided on the input side of the transformer circuit 4, or other sensors necessary for controlling the inverter circuit 2 may be provided. Further, the transformer circuit 4 may be provided on the input side of the filter circuit 3, or a so-called transformer-less system in which the transformer circuit 4 is not provided. Further, a DC / DC converter circuit may be provided between the DC power supply 1 and the inverter circuit 2.

  The DC power source 1 outputs DC power and includes, for example, a solar battery. A solar cell generates direct-current power by converting solar energy into electrical energy. The DC power source 1 outputs the generated DC power to the inverter circuit 2. Note that the DC power source 1 is not limited to one that generates DC power from a solar cell. For example, the DC power source 1 may be a fuel cell, a storage battery, an electric double layer capacitor, a lithium ion battery, or AC power generated by a diesel engine generator, a micro gas turbine generator, a wind turbine generator, or the like. It may be a device that converts to DC power and outputs it.

  The inverter circuit 2 converts a DC voltage input from the DC power source 1 into an AC voltage and outputs the AC voltage to the filter circuit 3. The inverter circuit 2 is a single-phase inverter and is a PWM control type inverter circuit provided with two sets of four switching elements (not shown). The inverter circuit 2 converts the DC voltage input from the DC power source 1 into an AC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. In addition, the inverter circuit 2 is not limited to this, For example, a multilevel inverter may be sufficient.

  The filter circuit 3 removes high frequency components due to switching from the AC voltage input from the inverter circuit 2. The filter circuit 3 includes a low pass filter including a reactor and a capacitor. The AC voltage from which the high frequency component has been removed by the filter circuit 3 is output to the transformer circuit 4. The configuration of the filter circuit 3 is not limited to this, and any known filter circuit for removing high frequency components may be used. The transformer circuit 4 boosts or lowers the AC voltage output from the filter circuit 3 to a level substantially the same as the system voltage.

  The current sensor 5 detects an alternating current output from the transformer circuit 4 (that is, an output current of the grid interconnection inverter system A). The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects the system voltage of the power system B. The detected voltage signal V is input to the control circuit 7. Note that the output voltage output by the grid interconnection inverter system A substantially matches the grid voltage.

  The control circuit 7 controls the inverter circuit 2 and is realized by, for example, a microcomputer. The control circuit 7 generates a PWM signal based on the current signal I input from the current sensor 5 and the voltage signal V input from the voltage sensor 6 and outputs the PWM signal to the inverter circuit 2. The control circuit 7 generates a command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A based on the detection signal input from each sensor, and is generated based on the command value signal. Output a pulse signal as a PWM signal. The inverter circuit 2 outputs an alternating voltage having a waveform corresponding to the command value signal by switching each switching element on and off based on the input PWM signal. The control circuit 7 controls the output current by changing the waveform of the command value signal to change the waveform of the output voltage of the grid interconnection inverter system A. Thereby, the control circuit 7 performs various feedback controls. In addition, the control circuit 7 suppresses harmonics by causing the inverter circuit 2 to output harmonics for canceling the harmonics input from the power system B.

  In FIG. 9, only the configuration for performing the output current control and the harmonic suppression control is described, and the configuration for the other control is omitted. Actually, the control circuit 7 performs DC voltage control (feedback control performed so that the input DC voltage becomes a preset voltage target value) and reactive power control (reactive power target value with preset output reactive power) (Feedback control to be performed) is also performed. The control method performed by the control circuit 7 is not limited to this. For example, output voltage control or active power control may be performed.

  The control circuit 7 includes a system counter component generation unit 72, a current controller 74, and a PWM signal generation unit 77.

  The system counter-part generating unit 72 receives the voltage signal V from the voltage sensor 6 and generates and outputs a system command value signal K. The system command value signal K serves as a reference for the command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A, and the system command value signal K is corrected by a correction value signal X described later. Thus, a command value signal is generated.

The current controller 74 receives a deviation between the current signal I detected by the current sensor 5 and the current target value, and generates a correction value signal X for current control. The current controller 74 is a controller designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (20). The processing performed by the current controller 74 is linear time invariant processing. Therefore, a control system design using linear control theory can be performed. As the angular frequency ω ₀ , the angular frequency of the fundamental wave of the system voltage (for example, ω ₀ = 120π [rad / sec] (60 [Hz])) is set in advance, and the real number k is also set in advance. Further, the current controller 74 performs a process for maximizing the stability margin, and the phase adjustment for correcting the phase delay in the control loop is also performed. The design method used for control system design is not limited to the H∞ loop shaping method, and for example, a robust control design method such as optimal control, H∞ control theory, and mixed sensitivity problem can be used.

  The current controller 74 controls the fundamental wave component included in the current signal I to a target value, and performs control to suppress the fifth, seventh, and eleventh harmonic components included in the current signal I.

In the present embodiment, the case where the current controller 74 performs suppression control of the fifth, seventh, and eleventh harmonics has been described, but the present invention is not limited to this. The frequency weight W may be set according to the harmonic order that needs to be suppressed. For example, when it is desired to suppress only the fifth harmonic, the denominator of the frequency weight W shown in the above equation (20) may be (s ² + ω ₀ ² ) · {s ² + (5ω ₀ ) ² } only. Further, if it is desired to further suppress the 13th harmonic, {s ² + (13ω ₀ ) ² } may be further applied to the denominator of the frequency weight W shown in the above equation (20).

  In the present embodiment, a current target value on the α-axis generated by stationary coordinate conversion of the d-axis current target value and the q-axis current target value is used as the current target value. A correction value for DC voltage control (not shown) is used for the d-axis current target value, and a correction value for reactive power control (not shown) is used for the q-axis current target value. When the current target value on the α axis is directly given, the target value may be used as it is.

  The system command value signal K output from the system counter-part generating unit 72 and the correction value signal X output from the current controller 74 are added to calculate the command value signal X ′ and input to the PWM signal generating unit 77. .

  The PWM signal generation unit 77 is based on an input command value signal X ′ and a signal obtained by inverting the command value signal X ′ and a carrier signal generated as a triangular wave signal having a predetermined frequency (for example, 4 kHz), PWM signals Pp and Pn are generated by a triangular wave comparison method. In the triangular wave comparison method, for example, the command value signal X ′ and the carrier signal are compared, and a pulse signal that becomes high level when the command value signal X ′ is larger than the carrier signal and becomes low level when the command value signal X ′ is smaller is the PWM signal Pp. Generated. Similarly, the inverted signal and the carrier signal are compared, and the PWM signal Pn is generated. The generated PWM signals Pp and Pn are output to the inverter circuit 2. The PWM signal generator 77 also outputs a signal obtained by inverting the PWM signals Pp and Pn to the inverter circuit 2.

  In the present embodiment, since the control circuit 7 does not need to generate a signal whose phase is delayed by 90 degrees from the current signal I, it is not necessary to provide the Hilbert transform unit 73 shown in FIG. Further, since only the current signal I needs to be processed (since it is not necessary to process a signal delayed by 90 degrees), the configuration is simplified. Furthermore, since the Hilbert transform unit 73 is unnecessary, there is no inconvenience of designing the order n of the FIR filter constituting the Hilbert transform unit 73.

  Further, the current controller 74 performs the same processing as the rotation coordinate conversion unit 78, the LPFs 74a and 75a, the PI control units 74b and 75b, the stationary coordinate conversion unit 79, and the harmonic compensation controller 800 (see FIG. 14) shown in FIG. It is carried out. Therefore, the control circuit 7 can perform high-speed and high-accuracy current control with a simple configuration, and can also perform harmonic compensation in the current control control system.

  Furthermore, the processing performed by the current controller 74 is linear time-invariant processing. Further, the control circuit 7 does not include a rotation coordinate conversion process and a static coordinate conversion process, which are nonlinear time-varying processes, and the control circuit 7 performs control in a static coordinate system. That is, the entire current control system is a linear time invariant system. That is, the control system for current control and the control system for each harmonic compensation are set as one control system, and the control system is set as a linear time-invariant process. Therefore, linear control theory can be used for control system design, and control system design can be facilitated.

  In the first embodiment, the case where the output current is controlled has been described, but the present invention is not limited to this. For example, the output voltage may be controlled. Hereinafter, the case of controlling the output voltage will be described as a second embodiment.

  FIG. 10 is a block diagram for explaining a control circuit according to the second embodiment. In the same figure, the same code | symbol is attached | subjected to the element which is the same as that of the grid connection inverter system A shown in FIG. 9, or similar.

  The inverter system A ′ shown in FIG. 10 differs from the grid-connected inverter system A (see FIG. 9) according to the first embodiment in that power is supplied to the load L instead of the power system B. Since it is necessary to control the voltage supplied to the load L, the control circuit 8 controls not the output current but the output voltage. The control circuit 8 differs from the control circuit 7 according to the first embodiment (see FIG. 9) in that a PWM signal is generated based on the voltage signal V input from the voltage sensor 6. The inverter system A ′ supplies power to the load L while controlling the output voltage to a target value by feedback control.

  The voltage controller 84 receives a deviation between the voltage signal V detected by the voltage sensor 6 and the voltage target value, and generates a correction value signal X for voltage control. The voltage controller 84 is a controller designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (20).

  Also in this embodiment, the control circuit 8 does not need to generate a signal whose phase is delayed by 90 degrees from the voltage signal V. Further, the voltage controller 84 uses a control system for voltage control and a control system for compensation of each harmonic as one control system, and the control system is a linear time-invariant process. Therefore, the same effect as that of the first embodiment can be obtained.

  In the said 1st or 2nd embodiment, although the case where the control circuit which concerns on this invention was used for the grid connection inverter system (inverter system) was demonstrated, it is not restricted to this. The present invention is also applied to a control circuit for controlling an inverter circuit that performs harmonic compensation used in, for example, a power active filter, a static reactive power compensator (SVC, SVG), an uninterruptible power supply (UPS), and the like. be able to. Moreover, the present invention is not limited to controlling an inverter circuit that converts direct current to single-phase alternating current. For example, it is also applicable to a control circuit such as a converter circuit that converts single-phase alternating current to direct current or a cycloconverter that converts the frequency of single-phase alternating current. Can be applied. Hereinafter, a case where the present invention is applied to a control circuit of a converter circuit will be described as a third embodiment.

  FIG. 11 is a block diagram for explaining a single-phase PWM converter system according to the third embodiment. In the same figure, the same code | symbol is attached | subjected to the element which is the same as that of the grid connection inverter system A shown in FIG. 9, or similar.

  A single-phase PWM converter system C shown in FIG. 11 converts AC power supplied from the power system B into DC power and supplies it to a load L ′. The load L ′ is a DC load. The single-phase PWM converter system C includes a transformer circuit 4, a filter circuit 3, a current sensor 5, a voltage sensor 6, a converter circuit 9, and a control circuit 7.

  The transformer circuit 4 boosts or steps down the AC voltage input from the power system B to a predetermined level. The filter circuit 3 removes a high frequency component from the AC voltage input from the transformer circuit 4 and outputs it to the converter circuit 9. The current sensor 5 detects an alternating current input to the converter circuit 9. The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects an AC voltage input to the converter circuit 9. The detected voltage signal V is input to the control circuit 7. The converter circuit 9 converts the input AC voltage into a DC voltage and outputs it to the load L ′. The converter circuit 9 is a single-phase PWM converter, and is a voltage type converter circuit including two sets of four switching elements (not shown). The converter circuit 9 converts the input AC voltage into a DC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. The converter circuit 9 is not limited to this, and may be a current type converter circuit.

  The control circuit 7 controls the converter circuit 9. The control circuit 7 generates a PWM signal and outputs it to the converter circuit 9 in the same manner as the control circuit 7 of the first embodiment. In FIG. 11, only the configuration for performing input current control and harmonic compensation is described, and the configuration for other control is omitted. Although not shown, the control circuit 7 also includes a DC voltage controller and a reactive power controller, and also controls the output voltage and the input reactive power. The control method performed by the control circuit 7 is not limited to this. For example, when the converter circuit 9 is a current-type converter circuit, output current control may be performed instead of output voltage control.

  Also in this embodiment, the same effect as in the case of the first embodiment can be obtained. That is, in the single-phase PWM converter system C, harmonic compensation for suppressing harmonics of the input current is required, but in the present embodiment, the control system for current control and each harmonic compensation are used. By using the control system as one control system and making the control system a linear time-invariant process, the control system design can be facilitated using linear control theory. Further, since it is not necessary to generate a signal whose phase is delayed by 90 degrees from the current signal I, there is no trouble of designing the order n of the FIR filter by providing the Hilbert transform unit 73 shown in FIG. Therefore, the configuration is simplified.

  The configuration of the single-phase PWM converter system C is not limited to the above. For example, the control circuit 8 may be used instead of the control circuit 7. Alternatively, an inverter circuit may be provided on the output side of the converter circuit 9 so as to convert the DC power into AC power and supply the AC load to the AC load.

  The control circuit according to the present invention, the grid interconnection inverter system using the control circuit, and the single-phase PWM converter system are not limited to the above-described embodiments. The specific configuration of each part of the control circuit according to the present invention, the grid-connected inverter system using this control circuit, and the single-phase PWM converter system can be variously modified.

A Grid-connected inverter system A 'Inverter system 1 DC power supply 2 Inverter circuit (power conversion circuit)
3 Filter circuit 4 Transformer circuit 5 Current sensor 6 Voltage sensor 7 and 8 Control circuit 72 System counter component generation unit 74 Current controller (control means)
84 Voltage controller (control means)
77 PWM signal generator 9 Converter circuit (power conversion circuit)
B Power system C Single-phase PWM converter system L, L 'Load

Claims (7)

A control circuit for controlling the driving of a plurality of switching means in a power conversion circuit for single-phase alternating current by means of a PWM signal,
Deviation signal generating means for generating a deviation signal that is a deviation between a signal based on the output or input of the power conversion circuit and a target value of the fundamental wave component;
Control means for controlling a fundamental wave component included in the deviation signal to zero and generating a correction value signal for suppressing and controlling a predetermined harmonic component included in the deviation signal;
PWM signal generating means for generating a PWM signal based on the correction value signal;
With
The control means generates the correction value signal by signal processing the deviation signal by a transfer function designed by a robust control design method using the frequency weight W,
When the angular frequency of the fundamental wave of the single-phase alternating current is set to ω0 and the nth harmonic is suppressed, the denominator of the frequency weight W is (s ² + ω ₀ ² ) and {s ² + (n Ω ₀ ) ² } is multiplied, and the numerator is not multiplied by (s ² + ω ₀ ² ) and {s ² + (n · ω ₀ ) ² } corresponding to each n ,
A control circuit characterized by that. The frequency weight W is a predetermined real number k,
The control circuit according to claim 1, wherein   The control circuit according to claim 1, wherein the signal based on the output current or the input current is a signal detected.   The control circuit according to claim 1, wherein the signal based on the output voltage or the input voltage is a signal detected.   The control circuit according to claim 1, wherein the transfer function is designed by a H∞ loop shaping method using a frequency weight W.   A grid-connected inverter system comprising an inverter circuit and the control circuit according to any one of claims 1 to 5.   A single-phase PWM converter system comprising a converter circuit and the control circuit according to claim 1.