JP5865090B2 - Control circuit for power conversion circuit, grid-connected inverter system using this control circuit, and single-phase PWM converter system - Google Patents

Description

  The present invention relates to a control circuit for controlling the output or input of a power conversion circuit, a grid-connected inverter system and a single-phase PWM converter system using the control circuit, and more particularly to a circuit for compensating for harmonics.

  2. Description of the Related Art Conventionally, a grid-connected inverter system has been developed that converts DC power generated by a solar cell or the like into AC power and supplies it to an electric power system.

  FIG. 13 is a block diagram for explaining a conventional grid-connected inverter system of a general single-phase power system. The grid interconnection inverter system A100 converts the power generated by the DC power supply 1 and supplies it to a single-phase power system B (hereinafter referred to as “power system B”).

  The inverter circuit 2 converts a DC voltage input from the DC power supply 1 into an AC voltage by switching a switching element (not shown). The filter circuit 3 removes a switching frequency component included in the AC voltage output from the inverter circuit 2. The transformer circuit 4 boosts (or steps down) the AC voltage output from the filter circuit 3 to the system voltage of the power system B. The control circuit 700 receives the current signal and the voltage signal detected by the current sensor 5 and the voltage sensor 6, generates a PWM signal based on the current signal and the voltage signal, and outputs the PWM signal to the inverter circuit 2. The inverter circuit 2 performs switching of the switching element based on the PWM signal input from the control circuit 700.

  FIG. 14 is a block diagram for explaining the internal configuration of the control circuit 700.

  The current signal input from the current sensor 5 is input to the Hilbert conversion unit 73 and the rotation coordinate conversion unit 78.

The Hilbert transform unit 73 delays the phase of the input current signal by π / 2 (90 degrees) by the Hilbert transform. An ideal Hilbert transform is represented by a transfer function H (ω) shown in the following equation (1). Where ω _S is the sampling angular frequency and j is the imaginary unit. That is, the Hilbert transform is a filter process in which the amplitude characteristic is constant regardless of the frequency and the phase characteristic is delayed by π / 2 (90 degrees) in a positive / negative frequency region. Since ideal Hilbert transform cannot be realized, it is approximately realized as, for example, a FIR (Finite impulse response) filter.

  The Hilbert transformed signal is obtained by delaying the phase of the original signal by π / 2. Therefore, since the so-called rotational coordinate conversion process (dq conversion process) can be performed by using the signal after the Hilbert transform and the original signal, the rotation can be performed even in the case of single-phase alternating current as in the case of three-phase alternating current. Control in the coordinate system can be performed.

  The rotation coordinate conversion unit 78 includes a current signal input from the current sensor 5 (hereinafter referred to as “α-axis current signal Iα”) and a current signal input from the Hilbert conversion unit 73 (hereinafter referred to as “β-axis current signal Iβ”). Is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system. The rotating coordinate system is an orthogonal coordinate system having orthogonal d-axis and q-axis and rotating in the same rotational direction at the same angular velocity as the fundamental wave of the system voltage of the power system B. As a concept opposite to the rotating coordinate system, a non-rotating coordinate system is a stationary coordinate system. The rotation coordinate conversion unit 78 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 78 is expressed by a determinant represented by the following expression (2).

  The LPF 74a and the LPF 75a are low-pass filters and pass only the DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. Through the rotation coordinate conversion process, the fundamental wave components of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. The PI control unit 74b and the PI control unit 75b perform PI control (proportional integration control) based on the deviation between the DC component of the d-axis current signal Id and the q-axis current signal Iq and the target value, respectively, and the fundamental compensation signal Xd and Xq are output. Since a DC component can be used as the target value, the PI control unit 74b and the PI control unit 75b can perform control with high accuracy.

  The stationary coordinate conversion unit 79 converts the fundamental wave compensation signals Xd and Xq input from the PI control unit 74b and the PI control unit 75b, respectively, into two fundamental wave compensation signals Xα and Xβ in the stationary coordinate system. The rotation coordinate conversion unit 78 performs a reverse conversion process. The stationary coordinate conversion unit 79 performs a so-called stationary coordinate conversion process (inverse dq conversion process), and the fundamental wave compensation signals Xd and Xq of the rotating coordinate system are converted to the fundamental wave compensation of the stationary coordinate system based on the phase θ. Converted to signals Xα and Xβ.

The conversion process performed by the stationary coordinate conversion unit 79 is expressed by a determinant represented by the following expression (3).

  The control circuit 700 has a function of suppressing harmonics input from the power system B and harmonics output from the inverter circuit 2. The harmonic compensation controller 800 extracts a harmonic component from the current signal input from the current sensor 5 and outputs a harmonic compensation signal for outputting a harmonic that cancels the harmonic component. The grid interconnection inverter system A100 outputs harmonics based on the harmonic compensation signal (that is, harmonics having a phase opposite to the detected harmonics) and cancels the harmonics, thereby suppressing harmonics.

  FIG. 15 is a block diagram for explaining the internal configuration of the harmonic compensation controller 800. In general, the harmonics from the power system B or the inverter circuit 2 are mostly the fifth harmonic, the seventh harmonic, and the eleventh harmonic. In order to suppress these harmonics, a fifth harmonic compensator 810 for suppressing the fifth harmonic, a seventh harmonic compensator 820 for suppressing the seventh harmonic, and an eleventh harmonic are provided. An eleventh harmonic compensation unit 830 for suppression is provided in the harmonic compensation controller 800. The fifth harmonic compensation unit 810 includes a rotation coordinate conversion unit 811, LPFs 812 and 813, I control units 814 and 815, and a stationary coordinate conversion unit 816. Since the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 have the same configuration as the fifth harmonic compensation unit 810, the description and description in FIG. 15 are omitted.

The rotation coordinate conversion unit 811 converts the α-axis current signal Iα input from the current sensor 5 and the β-axis current signal Iβ input from the Hilbert conversion unit 73 into a d-axis current signal Id ₅ and a q-axis current signal in the rotation coordinate system. Iq ₅ is converted. This rotating coordinate system is an orthogonal coordinate system that rotates in the opposite direction at an angular velocity that is five times the angular velocity of the fundamental wave of the system voltage. The rotation coordinate conversion unit 811 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 811 is represented by a determinant represented by the following expression (4).
Note that the rotational coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (4).

LPF 812 and LPF 813 are low-pass filters and pass only the DC components of d-axis current signal Id ₅ and q-axis current signal Iq ₅ , respectively. By the rotational coordinate conversion process, the fifth harmonics of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively. The I control unit 814 and the I control unit 815 perform I control (integration control) based on the DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively, and the fifth harmonic compensation signals Yd ₅ , Yq ₅ is output. Since a DC component can be used as the target value, the I control unit 814 and the I control unit 815 can perform highly accurate control.

The stationary coordinate conversion unit 816 converts the fifth harmonic compensation signals Yd ₅ and Yq ₅ input from the I control unit 814 and the I control unit 815, respectively, into two fifth harmonic compensation signals Yα ₅ and Yβ in the stationary coordinate system. ₅ is converted, and the conversion process reverse to that of the rotating coordinate conversion unit 811 is performed. The static coordinate conversion unit 816 performs so-called static coordinate conversion processing (inverse dq conversion processing), and converts the fifth-order harmonic compensation signals Yd ₅ and Yq ₅ of the rotating coordinate system into a static coordinate system based on the phase θ. of the fifth harmonic compensation signal Yarufa _5, converted into Ybeta _5.

The conversion process performed by the stationary coordinate conversion unit 816 is expressed by a determinant represented by the following expression (5).
Note that the stationary coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (5).

Similarly, the seventh harmonic compensation unit 820 generates _seventh harmonic compensation signals Yα ₇ and Yβ ₇ , and the eleventh harmonic compensation unit 830 generates _eleventh harmonic compensation signals Yα ₁₁ and Yβ ₁₁ . A harmonic compensation signal Yα obtained by adding the fifth harmonic compensation signal Yα ₅ , the seventh harmonic compensation signal Yα ₇ , and the eleventh harmonic compensation signal Yα ₁₁ is output from the harmonic compensation controller 800.

  The PWM signal generation unit 77 is based on a correction value signal X′α obtained by adding the fundamental wave compensation signal Xα output from the stationary coordinate conversion unit 79 and the harmonic compensation signal Yα output from the harmonic compensation controller 800. Generate and output a PWM signal.

Japanese Patent Laid-Open No. 2003-143860 Japanese Patent No. 4421700

"Control method of single-phase grid-connected inverter using Hilbert transform" Electron Theory D, Vol. 121, No. 10, 2001

  Since the Hilbert transform unit 73 cannot realize an ideal Hilbert transform, the Hilbert transform unit 73 is configured by, for example, an FIR filter to realize an approximate Hilbert transform. Since a group delay occurs in the FIR filter, the β-axis current signal Iβ output from the Hilbert transform unit 73 is delayed with respect to the α-axis current signal Iα. For this reason, although not shown in FIG. 14, in actual processing, it is necessary to provide a delay circuit between the current sensor 5 and the rotation coordinate conversion unit 78 to adjust the phase. The FIR filter constituting the Hilbert transform unit 73 has a characteristic that if the order n is increased to widen the conversion band, the ripple in the frequency characteristics can be reduced, but the group delay is increased and the detection is delayed. On the contrary, if the order n is reduced to narrow the conversion band, the group delay can be suppressed, but the ripple in the frequency characteristic becomes large and the detection accuracy is lowered.

  Therefore, when the Hilbert transform unit 73 is used, there is an inconvenience that the order n of the FIR filter to be configured must be designed in consideration of the tradeoff between the detection speed and the detection accuracy. In addition, in order to obtain a certain degree of detection accuracy, it is necessary to increase the order n of the FIR filter. As a result, the configuration of the Hilbert transform unit 73 becomes large, which causes a problem that the control circuit 700 is complicated.

  In addition, there is a problem that a great effort is required to design a control system for each harmonic compensation. In order to design a control system for each harmonic compensation, it is necessary to optimally design the parameters of the LPFs 812 and 813 and the integral gains of the I controllers 814 and 815. However, since the rotating coordinate conversion unit 811 and the stationary coordinate conversion unit 816 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  The present invention has been conceived under the circumstances described above, and is capable of performing high-speed and high-accuracy harmonic suppression control with a simple configuration and performing processing having linearity and time invariance. Its purpose is to provide a circuit.

  In order to solve the above problems, the present invention takes the following technical means.

A control circuit provided by the first aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to single-phase alternating current using a PWM signal, and outputs or inputs of the power conversion circuit PWM signal generating means for generating a PWM signal based on a predetermined harmonics compensation means for generating a harmonic compensation signal line Utame control to suppress a harmonic component, the harmonic compensation signal included in the signal based on the The harmonic compensation means performs signal processing on the signal based on the transfer function G (s), and further performs phase adjustment processing to generate the harmonic compensation signal, and the transmission The function G (s) is an n-th order harmonic where F (s) is a one-input one-output transfer function having an impulse response f (t), the angular frequency of the fundamental wave of the single-phase alternating current is ω ₀ , and the imaginary unit is j. Suppress the wave If you want,
It is characterized by being.

  In a preferred embodiment of the present invention, control means for generating a fundamental wave compensation signal by performing control for causing the fundamental wave component included in the signal based on the target value to follow a target value, and the fundamental wave compensation signal and the harmonic wave are generated. Correction value signal generation means for generating a correction value signal by adding the compensation signal is further provided, and the PWM signal generation means generates a PWM signal based on the correction value signal.

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _I / s (where K _I is an integral gain).

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _P + K _I / s (where K _P and K _I are proportional gain and integral gain, respectively). is there.

In a preferred embodiment of the present invention, the transfer function representing the predetermined control process is F (s) = K _P + K _I / s + K _D · s (where K _P , K _I and K _D are proportional to each other) Gain, integral gain, differential gain).

  In a preferred embodiment of the present invention, the signal based on the output current or the input current is a signal detected.

  In a preferred embodiment of the present invention, the signal based on the output voltage or the input voltage is detected.

  In a preferred embodiment of the present invention, the control system is designed using a robust control design.

  In a preferred embodiment of the present invention, the control system is designed using the H∞ loop shaping method.

  In a preferred embodiment of the present invention, based on the harmonic compensation signal output from the harmonic compensation means, divergence determining means for determining that the control for suppressing the harmonic component tends to diverge, When the divergence determining means determines that there is a divergence tendency, it further comprises output stop means for stopping the output of the harmonic compensation signal.

  In a preferred embodiment of the present invention, based on the harmonic compensation signal output from the harmonic compensation means, divergence determining means for determining that the control for suppressing the harmonic component tends to diverge, When it is determined by the divergence determining means that there is a tendency to diverge, it further comprises phase changing means for changing the phase of the harmonic compensation signal to a phase where control does not diverge.

  In a preferred embodiment of the present invention, the divergence determining means determines that the harmonic compensation signal has a divergence tendency when the harmonic compensation signal exceeds a predetermined threshold.

  The grid interconnection inverter system provided by the second aspect of the present invention is a grid interconnection inverter system including an inverter circuit and a control circuit provided by the first aspect of the present invention. To do.

  The single-phase PWM converter system provided by the third aspect of the present invention is a single-phase PWM converter system including a converter circuit and a control circuit provided by the first aspect of the present invention. To do.

  According to the present invention, since it is not necessary to generate a signal whose phase is delayed by 90 degrees from a signal based on an output or an input, there is no inconvenience of designing an FIR filter for Hilbert transform processing. Further, the configuration can be simplified.

Further, according to the present invention, a signal based on the transfer function G (s) is subjected to signal processing, and further, phase adjustment processing is performed to perform control and generate a harmonic compensation signal. The signal processing based on the transfer function G (s) is the same processing as that in which the harmonic compensation signal generated by performing the predetermined control processing after performing the rotational coordinate transformation is subjected to the stationary coordinate transformation. Therefore, high-speed and high-accuracy harmonic suppression control can be performed with a simple configuration. Moreover, the signal processing by the transfer function G (s) has linearity and time invariance. Therefore, a design method based on the linear control theory can be used, and the control system can be easily designed. System analysis can also be performed.

  Other features and advantages of the present invention will become more apparent from the detailed description given below with reference to the accompanying drawings.

It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process. It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process, and is represented by the matrix. It is a block diagram for demonstrating the calculation of a matrix. It is a block diagram which shows the process which performs static coordinate transformation after performing PI control after performing rotational coordinate transformation. It is a block diagram which shows the process which performs a stationary coordinate transformation after performing I control after performing a rotational coordinate transformation. Is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. It is a block diagram for demonstrating the grid connection inverter system which concerns on 1st Embodiment. It is a block diagram for demonstrating the control circuit which concerns on 2nd Embodiment. It is a Bode diagram which shows an example of the transfer function before and behind interconnection. It is a figure for demonstrating the harmonic compensation controller which concerns on 3rd Embodiment. It is a figure for demonstrating the other Example of the harmonic compensation controller which concerns on 3rd Embodiment. It is a block diagram for demonstrating the single phase PWM converter system which concerns on 4th Embodiment. It is a block diagram for demonstrating the grid connection inverter system of the conventional common single phase electric power system. It is a block diagram for demonstrating the internal structure of a control circuit. It is a block diagram for demonstrating the internal structure of a harmonic compensation controller.

  Embodiments of the present invention will be specifically described below with reference to the drawings.

  First, a method for converting processing involving rotational coordinate transformation and stationary coordinate transformation into linear time-invariant processing will be described.

  Fig.1 (a) is a figure for demonstrating the process accompanied by rotation coordinate transformation and stationary coordinate transformation. In this processing, first, the signals α and β are converted into signals d and q by rotational coordinate conversion. The signals d and q are each processed by a predetermined transfer function F (s), and signals d 'and q' are output. Next, the signals d ′ and q ′ are converted into signals α ′ and β ′ by stationary coordinate transformation. The nonlinear time-varying process shown in FIG. 1A is converted into a process using a matrix G of linear time-invariant transfer functions shown in FIG.

The rotational coordinate transformation shown in FIG. 1A is represented by a determinant of the following equation (6), and the stationary coordinate transformation is represented by a determinant of the following equation (7).

  Therefore, the process shown in FIG. 1A can be expressed as shown in FIG. 2A using a matrix. The matrix G shown in FIG. 1B can be calculated by calculating the product of the three matrices shown in FIG. 2A and making the calculated matrix a linear time-invariant matrix. At this time, the calculation is performed after converting the matrix of the stationary coordinate conversion and the rotation coordinate conversion into a matrix product.

The matrix of rotational coordinate conversion can be converted into the product of the right-hand side matrix shown in the following equation (8).
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

Moreover, the matrix of a static coordinate transformation | transformation can be converted into the product of the matrix of the right side shown in following (9) Formula. The matrix in the middle of the matrix product is a linear time-invariant matrix.
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

When the product of the three matrices shown in FIG. 2A is calculated using the above equations (8) and (9) and the matrix G is calculated, the following equation (10) is obtained.

If attention is paid to the element in the first row and the first column of the three matrixes in the center of the above equation (10) and this is expressed in a block diagram, the block diagram shown in FIG. 3 is obtained. When calculating the input / output characteristics of the block diagram shown in FIG.
It becomes. Where F (s) is a one-input one-output transfer function having an impulse response f (t).

Here, when θ (t) = ω ₀ t, θ (t) −θ (τ) = ω ₀ t−ω ₀ τ = ω ₀ (t−τ) = θ (t−τ). The input / output characteristics of the block diagram shown in FIG. 3 are equal to those of a linear time invariant system having an impulse response f (t) exp (−jω ₀ t). When the impulse response f (t) exp (−jω ₀ t) is Laplace transformed, a transfer function F (s + jω ₀ ) is obtained. The input / output characteristics when exp (jθ (t)) and exp (−jθ (t)) in the block diagram shown in FIG. 3 are interchanged are the input / output characteristics of the transfer function F (s−jω ₀ ). become.

Therefore, if the calculation is further advanced from the above equation (10),
Is calculated.

  Thereby, the process shown in FIG. 2A can be converted into the process shown in FIG. The process shown in FIG. 2B is equivalent to the process of performing the stationary coordinate conversion after performing the process represented by the predetermined transfer function F (s) after performing the rotational coordinate conversion. This system is a linear time-invariant system.

The transfer function of the PI control (proportional integral control) controller is expressed as F (s) = K _P + K _I / s, where K _P and K _I are the proportional gain and integral gain, respectively. Therefore, the transfer function matrix G _PI showing the process shown in FIG. 4, that is, the process equivalent to the process of performing the static coordinate conversion after performing the PI control after performing the rotational coordinate conversion, uses the above equation (11). Then, it is calculated as in the following equation (12).

Further, the transfer function of the I control (integral control) controller, the integral gain and K _I, represented by _{F (s) = K I /} s. Therefore, the processing shown in FIG. 5, i.e., the matrix G _I of the transfer function from performing a rotational coordinate transformation showing the processing of the processing equivalent to a still coordinate transformation after the I control, using the above equation (11) Thus, the following equation (13) is calculated.

Figure 6 is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. FIG (a) is first row and the first column elements (hereinafter, the same applies for. Other elements described as "(1,1) element".) The matrix G _I and (2,2) element of FIG. 4B shows the transfer function of the (1,2) element of the matrix G _I , and FIG. 4C shows the transfer function of the (2,1) element of the matrix G _I. Is shown. The figure shows the case where the frequency of the fundamental wave of the system voltage (hereinafter referred to as “center frequency”. Also, the angular frequency corresponding to the center frequency is referred to as “center angular frequency”) is 60 Hz (that is, angular frequency). ω ₀ = 120π), and the integral gain K _I is “0.1”, “1”, “10”, “100”.

FIG (a), the amplitude characteristic shown by (b) and (c) are all, there is a peak in the center frequency, the integral gain K _I is increased, the amplitude characteristic is increased. Further, the phase characteristic shown in FIG. 5A is 0 degree at the center frequency. In other words, the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without the signal of the center frequency (center angular frequency) changes the phase. The phase characteristic shown in FIG. 5B is 90 degrees at the center frequency. In other words, the transfer function of the (1,2) element of the matrix G _I causes the phase of the signal of the center frequency (center angular frequency) passes advancing 90 degrees. On the other hand, the phase characteristic shown in FIG. 4C is −90 degrees at the center frequency. In other words, the transfer function of (2,1) element of the matrix G _I is the phase of the signal of the center frequency (center angular frequency) passing delayed 90 degrees.

Consider a case where the phases are treated with 90-degree different signals α and the signal beta (the phase of the signal α is advanced 90 degrees from the signal beta.) The matrix G _I. Even if the processing shown in the transfer function of (1, 1) element of the matrix G _I in signal alpha, phase does not change (see FIG. 6 (a)). Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I in signal beta, advances the phase by 90 degrees (see Figure 6 (b)). Therefore, since the phase of both is the same as that of the signal α, the two are added together to strengthen each other. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I in signal alpha, phase is delayed 90 degrees (see Figure 6 (c)). Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in signal beta, phase does not change. Therefore, since both phases are the same as the signal β, they are strengthened by adding both.

The process shown in the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without changing the phase (see FIG. 6 (a)). Therefore, even if the matrix G ′ _I of the transfer function shown in the following equation (14) in which the (1,2) and (2,1) elements of the matrix G _I shown in the above equation (13) are set to “0” is used. A similar process can be performed. In the case of the following equation (14), since the (13) minutes is not constructive as compared with the case of expression, it is necessary to design the integral gain K _I in correspondingly large value.

That is, if the signal α and the signal β are processed by the transfer function G _I (s) shown in the following equation (15), the stationary coordinate conversion is performed after performing the I control after performing the rotational coordinate conversion. The same processing is performed. In the case of a single-phase system, it is only necessary to perform processing on the signal α, so that it is not necessary to generate the signal β whose phase is delayed by 90 degrees.

  Next, a method for controlling harmonic components will be described.

The transfer function G _I (s) shown in the above equation (15) is for controlling the fundamental wave component. The n-th harmonic is an angular frequency component obtained by multiplying the angular frequency of the fundamental wave by n. Therefore, the transfer function in the case of controlling the nth harmonic is the transfer function G _In (s) shown in the following equation (16) where ω ₀ is n · ω ₀ in the above equation (15).

Hereinafter, a case where the harmonic compensation controller that performs the processing represented by the transfer function G _In (s) in the above equation (16) is applied to the control circuit of the grid-connected inverter system will be described as a first embodiment of the present invention. explain.

  FIG. 7 is a block diagram for explaining the grid-connected inverter system according to the first embodiment.

  As shown in the figure, the grid-connected inverter system A includes a DC power source 1, an inverter circuit 2, a filter circuit 3, a transformer circuit 4, a current sensor 5, a voltage sensor 6, and a control circuit 7.

  The DC power source 1 is connected to the inverter circuit 2. The inverter circuit 2, the filter circuit 3, and the transformer circuit 4 are connected in series to the output line in this order, and are connected to the single-phase AC power system B. The current sensor 5 and the voltage sensor 6 are installed on the output side of the transformer circuit 4. The control circuit 7 is connected to the inverter circuit 2. The grid interconnection inverter system A converts the DC power output from the DC power supply 1 into AC power and supplies it to the power grid B. In addition, the structure of the grid connection inverter system A is not restricted to this. For example, the current sensor 5 and the voltage sensor 6 may be provided on the input side of the transformer circuit 4, or other sensors necessary for controlling the inverter circuit 2 may be provided. Further, the transformer circuit 4 may be provided on the input side of the filter circuit 3, or a so-called transformer-less system in which the transformer circuit 4 is not provided. Further, a DC / DC converter circuit may be provided between the DC power supply 1 and the inverter circuit 2.

  The DC power source 1 outputs DC power and includes, for example, a solar battery. A solar cell generates direct-current power by converting solar energy into electrical energy. The DC power source 1 outputs the generated DC power to the inverter circuit 2. Note that the DC power source 1 is not limited to one that generates DC power from a solar cell. For example, the DC power source 1 may be a fuel cell, a storage battery, an electric double layer capacitor, a lithium ion battery, or AC power generated by a diesel engine generator, a micro gas turbine generator, a wind turbine generator, or the like. It may be a device that converts to DC power and outputs it.

  The inverter circuit 2 converts a DC voltage input from the DC power source 1 into an AC voltage and outputs the AC voltage to the filter circuit 3. The inverter circuit 2 is a single-phase inverter and is a PWM control type inverter circuit provided with two sets of four switching elements (not shown). The inverter circuit 2 converts the DC voltage input from the DC power source 1 into an AC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. In addition, the inverter circuit 2 is not limited to this, For example, a multilevel inverter may be sufficient.

  The filter circuit 3 removes high frequency components due to switching from the AC voltage input from the inverter circuit 2. The filter circuit 3 includes a low pass filter including a reactor and a capacitor. The AC voltage from which the high frequency component has been removed by the filter circuit 3 is output to the transformer circuit 4. The configuration of the filter circuit 3 is not limited to this, and any known filter circuit for removing high frequency components may be used. The transformer circuit 4 boosts or lowers the AC voltage output from the filter circuit 3 to a level substantially the same as the system voltage.

  The current sensor 5 detects an alternating current output from the transformer circuit 4 (that is, an output current of the grid interconnection inverter system A). The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects the system voltage of the power system B. The detected voltage signal V is input to the control circuit 7. Note that the output voltage output by the grid interconnection inverter system A substantially matches the grid voltage.

  The control circuit 7 controls the inverter circuit 2 and is realized by, for example, a microcomputer. The control circuit 7 generates a PWM signal based on the current signal I input from the current sensor 5 and the voltage signal V input from the voltage sensor 6 and outputs the PWM signal to the inverter circuit 2. The control circuit 7 generates a command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A based on the detection signal input from each sensor, and is generated based on the command value signal. Output a pulse signal as a PWM signal. The inverter circuit 2 outputs an alternating voltage having a waveform corresponding to the command value signal by switching each switching element on and off based on the input PWM signal. The control circuit 7 controls the output current by changing the waveform of the command value signal to change the waveform of the output voltage of the grid interconnection inverter system A. Thereby, the control circuit 7 performs various feedback controls. In addition, the control circuit 7 suppresses harmonics by causing the inverter circuit 2 to output harmonics for canceling the harmonics input from the power system B.

  In FIG. 7, only the configuration for performing the output current control and the harmonic suppression control is described, and the configuration for the other control is omitted. Actually, the control circuit 7 performs DC voltage control (feedback control performed so that the input DC voltage becomes a preset voltage target value) and reactive power control (reactive power target value with preset output reactive power) (Feedback control to be performed) is also performed. The control method performed by the control circuit 7 is not limited to this. For example, output voltage control or active power control may be performed.

  The control circuit 7 includes a system counter component generation unit 72, a current controller 74, a PWM signal generation unit 77, and a harmonic compensation controller 8.

  The system counter-part generating unit 72 receives the voltage signal V from the voltage sensor 6 and generates and outputs a system command value signal K. The system command value signal K serves as a reference of the command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A, and the system command value signal K is corrected by a correction value signal X ′ described later. As a result, a command value signal is generated.

The current controller 74 receives a deviation between the current signal I detected by the current sensor 5 and the current target value, and generates a fundamental wave compensation signal X for current control. The current controller 74 performs processing represented by the transfer function G _I (s) in the above equation (13). That is, assuming that the deviation ΔI between the current signal I and the current target value is the process shown in the following equation (17). The angular frequency omega ₀ is the fundamental angular frequency of the system voltage _{(e.g., ω 0 = 120π [rad /} sec] (60 [Hz])) is set in advance, the integral gain K _I is pre-designed. Further, the current controller 74 performs a process for maximizing the stability margin, and the phase adjustment for correcting the phase delay in the control loop is also performed.

  In the present embodiment, a current target value on the α-axis generated by stationary coordinate conversion of the d-axis current target value and the q-axis current target value is used as the current target value. A correction value for DC voltage control (not shown) is used for the d-axis current target value, and a correction value for reactive power control (not shown) is used for the q-axis current target value. When the current target value on the α axis is directly given, the target value may be used as it is.

In the present embodiment, the current controller 74 is designed by the H∞ loop shaping method, which is one of linear control theory, using the transfer function G _I (s) as the frequency weight. Since the process performed by the current controller 74 is indicated by the transfer function G _I (s), it is a linear time-invariant process. Therefore, a control system design using linear control theory can be performed.

The current controller 74 is required to have a design specification that the output current follows the sine wave target value and that the output is returned to a predetermined ratio within a predetermined time (speed response) at the time of a sag. In order for the output of the system to completely follow a certain target value, the closed-loop system must have the same pole as the target generator, and the closed-loop system must be asymptotically stable (internal model principle). The pole of the sine wave target value is ± jωo, and the pole of the term 1 / (s ² + ω ₀ ² ) included in the transfer function G _I (s) is also ± jωo. Therefore, the poles of the closed loop system and the target generator are the same. In addition, if the H∞ loop shaping method is used, a controller in which the closed loop system becomes asymptotically stable can be designed. Therefore, by designing using the H∞ loop shaping method so as to satisfy the condition of rapid response, it is possible to easily design the most stable control system that meets the design specifications.

  Note that the design method used for designing the control system is not limited to this, and other linear control theories can also be used. For example, the design may be performed using a loop shaping method, optimal control, H∞ control, a mixed sensitivity problem, or the like.

  The harmonic compensation controller 8 receives the current signal I detected by the current sensor 5 and generates a harmonic compensation signal Y for harmonic suppression control. The harmonic compensation controller 8 suppresses the fifth harmonic compensation unit 81 for suppressing the fifth harmonic, the seventh harmonic compensation unit 82 for suppressing the seventh harmonic, and the eleventh harmonic. The 11th harmonic compensation part 83 for this is provided.

The fifth harmonic compensation unit 81 is for suppressing the fifth harmonic. The fifth-order harmonic compensator 81 is represented by a transfer function G _I5 (s) for controlling the fifth-order harmonic with n = 5 in the transfer function G _In (s) shown in the above equation (16). Process. That is, the fifth-order harmonic compensator 81 performs the processing shown in the following equation (18) and outputs the fifth-order harmonic compensation signal Y ₅ . As the angular frequency ω _{0, the} angular frequency of the fundamental wave of the system voltage (for example, ω ₀ = 120π [rad / sec] (60 [Hz])) is set in advance, and the integral gain K _I5 is designed in advance. The fifth-order harmonic compensator 81 also performs processing for maximizing the stability margin. Among these, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken.

In the present embodiment, the fifth-order harmonic compensator 81 is designed by the H∞ loop shaping method, which is one of linear control theories, using the transfer function G _I5 (s) as the frequency weight. Since the process performed by the fifth harmonic compensation unit 81 is indicated by the transfer function G _I5 (s), it is a linear time-invariant process. Therefore, a control system design using linear control theory can be performed.

Note that the design method used for designing the control system is not limited to this, and other linear control theories can also be used. For example, the design may be performed using a loop shaping method, optimal control, H∞ control, a mixed sensitivity problem, or the like. Alternatively, the phase θ ₅ for adjustment from the phase delay may be calculated and set in advance. For example, if the phase is delayed by 90 degrees in the controlled object, θ ₅ = −90 degrees may be set in order to delay the phase by 180 degrees.

The seventh harmonic compensation unit 82 is for suppressing the seventh harmonic. The seventh-order harmonic compensator 82 is represented by the transfer function G _I7 (s) for controlling the seventh-order harmonic with n = 7 in the transfer function G _In (s) shown in the above equation (16). Process. That is, the seventh-order harmonic compensation unit 82 performs processing shown in the following equation (19) and outputs the seventh-order harmonic compensation signal Y ₇ . As the angular frequency ω _0, the angular frequency of the fundamental wave of the system voltage is set in advance, and the integral gain K _I7 is designed in advance. The seventh harmonic compensation unit 82 also performs processing for maximizing the stability margin, and in this process, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken. The seventh harmonic compensation unit 82 is also designed in the same manner as the fifth harmonic compensation unit 81.

The 11th harmonic compensator 83 is for suppressing the 11th harmonic. The eleventh-order harmonic compensator 83 is represented by the transfer function G _I11 (s) for controlling the eleventh-order harmonic with n = 11 in the transfer function G _In (s) shown in the above equation (16). Process. That is, the 11th-order harmonic compensation unit 83 performs the processing shown in the following equation (20) and outputs the 11th-order harmonic compensation signal Y ₁₁ . As the angular frequency ω _0, the angular frequency of the fundamental wave of the system voltage is set in advance, and the integral gain K _I11 is designed in advance. The eleventh harmonic compensation unit 83 also performs processing for maximizing the stability margin, and in this process, the phase adjustment for correcting the phase delay in the control loop to make the phase opposite is also performed. It has been broken. The eleventh harmonic compensation unit 83 is also designed in the same manner as the fifth harmonic compensation unit 81.

The fifth harmonic compensation signal Y ₅ output from the fifth harmonic compensation unit 81, the seventh harmonic compensation signal Y ₇ output from the seventh harmonic compensation unit 82, and the eleventh harmonic compensation unit 83 output The eleventh harmonic compensation signal Y ₁₁ is added and output from the harmonic compensation controller 8 as the harmonic compensation signal Y. In the present embodiment, the case where the harmonic compensation controller 8 includes the fifth-order harmonic compensation unit 81, the seventh-order harmonic compensation unit 82, and the eleventh-order harmonic compensation unit 83 has been described. Not limited. The harmonic compensation controller 8 may be designed according to the harmonic order that needs to be suppressed. For example, when it is desired to suppress only the fifth harmonic, only the fifth harmonic compensator 81 needs to be provided. If it is desired to further suppress the 13th harmonic, the process represented by the transfer function G _I13 (s) with n = 13 is performed in the transfer function G _In (s) shown in the above equation (16) 13. A second harmonic compensator may be further provided.

  The fundamental wave compensation signal X output from the current controller 74 and the harmonic compensation signal Y output from the harmonic compensation controller 8 are added to calculate a correction value signal X ′. Then, the system command value signal K output from the system counter generation unit 72 is added to the correction value signal X ′ to calculate the command value signal X ″ and input to the PWM signal generation unit 77.

  The PWM signal generation unit 77 is based on an input command value signal X ″ and a signal obtained by inverting the command value signal X ″ and a carrier signal generated as a triangular wave signal having a predetermined frequency (for example, 4 kHz), PWM signals Pp and Pn are generated by a triangular wave comparison method. In the triangular wave comparison method, for example, the command value signal X ″ is compared with the carrier signal, and the pulse signal that becomes high level when the command value signal X ″ is larger than the carrier signal and becomes low level when the command value signal X ″ is smaller is the PWM signal Pp. Is generated as Similarly, the inverted signal and the carrier signal are compared, and the PWM signal Pn is generated. The generated PWM signals Pp and Pn are output to the inverter circuit 2. The PWM signal generator 77 also outputs a signal obtained by inverting the PWM signals Pp and Pn to the inverter circuit 2.

  In the present embodiment, since the control circuit 7 does not need to generate a signal whose phase is delayed by 90 degrees from the current signal I, it is not necessary to provide the Hilbert transform unit 73 shown in FIG. Further, since only the current signal I needs to be processed (since it is not necessary to process a signal delayed by 90 degrees), the configuration is simplified. Furthermore, since the Hilbert transform unit 73 is unnecessary, there is no inconvenience of designing the order n of the FIR filter constituting the Hilbert transform unit 73.

In the present embodiment, the fifth-order harmonic compensator 81 performs control in the stationary coordinate system without performing rotational coordinate conversion and stationary coordinate conversion. As described above, the transfer function G _I5 (s) is a transfer function showing the same process as the process of performing the stationary coordinate conversion after performing the I control after performing the rotational coordinate conversion. Therefore, the fifth-order harmonic compensator 81 that performs the process represented by the transfer function G _I5 (s) is the same process as the rotation coordinate converter 811, the stationary coordinate converter 816, and the I controller 814 shown in FIG. It is carried out. Further, the Bode diagram of the transfer function G _I5 (s) is the same as the Bode diagram shown in FIG. _6A with the center frequency set to five times the fundamental frequency. That is, the fifth-order harmonic compensator 81 has a high gain only for the frequency component five times the fundamental frequency. Therefore, there is no need to provide the LPF 812 shown in FIG. Therefore, the fifth-order harmonic compensator 81 can perform high-speed and high-accuracy fifth-order harmonic suppression control with a simple configuration.

The process performed by the fifth-order harmonic compensator 81 is a linear time-invariant process because it is represented by the transfer function G _I5 (s). Further, the control loop for the fifth harmonic compensation does not include the rotation coordinate conversion process and the stationary coordinate conversion process, which are nonlinear time-varying processes, and thus is a linear time-invariant system. Therefore, control system design and system analysis using linear control theory are possible. As described above, by using the transfer function G _I5 (s), the nonlinear process of performing the static coordinate conversion after performing the I control after performing the rotational coordinate conversion is reduced to the linear input / output invariant multi-input / output system. This facilitates system analysis and control system design.

  The same applies to the seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83. The seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83 include the seventh harmonic compensation unit 820 and the seventh harmonic compensation unit 820 shown in FIG. The same processing as that of the 11th harmonic compensation unit 830 is performed. In addition, the processing performed by the seventh harmonic compensation unit 82 and the eleventh harmonic compensation unit 83 is also a linear time-invariant processing, which enables control system design and system analysis using linear control theory.

  In the above embodiment, the case where the fifth harmonic compensation unit 81, the seventh harmonic compensation unit 82, and the eleventh harmonic compensation unit 83 are individually designed has been described. However, the present invention is not limited to this. The fifth-order harmonic compensator 81, the seventh-order harmonic compensator 82, and the eleventh-order harmonic compensator 83 may be designed at a time so that the integral gain is shared.

In the first embodiment, the case where the fifth-order harmonic compensation unit 81, the seventh-order harmonic compensation unit 82, and the eleventh-order harmonic compensation unit 83 perform control in place of I control has been described, but the present invention is not limited to this. . For example, control in place of PI control may be performed. In the first embodiment, when the fifth harmonic compensator 81, the seventh harmonic compensator 82, and the eleventh harmonic compensator 83 perform control instead of PI control, the above equation (12) is used. In the transfer function G _PI (s) of the (1, 1) element of the transfer function matrix G _PI shown, ω ₀ is n · ω ₀ and the transfer function G _PIn (s ) May be used.

When the fifth harmonic compensator 81 performs control in place of PI control, the matrix G _PI5 with n = 5 in the above equation (21) may be used, and the seventh harmonic compensator 82 performs PI control. When performing alternative control, the matrix G _PI7 with n = 7 in the above equation (21) may be used. When the 11th harmonic compensator 83 performs control in place of PI control, In the equation (21), a matrix G _PI11 with n = 11 may be used.

When performing control in place of PI control, there is a merit that a damping effect at the time of transition can be added by adjusting the proportional gain K _P , but there is a demerit that it is easily affected by modeling errors. Conversely, when performing control in place of I control, there is a demerit that a damping effect during transition cannot be added, but there is a merit that it is less susceptible to modeling errors.

Note that the fifth harmonic compensation unit 81, the seventh harmonic compensation unit 82, and the eleventh harmonic compensation unit 83 may perform control in place of control other than I control and PI control. In the transfer function G _n (s) shown in the following equation (22) (transfer function in which ω ₀ is n · ω ₀ in the transfer function of the (1,1) element of the matrix G shown in the above equation (11)): By using the transfer function F (s) as the transfer function of each control, it is possible to calculate a transfer function indicating the same process as the process of performing the static coordinate conversion after performing the control after performing the rotational coordinate conversion. .

Therefore, PID control (the transfer function is expressed as F (s) = K _P + K _I / s + K _D · s where K _P is a proportional gain, K _{I is} an integral gain, and K _D is a differential gain). it can be made to perform control to replace, D control (differential control:. transfer function, the differential gain and K _D, represented by _{F (s) = K D ·} s), P control (proportional Control: The transfer function is expressed by F (s) = K _P, where the proportional gain is K _P ), and can be controlled in place of PD control, ID control, and the like.

  In the first embodiment, the case where the output current is controlled has been described, but the present invention is not limited to this. For example, the output voltage may be controlled. Hereinafter, the case of controlling the output voltage will be described as a second embodiment.

  FIG. 8 is a block diagram for explaining a control circuit according to the second embodiment. In the figure, the same or similar elements as those in the grid interconnection inverter system A shown in FIG.

  The inverter system A ′ shown in FIG. 8 is different from the grid-connected inverter system A (see FIG. 7) according to the first embodiment in that power is supplied to the load L instead of the power system B. Since it is necessary to control the voltage supplied to the load L, the control circuit 9 controls the output voltage, not the output current. The control circuit 9 is different from the control circuit 7 according to the first embodiment (see FIG. 7) in that a PWM signal is generated based on the voltage signal V input from the voltage sensor 6. The inverter system A ′ supplies power to the load L while controlling the output voltage to a target value by feedback control.

The voltage controller 94 receives a deviation between the voltage signal V detected by the voltage sensor 6 and the voltage target value, and generates a fundamental wave compensation signal X for voltage control. The voltage controller 94 performs processing represented by the transfer function G _I (s) in the above equation (13).

Also in this embodiment, the harmonic compensation controller 8 is provided. The fifth-order harmonic compensator 81 receives the voltage signal V detected by the voltage sensor 6 and changes the phase in order to reverse the process represented by the transfer function G _I5 (s) and the output harmonics. The adjustment process is performed, and the fifth-order harmonic compensation signal Y ₅ is output. Similarly, the seventh-order harmonic compensator 82 performs a process represented by the transfer function G _I7 (s) and a process for adjusting the phase so that the output harmonics have an opposite phase, and the seventh-order harmonics The 11th-order harmonic compensator 83 outputs a wave compensation signal Y ₇ , and performs a process represented by the transfer function G _I11 (s) and a process for adjusting the phase so that the output harmonics have an opposite phase. go and outputs the 11-order harmonic compensation signal Y _11.

  In this embodiment, the same effect as that of the first embodiment can be obtained. Even when the output voltage is controlled, each control method described in the first embodiment can be used. Moreover, you may make it switch control of output voltage and control of output current combining 1st Embodiment and 2nd Embodiment. In other words, current control is performed to supply power to the power system B when connected to the power system B, and voltage control is performed to supply power to the load L when not connected to the power system B. May be.

  Next, a method for preventing the harmonic suppression control from diverging will be described.

  In the first embodiment (see FIG. 7), the fifth-order harmonic compensator 81 adjusts the phase to correct the phase delay in the control loop so that the phase is reversed. The phase to be adjusted is set based on the impedance of the grid-connected inverter system A before being linked to the power grid B (mainly due to the inductance of the reactor of the filter circuit 3 and the capacitance of the capacitor). When the grid-connected inverter system A is linked to the power grid B, if the load condition of the power grid B is different from the assumption, the resonance point of the filter circuit 3 may be shifted or increased. In this case, the phase to be adjusted is not appropriate, and control may diverge.

  FIG. 9 shows a Bode diagram of a transfer function from the output voltage of the inverter circuit 2 to the output current of the grid-connected inverter system A, and transmission before the grid-connected inverter system A is linked to the power system B. An example of the transfer function after being linked to the function is shown.

When the angular frequency ω ₀ of the fundamental wave of the system voltage is 120π [rad / sec] (60 [Hz]), the angular frequency of the fifth harmonic is 600π (≈1885) [rad / sec] (300 [Hz]). become. According to the figure, the phase of the fifth harmonic is delayed about 90 degrees before the interconnection, but is delayed about 270 degrees after the interconnection. That is, even if the phase is adjusted so that the control is the negative feedback control before the connection, the control becomes the positive feedback control after the connection and the control is diverged. Below, the case where the structure for preventing the divergence of such control is provided is demonstrated as 3rd Embodiment.

  FIG. 10 is a diagram for explaining a harmonic compensation controller according to the third embodiment, and shows a fifth-order harmonic compensation unit 81 and a divergence prevention unit provided in the subsequent stage.

The divergence prevention unit 84 is for preventing divergence in the suppression control of the fifth harmonic. When the divergence prevention unit 84 determines that the control tends to diverge, the divergence prevention unit 84 changes the phase of the _fifth harmonic compensation signal Y ₅ input from the fifth harmonic compensation unit 81 and outputs it. The divergence prevention unit 84 includes a divergence determination unit 841 and a phase change unit 842.

The divergence determination unit 841 determines that the control tends to diverge. The divergence determination unit 841 compares the _fifth harmonic compensation signal Y ₅ input from the fifth harmonic compensation unit 81 with a predetermined threshold, and when the fifth harmonic compensation signal Y ₅ is larger than the predetermined threshold. It is determined that the control tends to diverge. The divergence determining unit 841 outputs a determination signal to the phase changing unit 842 when it is determined that the control tends to diverge. Note that the method of determining the divergence tendency by the divergence determining unit 841 is not limited to this. For example, when the fifth harmonic compensation signal Y ₅ is larger than a predetermined threshold for a predetermined time, it may be determined that the control tends to diverge. Alternatively, the maximum value of the fifth-order harmonic compensation signal Y ₅ may be always stored, and it may be determined that there is a divergence tendency based on the gradient of the maximum value.

The phase changing unit 842 changes the phase of the fifth harmonic compensation signal Y ₅ input from the fifth harmonic compensation unit 81, and outputs the fifth harmonic compensation signal Y ′ ₅ after the phase change. The phase change amount Δθ ₅ is set to an initial value “0”, and is output as it is without changing the phase of the _fifth harmonic compensation signal Y ₅ until the determination signal is input from the divergence determination unit 841. . If input a determination signal from a divergent decision unit 841, the phase changing portion 842 by changing the phase change amount [Delta] [theta] _5, and outputs the fifth-order harmonic compensation signal Y _'5. When the determination signal is no longer input from the divergence determination unit 841, the phase change amount Δθ ₅ is fixed and the fifth-order harmonic compensation signal Y ′ ₅ is output. Phase changer 842 changes the phase change amount [Delta] [theta] ₅ in the reverse direction when the increased fifth harmonic compensation signal Y ₅ Gayori when changing the phase change amount [Delta] [theta] ₅ (eg, increased) (e.g., decrease) By doing so, the phase change amount Δθ ₅ for converging the control is searched.

If the control tends to diverge, the phase change amount Δθ ₅ is changed to search for an appropriate value. If the control does not tend to diverge, the phase change amount Δθ ₅ is fixed and the phase is appropriately changed. A fifth harmonic compensation signal Y ′ ₅ is output. Thereby, it can prevent that control diverges.

  The configuration for preventing control divergence is not limited to the above. Control divergence may be prevented by other methods.

  FIG. 11 is a diagram for explaining a harmonic compensation controller according to another example of the third embodiment, and shows a fifth-order harmonic compensation unit 81 and a divergence prevention unit provided in the subsequent stage.

The divergence prevention unit 84 ′ is for preventing divergence in the suppression control of the fifth harmonic. The divergence prevention unit 84 ′ stops the output of the fifth-order harmonic compensation signal Y ₅ input from the fifth-order harmonic compensation unit 81 when it is determined that the control tends to diverge. The divergence prevention unit 84 ′ is different from the divergence prevention unit 84 shown in FIG. 10 in that an output stop unit 843 is provided instead of the phase change unit 842.

The output stop unit 843 outputs the fifth-order harmonic compensation signal Y ₅ as it is while the determination signal is not input from the divergence determination unit 841, and the fifth-order harmonic compensation signal when the determination signal is input from the divergence determination unit 841. to stop the output of the Y _5.

When the control does not tend to diverge, the fifth harmonic compensation signal Y ₅ input from the fifth harmonic compensation unit 81 is output as it is, and when the control tends to diverge, the fifth harmonic compensation signal Y _5. Output is stopped. If the fifth-order harmonic compensation signal Y ₅ is not output, the fifth-order harmonic suppression control is not performed, so that control divergence can be prevented. Even if the suppression control of the fifth harmonic is not performed, other controls are not affected.

A phase change unit 842 shown in FIG. 10 is provided in front of the output stop unit 843, and the phase change amount Δθ ₅ is searched while the suppression control of the fifth harmonic is stopped, and the phase change amount Δθ is detected. _The output stop unit 843 may be stopped after ₅ is determined.

  In addition, the 7th harmonic compensation part 82 and the 11th harmonic compensation part 83 can also be provided with the same structure, and can prevent the divergence of each harmonic suppression control. Moreover, also in 2nd Embodiment, the divergence of each harmonic suppression control can be prevented similarly.

  In the first to third embodiments, the case where the control circuit according to the present invention is used in a grid-connected inverter system (inverter system) has been described. However, the present invention is not limited to this. The present invention controls, for example, an inverter circuit that performs harmonic compensation used in a harmonic compensation device, a power active filter, a static reactive power compensation device (SVC, SVG), an uninterruptible power supply (UPS), and the like. It can also be applied to circuits. For example, the harmonic compensation device has a function specialized in harmonic suppression by the harmonic compensation controller 8 by eliminating the current controller 74 or the voltage controller 94 in FIG. 7 or FIG. In addition, the present invention is not limited to controlling an inverter circuit that converts direct current to single-phase alternating current. Can be applied. Hereinafter, a case where the present invention is applied to a control circuit of a converter circuit will be described as a fourth embodiment.

  FIG. 12 is a block diagram for explaining a single-phase PWM converter system according to the fourth embodiment. In the figure, the same or similar elements as those in the grid interconnection inverter system A shown in FIG.

  A single-phase PWM converter system C shown in FIG. 12 converts AC power supplied from the power system B into DC power and supplies it to a load L ′. The load L ′ is a DC load. The single-phase PWM converter system C includes a transformer circuit 4, a filter circuit 3, a current sensor 5, a voltage sensor 6, a converter circuit 10, and a control circuit 7.

  The transformer circuit 4 boosts or steps down the AC voltage input from the power system B to a predetermined level. The filter circuit 3 removes a high frequency component from the AC voltage input from the transformer circuit 4 and outputs it to the converter circuit 10. The current sensor 5 detects an alternating current input to the converter circuit 10. The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects an AC voltage input to the converter circuit 10. The detected voltage signal V is input to the control circuit 7. The converter circuit 10 converts the input AC voltage into a DC voltage and outputs it to the load L ′. The converter circuit 10 is a single-phase PWM converter, and is a voltage type converter circuit including two sets and four switching elements (not shown). The converter circuit 10 converts the input AC voltage into a DC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. The converter circuit 10 is not limited to this, and may be a current type converter circuit.

  The control circuit 7 controls the converter circuit 10. The control circuit 7 generates a PWM signal and outputs it to the converter circuit 10 in the same manner as the control circuit 7 of the first embodiment. In FIG. 12, only the configuration for performing input current control and harmonic compensation is described, and the configuration for other control is omitted. Although not shown, the control circuit 7 also includes a DC voltage controller and a reactive power controller, and also controls the output voltage and the input reactive power. The control method performed by the control circuit 7 is not limited to this. For example, when the converter circuit 10 is a current-type converter circuit, output current control may be performed instead of output voltage control.

  Also in this embodiment, the same effect as in the case of the first embodiment can be obtained. In the single-phase PWM converter system C, harmonic compensation is required to suppress harmonics of the input current. In this embodiment, a control system design for harmonic compensation is easily performed using linear control theory. It can be performed.

  The configuration of the single-phase PWM converter system C is not limited to the above. For example, the control circuit 9 may be used instead of the control circuit 7. Alternatively, an inverter circuit may be provided on the output side of the converter circuit 10 to convert the DC power into AC power and supply the AC load to the AC load.

  The control circuit according to the present invention, the grid interconnection inverter system using the control circuit, and the single-phase PWM converter system are not limited to the above-described embodiments. The specific configuration of each part of the control circuit according to the present invention, the grid-connected inverter system using this control circuit, and the single-phase PWM converter system can be variously modified.

A Grid-connected inverter system A 'Inverter system 1 DC power supply 2 Inverter circuit (power conversion circuit)
DESCRIPTION OF SYMBOLS 3 Filter circuit 4 Transformer circuit 5 Current sensor 6 Voltage sensor 7, 9 Control circuit 72 System | strain opposing part production | generation part 74 Current controller (control means)
77 PWM signal generation unit 8 Harmonic compensation controller 81 5th harmonic compensation unit (harmonic compensation means)
82 7th harmonic compensation part (harmonic compensation means)
83 11th harmonic compensation part (harmonic compensation means)
84, 84 'divergence prevention unit 841 divergence determination unit 842 phase change unit 843 output stop unit 94 voltage controller (control means)
10 Converter circuit (Power conversion circuit)
B Power system C Single-phase PWM converter system L, L 'Load

Claims (14)

A control circuit for controlling the driving of a plurality of switching means in a power conversion circuit for single-phase alternating current by means of a PWM signal,
A harmonic compensation means for generating a harmonic compensation signal line Utame control for suppressing a predetermined harmonic component contained in the signal based on the output or input of the power converter circuit,
PWM signal generating means for generating a PWM signal based on the harmonic compensation signal;
With
The harmonic compensation means includes
The harmonic signal is generated by performing signal processing on the signal based on the transfer function G (s) and further performing phase adjustment processing;
The transfer function G (s) is a one-input one-output transfer function having an impulse response f (t), F (s), an angular frequency of the fundamental wave of the single-phase alternating current is ω ₀ , an imaginary unit is j, and n When suppressing the second harmonic,
Is,
A control circuit characterized by that. Control means for generating a fundamental wave compensation signal by performing control to cause the fundamental wave component included in the signal based on the target value to follow.
Correction value signal generation means for generating a correction value signal by adding the fundamental wave compensation signal and the harmonic compensation signal;
Further comprising
The control circuit according to claim 1, wherein the PWM signal generation unit generates a PWM signal based on the correction value signal. The control circuit according to claim 1, wherein a transfer function representing the predetermined control processing is F (s) = K _I / s (where K _I is an integral gain). The transfer function representing the predetermined control process is F (s) = K _P + K _I / s (where K _P and K _I are a proportional gain and an integral gain, respectively). Control circuit. The transfer function representing the predetermined control processing is F (s) = K _P + K _I / s + K _D · s (where K _P , K _I and K _D are proportional gain, integral gain and differential gain, respectively). The control circuit according to claim 1 or 2.   The control circuit according to claim 1, wherein the signal based on the output current or the input current is a signal detected.   The control circuit according to claim 1, wherein the signal based on the output voltage or the input voltage is a signal detected.   The control circuit according to claim 1, wherein the control system is designed using a robust control design.   The control circuit according to claim 8, wherein the control system is designed using the H∞ loop shaping method. Based on the harmonic compensation signal output from the harmonic compensation means, divergence determination means for determining that the control for suppressing the harmonic component tends to diverge,
When it is determined by the divergence determining means that there is a divergence tendency, output stopping means for stopping the output of the harmonic compensation signal;
Further equipped with,
The control circuit according to claim 1. Based on the harmonic compensation signal output from the harmonic compensation means, divergence determination means for determining that the control for suppressing the harmonic component tends to diverge,
When it is determined by the divergence determining means that there is a divergence tendency, the phase change means for changing the phase of the harmonic compensation signal to a phase where the control does not diverge;
Further equipped with,
The control circuit according to claim 1. The divergence determination means determines that the harmonic compensation signal is in a divergence tendency when the harmonic compensation signal exceeds a predetermined threshold.
The control circuit according to claim 10 or 11.   A grid-connected inverter system comprising an inverter circuit and the control circuit according to any one of claims 1 to 12.   A single-phase PWM converter system comprising a converter circuit and the control circuit according to claim 1.