JP2013138543A - Power conversion circuit control circuit, grid connected inverter system, and three phase pwm converter system using control circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a control circuit which, in a control system for current control, performs harmonics compensation as well, the control system having linearity and time invariance.SOLUTION: A control circuit 7 comprises: a three-phase to two-phase conversion unit 73 which converts a three-phase current signal into an α axis and a β axis current signal; an α axis current controller 74 which performs signal processing on a deviation signal between the α axis current signal and a target value to generate a first correction value signal; a β axis current controller 75 which performs signal processing on a deviation signal between the β axis current signal and a target value to generate a second correction value signal; a two-phase to three-phase conversion unit 76 which converts the first and the second correction value signals into three correction value signals; and a PWM signal generation unit 77 which generates a PWM signal on the basis of the three correction value signals. The α axis current controller 74 and the β axis current controller 75 performs signal processing with a transfer function designed by a H∞ control theory using a frequency weight W having (s+ω)×{s+(n×ω)} included in the denominator.

Description

  The present invention relates to a control circuit for controlling the output or input of a power conversion circuit, a grid-connected inverter system and a three-phase PWM converter system using the control circuit, and more particularly to a circuit for compensating for harmonics.

  2. Description of the Related Art Conventionally, a grid-connected inverter system has been developed that converts DC power generated by a solar cell or the like into AC power and supplies it to an electric power system.

  FIG. 14 is a block diagram for explaining a conventional general grid-connected inverter system.

  The grid interconnection inverter system A100 converts the power generated by the DC power supply 1 and supplies it to the three-phase power system B. Hereinafter, the three phases are referred to as a U phase, a V phase, and a W phase.

  The inverter circuit 2 converts a DC voltage input from the DC power supply 1 into an AC voltage by switching a switching element (not shown). The filter circuit 3 removes a switching frequency component included in the AC voltage output from the inverter circuit 2. The transformer circuit 4 boosts (or steps down) the AC voltage output from the filter circuit 3 to the system voltage of the three-phase power system B. The control circuit 700 receives the current signal and the voltage signal detected by the current sensor 5 and the voltage sensor 6, generates a PWM signal based on the current signal and the voltage signal, and outputs the PWM signal to the inverter circuit 2. The inverter circuit 2 performs switching of the switching element based on the PWM signal input from the control circuit 700.

  FIG. 15 is a block diagram for explaining the internal configuration of the control circuit 700.

  The current signal of each phase input from the current sensor 5 is input to the three-phase / two-phase converter 73.

  The three-phase / two-phase conversion unit 73 converts the three input current signals Iu, Iv, and Iw into an α-axis current signal Iα and a β-axis current signal Iβ. The three-phase / two-phase conversion unit 73 performs a so-called three-phase / two-phase conversion process (αβ conversion process). The current signals Iu, Iv, Iw are respectively converted into an α-axis component and a β-axis component that are orthogonal to each other. The α-axis current signal Iα and the β-axis current signal Iβ are generated by decomposing and collecting the respective axis components.

The conversion process performed by the three-phase / two-phase conversion unit 73 is represented by a determinant represented by the following equation (1).

  The rotation coordinate conversion unit 78 converts the α-axis current signal Iα and the β-axis current signal Iβ input from the three-phase / two-phase conversion unit 73 into a d-axis current signal Id and a q-axis current signal Iq in the rotation coordinate system. Is. The rotating coordinate system is an orthogonal coordinate system having orthogonal d-axis and q-axis and rotating in the same direction at the same angular velocity as the fundamental wave of the system voltage of the three-phase power system B. As a concept opposite to the rotating coordinate system, a non-rotating coordinate system is a stationary coordinate system. The rotation coordinate conversion unit 78 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id and a q-axis current signal Iq in the rotating coordinate system.

The conversion process performed by the rotation coordinate conversion unit 78 is expressed by a determinant represented by the following expression (2).

  The LPF 74a and the LPF 75a are low-pass filters and pass only the DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. Through the rotation coordinate conversion process, the fundamental wave components of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id and the q-axis current signal Iq, respectively. The PI control unit 74b and the PI control unit 75b perform PI control (proportional integration control) based on the deviation between the DC component of the d-axis current signal Id and the q-axis current signal Iq and the target value, respectively, and the fundamental compensation signal Xd and Xq are output. Since a DC component can be used as the target value, the PI control unit 74b and the PI control unit 75b can perform control with high accuracy.

  The stationary coordinate conversion unit 79 converts the fundamental wave compensation signals Xd and Xq input from the PI control unit 74b and the PI control unit 75b, respectively, into two fundamental wave compensation signals Xα and Xβ in the stationary coordinate system. The rotation coordinate conversion unit 78 performs a reverse conversion process. The static coordinate conversion unit 79 performs a so-called static coordinate conversion process (inverse dq conversion process), and uses the fundamental wave compensation signals Xd and Xq of the rotating coordinate system based on the phase θ to compensate the fundamental wave of the static coordinate system The signals are converted into signals Xα and Xβ.

The conversion process performed by the stationary coordinate conversion unit 79 is expressed by a determinant represented by the following expression (3).

  The two-phase / three-phase conversion unit 76 adds the harmonic compensation signals Yα and Yβ output from the harmonic compensation controller 800 described later to the fundamental wave compensation signals Xα and Xβ output from the stationary coordinate conversion unit 79. The value signals X′α, X′β are converted into three correction value signals Xu, Xv, Xw. The two-phase / three-phase conversion unit 76 performs a so-called two-phase / three-phase conversion process (reverse αβ conversion process), and performs a conversion process opposite to the three-phase / two-phase conversion unit 73.

The conversion process performed by the two-phase / three-phase conversion unit 76 is expressed by a determinant represented by the following equation (4).

  The PWM signal generation unit 77 generates and outputs a PWM signal based on the correction value signals Xu, Xv, and Xw output from the two-phase / three-phase conversion unit 76.

  The control circuit 700 has a function of suppressing harmonics input from the three-phase power system B and harmonics output from the inverter circuit 2. The harmonic compensation controller 800 extracts a harmonic component from each phase current signal input from the current sensor 5 and outputs a harmonic compensation signal for outputting a harmonic that cancels the harmonic component. The grid interconnection inverter system A100 outputs harmonics based on the harmonic compensation signal (that is, harmonics having a phase opposite to the detected harmonics) and cancels the harmonics, thereby suppressing harmonics.

  FIG. 16 is a block diagram for explaining the internal configuration of the harmonic compensation controller 800. In general, the harmonics from the three-phase power system B or the inverter circuit 2 are mostly the fifth harmonic, the seventh harmonic, and the eleventh harmonic. In order to suppress these harmonics, a fifth harmonic compensator 810 for suppressing the fifth harmonic, a seventh harmonic compensator 820 for suppressing the seventh harmonic, and an eleventh harmonic are provided. An eleventh harmonic compensation unit 830 for suppression is provided in the harmonic compensation controller 800. The fifth harmonic compensation unit 810 includes a rotation coordinate conversion unit 811, LPFs 812 and 813, I control units 814 and 815, and a stationary coordinate conversion unit 816. Since the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 have the same configuration as the fifth harmonic compensation unit 810, the description and description in FIG. 16 are omitted.

The rotation coordinate conversion unit 811 converts the α-axis current signal Iα and β-axis current signal Iβ input from the three-phase / two-phase conversion unit 73 into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotation coordinate system. To convert. This rotating coordinate system is an orthogonal coordinate system that rotates in the opposite direction at an angular velocity that is five times the angular velocity of the fundamental wave of the system voltage. The rotation coordinate conversion unit 811 performs a so-called rotation coordinate conversion process (dq conversion process). The α-axis current signal Iα and the β-axis current signal Iβ of the stationary coordinate system are converted into the system voltage detected by the phase detection unit 71. Based on the phase θ of the fundamental wave, it is converted into a d-axis current signal Id ₅ and a q-axis current signal Iq ₅ in the rotating coordinate system.

The conversion process performed by the rotating coordinate conversion unit 811 is represented by a determinant represented by the following expression (5).
Note that the rotation coordinate conversion units of the seventh harmonic compensation unit 820 and the eleventh harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (5).

LPF 812 and LPF 813 are low-pass filters and pass only the DC components of d-axis current signal Id ₅ and q-axis current signal Iq ₅ , respectively. By the rotational coordinate conversion process, the fifth harmonics of the α-axis current signal Iα and the β-axis current signal Iβ are converted into DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively. The I control unit 814 and the I control unit 815 perform I control (integration control) based on the DC components of the d-axis current signal Id ₅ and the q-axis current signal Iq ₅ , respectively, and the fifth harmonic compensation signals Yd ₅ , Yq ₅ is output. Since a DC component can be used as the target value, the I control unit 814 and the I control unit 815 can perform highly accurate control.

The stationary coordinate conversion unit 816 converts the fifth harmonic compensation signals Yd ₅ and Yq ₅ input from the I control unit 814 and the I control unit 815, respectively, into two fifth harmonic compensation signals Yα ₅ and Yβ in the stationary coordinate system. ₅ is converted, and the conversion process reverse to that of the rotating coordinate conversion unit 811 is performed. The static coordinate conversion unit 816 performs so-called static coordinate conversion processing (inverse dq conversion processing), and converts the fifth-order harmonic compensation signals Yd ₅ and Yq ₅ of the rotating coordinate system into a static coordinate system based on the phase θ. of the fifth harmonic compensation signal Yarufa _5, converted into Ybeta _5.

The conversion process performed by the stationary coordinate conversion unit 816 is represented by a determinant represented by the following expression (6).
Note that the stationary coordinate conversion units of the seventh-order harmonic compensation unit 820 and the eleventh-order harmonic compensation unit 830 perform processing in which (−5θ) is set to 7θ and (−11θ), respectively, in the above equation (6).

Similarly, the seventh harmonic compensator 820 7 harmonic compensation signal Yarufa _7, generates and outputs a Ybeta _7, 11 harmonic compensator 830 11 harmonic compensation signal Yarufa _11, the Ybeta ₁₁ Generate and output. 5 harmonic compensation signal Yarufa _5, 7 harmonic compensation signal Yα _7, 11 th and harmonic compensation signal Yarufa the harmonic compensation signal Yarufa ₁₁ by adding, fifth harmonic compensation signal Ybeta _5, 7 harmonic compensation The harmonic compensation signal Yβ obtained by adding the signal Yβ ₇ and the 11th-order harmonic compensation signal Yβ ₁₁ is output from the harmonic compensation controller 800 to the fundamental wave compensation signals Xα and Xβ output from the stationary coordinate converter 79, respectively. The added values are input to the two-phase / three-phase converter 76 as correction value signals X′α and X′β.

Japanese Patent No. 4421700 JP 2009-44897 A

  However, there is a problem that a great effort is required to design a control system for current control. Recent grid-connected inverter systems are required to have high-speed responsiveness in control, such as returning the output within a predetermined time with respect to the instantaneous drop. In order to design the control system so as to satisfy such requirements, it is necessary to optimally design the parameters of the LPF 74a and the LPF 75a and the proportional gain and integral gain of the PI control unit 74b and the PI control unit 75b. However, since the rotating coordinate conversion unit 78 and the stationary coordinate conversion unit 79 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  Further, in order to design a control system for each harmonic compensation, it is necessary to optimally design the parameters of the LPFs 812 and 813 and the integral gains of the I control units 814 and 815. However, since the rotating coordinate conversion unit 811 and the stationary coordinate conversion unit 816 perform nonlinear time-varying processing, it has not been possible to design a control system using linear control theory. Moreover, since the control system includes nonlinear time-varying processing, system analysis cannot be performed.

  The present invention has been conceived under the above circumstances, and a harmonic compensation is also performed in a current control system, and the control system has linearity and time invariance. Its purpose is to provide.

  In order to solve the above problems, the present invention takes the following technical means.

A control circuit provided by the first aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to three-phase alternating current using a PWM signal, and includes a three-phase control circuit. Three-phase to two-phase conversion means for converting three signals based on an output or input into a first signal and a second signal; a target value of each of the first signal, the second signal, and each fundamental wave component; Deviation signal generation means for generating a first deviation signal and a second deviation signal, which are deviations of the first deviation signal, and a fundamental wave component included in each of the first deviation signal and the second deviation signal are controlled to zero. And control means for generating a first correction value signal and a second correction value signal for suppressing and controlling predetermined harmonic components contained in the first deviation signal and the second deviation signal, respectively. The first Two-phase / three-phase conversion means for converting the correction value signal and the second correction value signal into three correction value signals, and a PWM signal generation means for generating a PWM signal based on the three correction value signals. And the control means performs signal processing on the first deviation signal and the second deviation signal by a transfer function designed by a robust control design method using the frequency weight W, so that the first deviation signal is processed. When the correction value signal and the second correction value signal are generated, the angular frequency of the fundamental wave of the three-phase alternating current is set to ω ₀ , and the nth harmonic is suppressed, the denominator of the frequency weight W is ( s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } is included.

A control circuit provided by the second aspect of the present invention is a control circuit that controls driving of a plurality of switching means in a power conversion circuit related to three-phase alternating current by using a PWM signal, Deviation signal generating means for generating three deviation signals, which are deviations between the three signals based on the output or input and the target values of the respective fundamental wave components, and the three deviation signals as the first deviation signal and the second deviation signal Three-phase to two-phase conversion means for converting to a deviation signal, and control the fundamental wave components included in the first deviation signal and the second deviation signal to zero, respectively, and the first deviation signal and the Control means for generating a first correction value signal and a second correction value signal for suppressing and controlling predetermined harmonic components respectively included in the second deviation signal; the first correction value signal; 2 Two-phase three-phase conversion means for converting the correction value signal into three correction value signals, and PWM signal generation means for generating a PWM signal based on the three correction value signals, the control means, By performing signal processing on the first deviation signal and the second deviation signal by a transfer function designed by the robust control design method using the frequency weight W, the first correction value signal and the second correction signal When a correction value signal is generated and the angular frequency of the fundamental wave of the three-phase alternating current is set to ω ₀ and the nth-order harmonic is suppressed, the denominator of the frequency weight W is (s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } is included.

In a preferred embodiment of the present invention, when the frequency weight W is a predetermined real number k,
It is.

  In a preferred embodiment of the present invention, the three signals are signals obtained by detecting an output current or an input current of each phase.

  In a preferred embodiment of the present invention, the three signals are signals obtained by detecting an output voltage or an input voltage of each phase.

  In a preferred embodiment of the present invention, the transfer function is designed by the H∞ loop shaping method using the frequency weight W.

  The grid-connected inverter system provided by the third aspect of the present invention includes an inverter circuit and a control circuit provided by the first or second aspect of the present invention.

  A three-phase PWM converter system provided by the fourth aspect of the present invention includes a converter circuit and a control circuit provided by the first or second aspect of the present invention.

According to the present invention, a first correction value signal and a second correction value signal are generated by performing signal processing on the first deviation signal and the second deviation signal, respectively, using a transfer function. The transfer function is designed using a frequency weight W including (s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } in the denominator. Are controlled simultaneously. Further, since the control system has linearity and time invariance, the design of the control system can be facilitated using a design method based on the linear control theory.

  Other features and advantages of the present invention will become more apparent from the detailed description given below with reference to the accompanying drawings.

It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process. It is a block diagram for demonstrating the method to convert the process accompanied by a rotation coordinate transformation and a stationary coordinate transformation into a linear time invariant process, and is represented by the matrix. It is a block diagram for demonstrating the calculation of a matrix. It is a block diagram which shows the process which performs static coordinate transformation after performing PI control after performing rotational coordinate transformation. It is a block diagram which shows the process which performs a stationary coordinate transformation after performing I control after performing a rotational coordinate transformation. Is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. It is a figure for demonstrating the signal for a positive phase, and the signal for a reverse phase. It is a Bode diagram for analyzing the frequency weight W. It is a Bode diagram for analyzing a transfer function K. It is a block diagram for demonstrating the grid connection inverter system which concerns on 1st Embodiment. It is a figure for demonstrating a simulation result. It is a block diagram for demonstrating the control circuit which concerns on 2nd Embodiment. It is a block diagram for demonstrating the three-phase PWM converter system which concerns on 3rd Embodiment. It is a block diagram for demonstrating the conventional general grid connection inverter system. It is a block diagram for demonstrating the internal structure of a control circuit. It is a block diagram for demonstrating the internal structure of a harmonic compensation controller.

  Embodiments of the present invention will be specifically described below with reference to the drawings.

  First, a method for converting processing involving rotational coordinate transformation and stationary coordinate transformation into linear time-invariant processing will be described.

  Fig.1 (a) is a figure for demonstrating the process accompanied by rotation coordinate transformation and stationary coordinate transformation. In this processing, first, the signals α and β are converted into signals d and q by rotational coordinate conversion. The signals d and q are each processed by a predetermined transfer function F (s), and signals d 'and q' are output. Next, the signals d ′ and q ′ are converted into signals α ′ and β ′ by stationary coordinate transformation. The nonlinear time-varying process shown in FIG. 1A is converted into a process using a matrix G of linear time-invariant transfer functions shown in FIG.

The rotational coordinate transformation shown in FIG. 1A is represented by a determinant of the following equation (7), and the stationary coordinate transformation is represented by a determinant of the following equation (8).

  Therefore, the process shown in FIG. 1A can be expressed as shown in FIG. 2A using a matrix. The matrix G shown in FIG. 1B can be calculated by calculating the product of the three matrices shown in FIG. 2A and making the calculated matrix a linear time-invariant matrix. At this time, the calculation is performed after converting the matrix of the stationary coordinate conversion and the rotation coordinate conversion into a matrix product.

The matrix for rotational coordinate conversion can be converted into a product of the right-hand side matrix shown in the following equation (9).
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

Also, the matrix of the static coordinate conversion can be converted into the product of the matrix on the right side shown in the following equation (10). The matrix in the middle of the matrix product is a linear time-invariant matrix.
Where j is the imaginary unit, exp () is the exponential function of the base e of the natural logarithm,
It is. T ⁻¹ is an inverse matrix of T.

From the Euler's formula, calculating by substituting exp (jθ) = cos θ + jsin θ and exp (−jθ) = cos θ−jsin θ,
It can be confirmed that

When the product of the three matrices shown in FIG. 2A is calculated using the above equations (9) and (10), and the matrix G is calculated, the following equation (11) is calculated.

When attention is paid to the element in the first row and the first column of the three matrixes in the center of the above equation (11) and this is expressed in a block diagram, the block diagram shown in FIG. 3 is obtained. When calculating the input / output characteristics of the block diagram shown in FIG.
It becomes. Where F (s) is a one-input one-output transfer function having an impulse response f (t).

Here, when θ (t) = ω ₀ t, θ (t) −θ (τ) = ω ₀ t−ω ₀ τ = ω ₀ (t−τ) = θ (t−τ). The input / output characteristics of the block diagram shown in FIG. 3 are equal to those of a linear time invariant system having an impulse response f (t) exp (−jω ₀ t). When the impulse response f (t) exp (−jω ₀ t) is Laplace transformed, a transfer function F (s + jω ₀ ) is obtained. The input / output characteristics when exp (jθ (t)) and exp (−jθ (t)) in the block diagram shown in FIG. 3 are interchanged are the input / output characteristics of the transfer function F (s−jω ₀ ). become.

Therefore, if the calculation is further advanced from the above equation (11),
Is calculated.

  Thereby, the process shown in FIG. 2A can be converted into the process shown in FIG. The process shown in FIG. 2B is equivalent to the process of performing the stationary coordinate conversion after performing the process represented by the predetermined transfer function F (s) after performing the rotational coordinate conversion. This system is a linear time-invariant system.

The transfer function of the PI control (proportional integral control) controller is expressed as F (s) = K _P + K _I / s, where K _P and K _I are the proportional gain and integral gain, respectively. Therefore, the transfer function matrix G _PI indicating the process shown in FIG. 4, that is, the process equivalent to the process of performing the stationary coordinate conversion after performing the PI control after performing the rotational coordinate conversion, uses the above equation (12). Thus, the following equation (13) is calculated.

Further, the transfer function of the I control (integral control) controller, the integral gain and K _I, represented by _{F (s) = K I /} s. Therefore, the processing shown in FIG. 5, i.e., the matrix G _I of the transfer function from performing a rotational coordinate transformation showing the processing of the processing equivalent to a still coordinate transformation after the I control, using the equation (12) Is calculated as in the following equation (14).

Figure 6 is a Bode diagram for analyzing a transfer function which is the elements of the matrix G _I. FIG (a) is first row and the first column elements (hereinafter, the same applies for. Other elements described as "(1,1) element".) The matrix G _I and (2,2) element of FIG. 4B shows the transfer function of the (1,2) element of the matrix G _I , and FIG. 4C shows the transfer function of the (2,1) element of the matrix G _I. Is shown. The figure shows the case where the frequency of the fundamental wave of the system voltage (hereinafter referred to as “center frequency”. Also, the angular frequency corresponding to the center frequency is referred to as “center angular frequency”) is 60 Hz (that is, angular frequency). ω ₀ = 120π), and the integral gain K _I is “0.1”, “1”, “10”, “100”.

FIG (a), the amplitude characteristic shown by (b) and (c) are all, there is a peak in the center frequency, the integral gain K _I is increased, the amplitude characteristic is increased. Further, the phase characteristic shown in FIG. 5A is 0 degree at the center frequency. In other words, the transfer function of (1, 1) element and (2,2) element of the matrix G _I is passed through without the signal of the center frequency (center angular frequency) changes the phase. The phase characteristic shown in FIG. 5B is 90 degrees at the center frequency. In other words, the transfer function of the (1,2) element of the matrix G _I causes the phase of the signal of the center frequency (center angular frequency) passes advancing 90 degrees. On the other hand, the phase characteristic shown in FIG. 4C is −90 degrees at the center frequency. In other words, the transfer function of (2,1) element of the matrix G _I is the phase of the signal of the center frequency (center angular frequency) passing delayed 90 degrees.

The matrix G (G _PI , G _I ) of the transfer function described above is for controlling the positive phase component of the fundamental wave component. Next, a method for controlling the reverse phase will be described.

  FIG. 7 is a diagram for explaining the signal for the positive phase and the signal for the negative phase of the fundamental wave. FIG. 4A shows the signal for the positive phase of the fundamental wave, and FIG. 4B shows the signal for the reverse phase of the fundamental wave.

In FIG. 5A, the positive phase signals of the fundamental wave components of the current signals Iu, Iv, Iw are indicated by the vectors u, v, w of broken line arrows. The vectors u, v, and w have directions different from each other by 120 degrees, and are arranged in a clockwise order and rotated counterclockwise at an angular frequency ω ₀ . The α-axis signal and β-axis signal obtained by converting the positive phase signal into three-phase / two-phase signals are indicated by vectors α and β of solid arrows. The vectors α and β are different in direction by 90 degrees in the clockwise order, and are rotated counterclockwise at the angular frequency ω ₀ .

That is, the α-axis signal is 90 degrees ahead of the β-axis signal. even if the processing shown in the transfer function of (1, 1) element of the matrix G _I to α-axis signal, the phase does not change (see FIG. 6 (a)). Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I to β-axis signal, it advances the phase by 90 degrees (see Figure 6 (b)). Therefore, since both phases become the same phase as the α-axis signal, they are strengthened by adding both. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I to α-axis signal, the phase is delayed 90 degrees (see Figure 6 (c)). Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in β axis signal, the phase does not change. Therefore, since both phases become the same phase as the β-axis signal, they are strengthened by adding both.

The reverse phase component is a component whose phase sequence is opposite to the normal phase component. In FIG. 7 (b), the antiphase component signals of the fundamental components of the current signals Iu, Iv, Iw are indicated by vectors u, v, w of broken line arrows. The vectors u, v, and w have directions different from each other by 120 degrees, and are arranged in the counterclockwise order and rotated in the counterclockwise direction at the angular frequency ω ₀ . An α-axis signal and a β-axis signal obtained by three-phase / two-phase conversion of the antiphase signal are indicated by vectors α and β of solid arrows. The vectors α and β are different in the direction of 90 degrees in the counterclockwise order, and rotate in the counterclockwise direction at the angular frequency ω ₀ .

That is, the α-axis signal is 90 degrees behind the β-axis signal. even if the processing shown in the transfer function of (1, 1) element of the matrix G _I to α-axis signal, the phase does not change. Further, when the processing shown in the transfer function of the (1,2) element of the matrix G _I to β-axis signal, it advances the phase by 90 degrees. Therefore, since the phases of both are reversed, adding them together cancels each other. On the other hand, when the processing shown in the transfer function of (2,1) element of the matrix G _I to α-axis signal, the phase is delayed 90 degrees. Further, even if the processing shown in the transfer function of the (2,2) element of the matrix G _I in β axis signal, the phase does not change. Therefore, since the phases of both are reversed, adding them together cancels each other. Accordingly, the matrix G _I of the transfer function, performs control of the positive phase component, control of the reverse phase is not performed.

If interchanged and (1,2) element of the matrix G _I of the transfer function and (2,1) element, contrary to the above, so that cancel each other positive phase component, reverse-phase components constructive. Therefore, when controlling the reversed phase, the (1,2) of the matrix G _I of the transfer function element and the (2,1) element and matrix may be used with interchanged. The same applies to the transfer function matrices G and _GPI .

  Next, a method for controlling both the positive phase portion and the reverse phase portion will be described.

Matrix processing shown in the transfer function of (1, 1) element and (2,2) element of G _I is passed through without changing the positive phase component and negative phase of the phase (see FIG. 6 (a)). Therefore, when the matrix G ′ _I of the transfer function shown in the following equation (15) in which the (1,2) and (2,1) elements of the matrix G _I shown in the above equation (14) are set to “0” is used, Both the normal phase and the reverse phase can be controlled. The same applies to the transfer function matrix _GPI , and the transfer function matrix G ′ shown in the following expression (16) in which the (1,2) element and the (2,1) element are set to “0” in the above expression (13). By using _PI , it is possible to control both the positive phase portion and the reverse phase portion.

  Next, a method for controlling harmonic components will be described.

The matrix G of the transfer function shown in the above equation (12) is for controlling the fundamental wave component. The n-th harmonic is an angular frequency component obtained by multiplying the angular frequency of the fundamental wave by n. When three-phase / two-phase conversion is performed on the positive phase component of the n-th harmonic, there are cases where the phase of the α-axis signal is advanced or delayed. In the case of n = 3k + 1 (k = 1, 2,...), the order of the phase of the positive phase signal of the n-th harmonic matches the order of the phases of the positive phase signal of the fundamental wave. That is, if the U, V, and W phase signals for the positive phase of the fundamental wave are Vu = Vcos θ, Vv = Vcos (θ-2π / 3), and Vw = Vcos (θ-4π / 3), for example, 7 positive phase of U of harmonics, V, each signal of the W _{phase, Vu 7 = V 7 cos7θ,} Vv 7 = V 7 cos (7θ-14π / 3) = V 7 cos (7θ-2π / 3) , the _{Vw 7 = V 7 cos (7θ} -28π / 3) = V 7 cos (7θ-4π / 3). In this case, the phase order matches the phase order of the positive phase signal of the fundamental wave, and the phase of the α-axis signal advances 90 degrees from the phase of the β-axis signal, as in FIG. Therefore, the matrix of the transfer function when controlling the positive phase component of the nth harmonic (n = 3k + 1) is the transfer function shown in the following equation (17) where ω ₀ is n · ω ₀ in the above equation (12). A matrix G _n is obtained. On the other hand, in the case of n = 3k + 2 (k = 0, 1, 2,...), The order of the phases of the n-order harmonics of the positive phase signal coincides with the order of the phases of the negative-phase signal of the fundamental wave. That is, when the U, V, and W phase signals Vu, Vv, and Vw for the positive phase of the fundamental wave are as described above, for example, the U, V, and W phase signals for the positive phase of the fifth harmonic are respectively Vu ₅ = V ₅ cos 5θ, Vv ₅ = V ₅ cos (5θ-10π / 3) = V ₅ cos (5θ-4π / 3), Vw ₅ = V ₅ cos (5θ-20π / 3) = V ₅ cos ( 5θ-2π / 3). In this case, the phase order coincides with the phase order of the anti-phase signal of the fundamental wave, and the phase of the α-axis signal is delayed by 90 degrees from the phase of the β-axis signal, as in FIG. Therefore, the matrix of the transfer function in the case of controlling the positive phase component of the n-th harmonic (n = 3k + 2) is represented by the above equation (12) where ω ₀ is n · ω ₀ and the (1,2) element and (2, 1) The transfer function matrix G _n shown in the following equation (17 ′) with elements replaced is obtained.

When controlling both the positive phase component and the reverse phase component, in the matrix G _n shown in the above equations (17) and (17 ′), the (1,2) and (2,1) elements are set to “0”. You can do it. In this case, in the case of n = 3k + 1 (k = 1, 2,...) And n = 3k + 2 (k = 0, 1, 2,...), The transfer function matrix G ′ _n shown in the following equation (18) is used. Become.

Further, the transfer function matrix G ′ _In for performing I control on both the positive phase component and the reverse phase component is expressed by the following equation (14) where ω ₀ is n · ω ₀ , and the (1,2) element and (2 , 1) Assuming that the element is “0”, the following equation (19) is calculated. The following equation (19) can also be calculated as F (s) = K _I / s in the above equation (18). Similarly, a transfer function matrix G ′ _PIn for PI control of both the positive phase component and the reverse phase _component is calculated as in the following equation (20).

As shown in the above equations (15) and (16), when controlling both the positive phase component and the negative phase component of the fundamental wave, the H∞ loop is used using the frequency weight W ₁ shown in the following equation (21). Controllers can be designed by shaping methods. By using the H∞ loop shaping method, the most stable controller that satisfies the design specifications can be designed.

Further, as shown in (19) and (20), n-th harmonic positive phase of, when performing control of both reversed phase, using the frequency weight W _n shown in the following (22) The controller can be designed by the H∞ loop shaping method.

When controlling both the positive and negative phase components of the fundamental wave, and controlling both the positive and negative phase components of the nth harmonic, the frequency weight W shown in the following equation (23) is used. The controller may be designed using the H∞ loop shaping method. Note that k is a real number corresponding to the response speed.

When a plurality of harmonics are controlled, a corresponding order term may be added to the denominator of the equation (23). For example, when controlling the fifth, seventh, and eleventh harmonics, the frequency weight W shown in the following equation (24) may be used.

FIG. 8 is a Bode diagram for analyzing the frequency weight W shown in the equation (24). This figure shows a case where the angular frequency ω ₀ = 120π and k = 10 ²⁴ . As shown in the figure, the frequency weights W are ω ₀ (= 120π≈377 [rad / sec]), 5ω ₀ (= 600π≈1884 [rad / sec]), 7ω ₀ (= 840π≈2638 [rad / sec]. sec]), 11ω ₀ (= 1320π≈4145 [rad / sec]).

When a controller is designed by the H∞ loop shaping method using this frequency weight W, a transfer function K shown in the following equation (25) is calculated. It is calculated when the inductance of the reactor of the LC filter included in the filter circuit 3 of FIG. 10 described later is L = 1000 μH, the capacitance of the capacitor is C = 20 μF, and the leakage inductance of the transformer circuit 4 is LT = 500 μH ing.

FIG. 9 is a Bode diagram for analyzing the transfer function K shown in the equation (25). As shown in the figure, the transfer function K inherits the characteristic that peaks at ω ₀ , 5ω ₀ , 7ω ₀ , and 11ω ₀ .

  Hereinafter, a case where a controller designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (24) is applied to a control circuit of a grid-connected inverter system will be described as a first embodiment of the present invention. .

  FIG. 10 is a block diagram for explaining the grid-connected inverter system according to the first embodiment.

  As shown in the figure, the grid-connected inverter system A includes a DC power source 1, an inverter circuit 2, a filter circuit 3, a transformer circuit 4, a current sensor 5, a voltage sensor 6, and a control circuit 7.

  The DC power source 1 is connected to the inverter circuit 2. The inverter circuit 2, the filter circuit 3, and the transformer circuit 4 are connected in series to the output lines of the U-phase, V-phase, and W-phase output voltages in this order, and connected to the three-phase AC power system B. Yes. The current sensor 5 and the voltage sensor 6 are installed on the output side of the transformer circuit 4. The control circuit 7 is connected to the inverter circuit 2. The grid interconnection inverter system A converts the DC power output from the DC power supply 1 into AC power and supplies it to the power grid B. In addition, the structure of the grid connection inverter system A is not restricted to this. For example, the current sensor 5 and the voltage sensor 6 may be provided on the input side of the transformer circuit 4, or other sensors necessary for controlling the inverter circuit 2 may be provided. Further, the transformer circuit 4 may be provided on the input side of the filter circuit 3, or a so-called transformer-less system in which the transformer circuit 4 is not provided. Further, a DC / DC converter circuit may be provided between the DC power supply 1 and the inverter circuit 2.

  The DC power source 1 outputs DC power and includes, for example, a solar battery. A solar cell generates direct-current power by converting solar energy into electrical energy. The DC power source 1 outputs the generated DC power to the inverter circuit 2. Note that the DC power source 1 is not limited to one that generates DC power from a solar cell. For example, the DC power source 1 may be a fuel cell, a storage battery, an electric double layer capacitor, a lithium ion battery, or AC power generated by a diesel engine generator, a micro gas turbine generator, a wind turbine generator, or the like. It may be a device that converts to DC power and outputs it.

  The inverter circuit 2 converts a DC voltage input from the DC power source 1 into an AC voltage and outputs the AC voltage to the filter circuit 3. The inverter circuit 2 is a three-phase inverter, and is a PWM control type inverter circuit including three sets of six switching elements (not shown). The inverter circuit 2 converts the DC voltage input from the DC power source 1 into an AC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. In addition, the inverter circuit 2 is not limited to this, For example, a multilevel inverter may be sufficient.

  The filter circuit 3 removes high frequency components due to switching from the AC voltage input from the inverter circuit 2. The filter circuit 3 includes a low pass filter including a reactor and a capacitor. The AC voltage from which the high frequency component has been removed by the filter circuit 3 is output to the transformer circuit 4. The configuration of the filter circuit 3 is not limited to this, and any known filter circuit for removing high frequency components may be used. The transformer circuit 4 boosts or lowers the AC voltage output from the filter circuit 3 to a level substantially the same as the system voltage.

  The current sensor 5 detects the alternating current of each phase output from the transformer circuit 4 (that is, the output current of the grid interconnection inverter system A). The detected current signal I (Iu, Iv, Iw) is input to the control circuit 7. The voltage sensor 6 detects the system voltage of each phase of the power system B. The detected voltage signal V (Vu, Vv, Vw) is input to the control circuit 7. Note that the output voltage output by the grid interconnection inverter system A substantially matches the grid voltage.

  The control circuit 7 controls the inverter circuit 2 and is realized by, for example, a microcomputer. The control circuit 7 generates a PWM signal based on the current signal I input from the current sensor 5 and the voltage signal V input from the voltage sensor 6 and outputs the PWM signal to the inverter circuit 2. The control circuit 7 generates a command value signal for commanding the waveform of the output voltage output from the grid interconnection inverter system A based on the detection signal input from each sensor, and is generated based on the command value signal. Output a pulse signal as a PWM signal. The inverter circuit 2 outputs an alternating voltage having a waveform corresponding to the command value signal by switching each switching element on and off based on the input PWM signal. The control circuit 7 controls the output current by changing the waveform of the command value signal to change the waveform of the output voltage of the grid interconnection inverter system A. Thereby, the control circuit 7 performs various feedback controls. In addition, the control circuit 7 suppresses harmonics by causing the inverter circuit 2 to output harmonics for canceling the harmonics input from the power system B.

  In FIG. 10, only the structure for performing output current control and harmonic suppression control is described, and the structure for other controls is omitted. Actually, the control circuit 7 performs DC voltage control (feedback control performed so that the input DC voltage becomes a preset voltage target value) and reactive power control (reactive power target value with preset output reactive power) (Feedback control to be performed) is also performed. The control method performed by the control circuit 7 is not limited to this. For example, output voltage control or active power control may be performed.

  The control circuit 7 includes a system counter component generation unit 72, a three-phase / two-phase conversion unit 73, an α-axis current controller 74, a β-axis current controller 75, a two-phase / three-phase conversion unit 76, and a PWM signal generation unit 77. I have.

  The system counter-part generating unit 72 receives the voltage signal V from the voltage sensor 6 and generates and outputs system command value signals Ku, Kv, Kw. The system command value signals Ku, Kv, Kw serve as a reference for the command value signal for instructing the waveform of the output voltage output from the system interconnection inverter system A. The system command value signals Ku, Kv, Kw are described later. The command value signal is generated by correcting with the correction value signals Xu, Xv, and Xw.

  The three-phase / two-phase conversion unit 73 is the same as the three-phase / two-phase conversion unit 73 shown in FIG. 15, and the three current signals Iu, Iv, Iw input from the current sensor 5 are converted into α-axis current signals. It converts to Iα and β-axis current signal Iβ. The conversion process performed by the three-phase / two-phase conversion unit 73 is represented by the determinant shown in the above equation (1).

  The α-axis current controller 74 receives the deviation between the α-axis current signal Iα output from the three-phase / two-phase converter 73 and the α-axis current target value that is the target value of the fundamental component, and receives the correction value signal X ′. α is generated. The β-axis current controller 75 receives the deviation between the β-axis current signal Iβ output from the three-phase / two-phase converter 73 and the β-axis current target value that is the target value of the fundamental component, and receives the correction value signal X ′. β is generated.

The α-axis current controller 74 and the β-axis current controller 75 are controllers designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (24). Processing performed by the α-axis current controller 74 and the β-axis current controller 75 is linear time-invariant processing. Therefore, a control system design using linear control theory can be performed. As the angular frequency ω ₀ , the angular frequency of the fundamental wave of the system voltage (for example, ω ₀ = 120π [rad / sec] (60 [Hz])) is set in advance, and the real number k is also set in advance. In addition, the α-axis current controller 74 and the β-axis current controller 75 perform processing for maximizing the stability margin, and in this process, the phase is adjusted to correct the phase delay in the control loop. ing. The design method used for control system design is not limited to the H∞ loop shaping method, and for example, a robust control design method such as optimal control, H∞ control theory, and mixed sensitivity problem can be used.

  The α-axis current controller 74 and the β-axis current controller 75 control the fundamental wave components included in the α-axis current signal Iα and the β-axis current signal Iβ to target values, respectively, and the α-axis current signal Iα and β-axis current signal Control is performed to suppress the fifth, seventh and eleventh harmonic components contained in Iβ.

In the present embodiment, the case where the α-axis current controller 74 and the β-axis current controller 75 perform suppression control of the fifth-order, seventh-order, and eleventh-order harmonics has been described, but the present invention is not limited thereto. The frequency weight W may be set according to the harmonic order that needs to be suppressed. For example, when it is desired to suppress only the fifth harmonic, the denominator of the frequency weight W shown in the above equation (24) may be (s ² + ω ₀ ² ) · {s ² + (5ω ₀ ) ² } only. Further, if it is desired to further suppress the 13th harmonic, {s ² + (13ω ₀ ) ² } is further applied to the denominator of the frequency weight W shown in the above equation (24).

  In the present embodiment, as the α-axis current target value and the β-axis current target value, those obtained by converting the d-axis current target value and the q-axis current target value into stationary coordinates are used. A correction value for DC voltage control (not shown) is used for the d-axis current target value, and a correction value for reactive power control (not shown) is used for the q-axis current target value. When a three-phase current target value is given, the target value may be converted into three-phase / two-phase to obtain an α-axis current target value and a β-axis current target value. Further, the respective deviations between the three current signals Iu, Iv, Iw and the three-phase current target value are calculated in advance, and the three deviation signals are converted into a three-phase / two-phase to obtain an α-axis current controller 74 and You may make it input into the beta-axis current controller 75. FIG. When the α-axis current target value and the β-axis current target value are directly given, the target values may be used as they are.

  Returning to FIG. 10, the two-phase / three-phase converter 76 is the same as the two-phase / three-phase converter 76 shown in FIG. 15, and the correction value signals X′α and X′β are converted into three correction values. The signals are converted into signals Xu, Xv, and Xw. The conversion process performed by the two-phase / three-phase conversion unit 76 is expressed by the determinant shown in the above equation (4).

  The system command value signals Ku, Kv, Kw output from the system counter-part generating unit 72 and the correction value signals Xu, Xv, Xw output from the two-phase / three-phase conversion unit 76 are added, respectively, and the command value signal X 'u, X'v, X'w are calculated and input to the PWM signal generation unit 77.

  The PWM signal generation unit 77 performs a triangular wave comparison based on the input command value signals X′u, X′v, and X′w and a carrier signal generated as a triangular wave signal having a predetermined frequency (for example, 4 kHz). PWM signals Pu, Pv, Pw are generated by the method. In the triangular wave comparison method, the command value signals X′u, X′v, and X′w are respectively compared with the carrier signal. For example, when the command value signal X′u is larger than the carrier signal, the high level is obtained. A low-level pulse signal is generated as the PWM signal Pu. The generated PWM signals Pu, Pv, Pw are output to the inverter circuit 2.

  FIG. 11 is a diagram for explaining a simulation result.

  In the grid-connected inverter system A (see FIG. 10), the transfer function K shown in the above equation (25) is used as the transfer function of the α-axis current controller 74 and the β-axis current controller 75, and the unbalanced disturbance and Simulation was performed by adding fifth, seventh and eleventh harmonics. FIG. 11 shows current signals Iu, Iv, and Iw obtained by detecting the output current of each phase by the current sensor 5. As shown in the figure, the current signals Iu, Iv, and Iw are in an equilibrium state in which each harmonic component is suppressed and the positive phase component of the fundamental wave follows the target value. That is, the α-axis current controller 74 and the β-axis current controller 75 appropriately control the fundamental, fifth, seventh, and eleventh harmonics.

  In the present embodiment, the α-axis current controller 74 and the β-axis current controller 75 include a rotation coordinate conversion unit 78, LPFs 74a and 75a, PI control units 74b and 75b, a stationary coordinate conversion unit 79, and harmonic compensation shown in FIG. The same processing as that of the controller 800 (see FIG. 16) is performed. The processes performed by the α-axis current controller 74 and the β-axis current controller 75 are linear time-invariant processes. Further, the control circuit 7 does not include a rotation coordinate conversion process and a static coordinate conversion process, which are nonlinear time-varying processes, and the control circuit 7 performs control in a static coordinate system. That is, the entire current control system is a linear time invariant system. That is, the control system for current control and the control system for each harmonic compensation are set as one control system, and the control system is set as a linear time-invariant process. Therefore, linear control theory can be used for control system design, and control system design can be facilitated.

  In the above embodiment, the frequency weight W used for the design of the α-axis current controller 74 and the β-axis current controller 75 is described as being common. However, the α-axis current controller 74 and the β-axis current controller 75 Different frequency weights W may be used. For example, different real numbers k may be set for the α-axis current controller 74 and the β-axis current controller 75 to improve the speed response of the α-axis component, for example. It can also be designed to give additional properties such as increasing the stability of one.

  In the first embodiment, the case where the output current is controlled has been described, but the present invention is not limited to this. For example, the output voltage may be controlled. Hereinafter, the case of controlling the output voltage will be described as a second embodiment.

  FIG. 12 is a block diagram for explaining a control circuit according to the second embodiment. In the same figure, the same code | symbol is attached | subjected to the element which is the same as that of the grid connection inverter system A shown in FIG. 10, or similar.

  The inverter system A ′ shown in FIG. 12 is different from the grid-connected inverter system A (see FIG. 10) according to the first embodiment in that power is supplied to the load L instead of the power system B. Since it is necessary to control the voltage supplied to the load L, the control circuit 8 controls not the output current but the output voltage. The control circuit 8 is different from the control circuit 7 according to the first embodiment (see FIG. 10) in that a PWM signal is generated based on the voltage signal V input from the voltage sensor 6. The inverter system A ′ supplies power to the load L while controlling the output voltage to a target value by feedback control.

The three-phase / two-phase converter 83 converts the three voltage signals Vu, Vv, Vw input from the voltage sensor 6 into an α-axis voltage signal Vα and a β-axis voltage signal Vβ. The conversion process performed by the three-phase / two-phase conversion unit 83 is represented by a determinant represented by the following equation (26).
The voltage signals Vu, Vv, and Vw are phase voltage signals for each phase, but a line voltage signal may be detected and used. In this case, after converting the line voltage signal into the phase voltage signal, the determinant shown in the above equation (26) is used, or instead of the matrix shown in the above equation (26), the line voltage signal is converted to the α axis. The matrix may be converted into the voltage signal Vα and the β-axis voltage signal Vβ.

  The α-axis voltage controller 84 receives the deviation between the α-axis voltage signal Vα output from the three-phase / two-phase converter 83 and the α-axis voltage target value that is the target value of the fundamental component, and receives the correction value signal X ′. α is generated. The β-axis voltage controller 85 receives the deviation between the β-axis voltage signal Vβ output from the three-phase / two-phase converter 83 and the β-axis voltage target value that is the target value of the fundamental component, and receives the correction value signal X ′. β is generated. The α-axis voltage controller 84 and the β-axis voltage controller 85 are controllers designed by the H∞ loop shaping method using the frequency weight W shown in the above equation (24).

  Also in this embodiment, the α-axis voltage controller 84 and the β-axis voltage controller 85 perform linear time-invariant processing, and the control system for voltage control and the control system for each harmonic compensation are set as one control system. The control system is a linear time-invariant process. Therefore, the same effect as that of the first embodiment can be obtained.

  In the said 1st or 2nd embodiment, although the case where the control circuit which concerns on this invention was used for the grid connection inverter system (inverter system) was demonstrated, it is not restricted to this. The present invention controls, for example, an inverter circuit that performs harmonic compensation used in a power active filter, an unbalance compensation device, a static reactive power compensation device (SVC, SVG), an uninterruptible power supply (UPS), and the like. It can also be applied to circuits. Moreover, the present invention is not limited to controlling an inverter circuit that converts direct current to three-phase alternating current. For example, control circuits such as a converter circuit that converts three-phase alternating current to direct current and a cycloconverter that converts the frequency of three-phase alternating current Can be applied. Hereinafter, a case where the present invention is applied to a control circuit of a converter circuit will be described as a third embodiment.

  FIG. 13 is a block diagram for explaining a three-phase PWM converter system according to the third embodiment. In the same figure, the same code | symbol is attached | subjected to the element which is the same as that of the grid connection inverter system A shown in FIG. 10, or similar.

  A three-phase PWM converter system C shown in FIG. 13 converts AC power supplied from the power system B into DC power and supplies it to a load L ′. The load L ′ is a DC load. The three-phase PWM converter system C includes a transformer circuit 4, a filter circuit 3, a current sensor 5, a voltage sensor 6, a converter circuit 9, and a control circuit 7.

  The transformer circuit 4 boosts or steps down the AC voltage input from the power system B to a predetermined level. The filter circuit 3 removes a high frequency component from the AC voltage input from the transformer circuit 4 and outputs it to the converter circuit 9. The current sensor 5 detects the alternating current of each phase input to the converter circuit 9. The detected current signal I is input to the control circuit 7. The voltage sensor 6 detects an AC voltage of each phase input to the converter circuit 9. The detected voltage signal V is input to the control circuit 7. The converter circuit 9 converts the input AC voltage into a DC voltage and outputs it to the load L ′. The converter circuit 9 is a three-phase PWM converter, and is a voltage type converter circuit including three sets and six switching elements (not shown). The converter circuit 9 converts the input AC voltage into a DC voltage by switching each switching element on and off based on the PWM signal input from the control circuit 7. The converter circuit 9 is not limited to this, and may be a current type converter circuit.

  The control circuit 7 controls the converter circuit 9. The control circuit 7 generates a PWM signal and outputs it to the converter circuit 9 in the same manner as the control circuit 7 of the first embodiment. In FIG. 13, only the configuration for performing input current control and harmonic compensation is described, and the configuration for other control is omitted. Although not shown, the control circuit 7 also includes a DC voltage controller and a reactive power controller, and also controls the output voltage and the input reactive power. The control method performed by the control circuit 7 is not limited to this. For example, when the converter circuit 9 is a current-type converter circuit, output current control may be performed instead of output voltage control.

  Also in this embodiment, the same effect as in the case of the first embodiment can be obtained. That is, in the three-phase PWM converter system C, harmonic compensation is required to suppress harmonics of the input current, but in this embodiment, a control system for current control and each harmonic compensation are used. By using the control system as one control system and making the control system a linear time-invariant process, the control system design can be facilitated using linear control theory.

  The configuration of the three-phase PWM converter system C is not limited to the above. For example, the control circuit 8 may be used instead of the control circuit 7. Alternatively, an inverter circuit may be provided on the output side of the converter circuit 9 so as to convert the DC power into AC power and supply the AC load to the AC load.

  The control circuit according to the present invention, the grid interconnection inverter system using the control circuit, and the three-phase PWM converter system are not limited to the above-described embodiments. The specific configuration of each part of the control circuit according to the present invention, the grid-connected inverter system using the control circuit, and the three-phase PWM converter system can be variously modified.

A Grid-connected inverter system A 'Inverter system 1 DC power supply 2 Inverter circuit (power conversion circuit)
3 Filter circuit 4 Transformer circuit 5 Current sensor 6 Voltage sensor 7 and 8 Control circuit 72 System counter component generation unit 73 and 83 Three-phase / two-phase conversion unit 74 α-axis current controller (control means)
75 β-axis current controller (control means)
84 α-axis voltage controller (control means)
85 β-axis voltage controller (control means)
76 Two-phase / three-phase converter 77 PWM signal generator 9 Converter circuit (power converter circuit)
B Power system C Three-phase PWM converter system L, L 'Load

Claims (8)

A control circuit that controls driving of a plurality of switching means in a power conversion circuit relating to three-phase alternating current using a PWM signal,
Three-phase two-phase conversion means for converting three signals based on the three-phase output or input of the power conversion circuit into a first signal and a second signal;
Deviation signal generating means for generating a first deviation signal and a second deviation signal, which are deviations between the first signal and the second signal and the target values of the respective fundamental wave components;
The fundamental wave component included in each of the first deviation signal and the second deviation signal is controlled to zero, and predetermined harmonics included in the first deviation signal and the second deviation signal, respectively. Control means for generating a first correction value signal and a second correction value signal for suppressing the component;
Two-phase three-phase conversion means for converting the first correction value signal and the second correction value signal into three correction value signals;
PWM signal generating means for generating a PWM signal based on the three correction value signals;
With
The control means performs the signal processing on the first deviation signal and the second deviation signal by a transfer function designed by a robust control design method using the frequency weight W, whereby the first correction value signal And generating the second correction value signal,
When the angular frequency of the fundamental wave of the three-phase alternating current is set to ω ₀ and the nth harmonic is suppressed, the denominator of the frequency weight W is (s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } included,
A control circuit characterized by that. A control circuit that controls driving of a plurality of switching means in a power conversion circuit relating to three-phase alternating current using a PWM signal,
Deviation signal generating means for generating three deviation signals that are deviations between the three signals based on the three-phase outputs or inputs of the power conversion circuit and the target values of the respective fundamental wave components;
Three-phase to two-phase conversion means for converting the three deviation signals into a first deviation signal and a second deviation signal;
The fundamental wave component included in each of the first deviation signal and the second deviation signal is controlled to zero, and predetermined harmonics included in the first deviation signal and the second deviation signal, respectively. Control means for generating a first correction value signal and a second correction value signal for suppressing the component;
Two-phase three-phase conversion means for converting the first correction value signal and the second correction value signal into three correction value signals;
PWM signal generating means for generating a PWM signal based on the three correction value signals;
With
The control means performs the signal processing on the first deviation signal and the second deviation signal by a transfer function designed by a robust control design method using the frequency weight W, whereby the first correction value signal And generating the second correction value signal,
When the angular frequency of the fundamental wave of the three-phase alternating current is set to ω ₀ and the nth harmonic is suppressed, the denominator of the frequency weight W is (s ² + ω ₀ ² ) · {s ² + (n · ω ₀ ) ² } included,
A control circuit characterized by that. The frequency weight W is a predetermined real number k,
The control circuit according to claim 1, wherein   The control circuit according to claim 1, wherein the three signals are signals obtained by detecting an output current or an input current of each phase.   The control circuit according to claim 1, wherein the three signals are signals obtained by detecting an output voltage or an input voltage of each phase.   The control circuit according to claim 1, wherein the transfer function is designed by an H∞ loop shaping method using a frequency weight W.   A grid-connected inverter system comprising an inverter circuit and the control circuit according to any one of claims 1 to 6.   A three-phase PWM converter system comprising a converter circuit and the control circuit according to any one of claims 1 to 6.