JPH10108474A - Multilevel switching type power converter - Google Patents

Abstract

PROBLEM TO BE SOLVED: To surely reduce a loss of a multi-level switching-type power converter with a simplified structure. SOLUTION: Series-connected first to fourth self-extinguishing type switching elements (IGBT 7-10) are connected in parallel to a DC power supply 1, and a pulse potential which changes between positive potential +V and neutral point potential 0V can be output from an output terminal 6 through repetition of the on state of IGBT 7, 8 and on stage of IGBT 8, 9. Moreover, a pulse potential which changes between negative potential -V and neutral point potential 0V can also be output from the output terminal 6, through repetition of the on state of IGBT 9, 10 and on state of IGBT 8, 9. In this case, since the characteristic is set so that the one voltage of IGBT 8, 9 having less number of times of the switching operation is lower than that of IGBT 7, 10, loss of arms 4, 5 can be reduced by reducing normal loss of IGBT 8, 9.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a multi-level switching type power converter for outputting a pulse potential varying between predetermined potentials from an output terminal connected to an upper arm and a lower arm comprising switching elements connected in series. About.

[0002]

2. Description of the Related Art In a conventional neutral point clamped inverter of PWM control, a loss generated by a self-extinguishing type switching element constituting an inverter includes a steady loss (on voltage at normal time × conducting current) and a switching loss. There is an integrated value （of (current supplied × voltage between main circuit terminals) between a period when the element shifts from off to on and a period when the element shifts from on to off.

[0003] In this type of neutral point clamp type inverter, the number of switching times for each switching element,
Since the operation periods are different, the switching loss is also different for each switching element.

Accordingly, Japanese Patent Application Laid-Open No. Hei 5-3287 shown in FIG.
No. 31 has been proposed. This converter converts the DC power of the main DC power supplies E1 and E2 connected in series into AC power. The switching circuits S1 to S4 are connected in series between the main DC power supplies E1 and E2, and the switching circuit S1 and the switching circuit S1 are switched. A DC / DC converter CNV1 for generating an output voltage E3 is connected to a connection point with the circuit S2 via a diode D1 having a polarity shown in FIG.
A DC / DC converter CNV2 for generating an output voltage E4 is connected to a connection point between the switching circuit 3 and the switching circuit S4 via a diode D2 having the polarity shown in the figure. in this case,
Assuming that the voltages of the main DC power supplies E1 and E2 are E / 2, the outputs of the DC / DC converters CNV1 and CNV2 are controlled around E / 4.

When the difference between the modulation voltage proportional to the modulation current flowing through the load according to the output voltage and the DC power supply voltage is small, the output voltages E3 and E4 of the DC / DC converters CNV1 and CNV2 are changed to E / 4. By further increasing, the switching frequency of each of the switching circuits S1 to S4 is equalized, and the switching loss is equalized. If the difference between the output voltage and the DC power supply voltage is large, the output voltages E3 and E3 of the DC / DC converters CNV1 and CNV2 are used.
By making 4 smaller than E / 4, the sum of the switching losses of the switching circuits S1 and S4 and the sum of the switching losses of the switching circuits S2 and S3 are equalized. Accordingly, it is possible to prevent the switching element from being destroyed due to the unbalanced switching loss.

[0006]

However, the one disclosed in Japanese Patent Application Laid-Open No. Hei 5-328873 discloses a DC / DC converter CNV1, CNV according to the difference between the modulation voltage and the DC power supply voltage.
2 is configured to increase and decrease the output voltages E3 and E4 around E / 4, so that the DC / DC converters CNV1 and CNV2
Has the disadvantage that the configuration is complicated and the cost is high. Also,
The steady loss is not improved at all, and the loss is not sufficiently reduced.

SUMMARY OF THE INVENTION The present invention has been made in view of the above circumstances, and has as its object to provide a multi-level switching power converter which can reliably reduce the loss of a self-extinguishing type switching element with a simple configuration. It is in.

[0008]

According to the first aspect of the present invention, a connection between the upper arm and the lower arm is set at a specific potential between the upper arm and the lower arm in accordance with ON / OFF of the switching elements constituting the upper arm and the lower arm. A changing potential is output.

Here, at least one of the plurality of switching elements constituting each arm is:
Since different types or different characteristics are set for other switching elements in the same arm, the sum of switching loss and steady state loss in each arm can be reduced.

According to the second aspect of the present invention, as the switching element becomes closer to the connection point between the upper arm and the lower arm,
The ON voltage is set small. The closer the switching element is to the connection point between the upper arm and the lower arm, the smaller the number of switching operations and the longer the on-time, so that the above-described configuration can reduce the steady-state loss. Further, the switching loss does not increase so much because the number of times of switching is small. As a result, the sum of the switching loss and the steady loss in each arm can be reduced.

According to the third aspect of the present invention, as the switching element becomes farther from the connection point between the upper arm and the lower arm,
Switching speed is set fast. The closer the switching element is to the connection point between the upper arm and the lower arm, the smaller the number of switching operations and the longer the on-time, so that the switching loss can be reduced as compared with the above configuration. Further, the steady loss does not increase so much because the ON time is short.
As a result, the sum of the switching loss and the steady loss in each arm can be reduced.

According to the fourth aspect of the present invention, the ON voltage of the IGBT can be set low by the lime time control. Therefore, the loss is reduced by reducing the ON voltage of the predetermined IGBT constituting each arm. be able to.

According to the fifth aspect of the present invention, the MOSFE
T is a unipolar element and has a shorter switching time than an IGBT, so that it can be used as an element for reducing switching loss.

According to the sixth aspect of the present invention, the ON voltage can be set low by increasing the effective area of the switching element, so that the ON voltage of the switching element can be reduced to reduce the loss.

According to the seventh aspect of the present invention, since the built-in (parasitic) diode of the MOSFET can be used as the freewheel diode, the whole can be miniaturized.

According to the eighth aspect of the present invention, since the heating values of the plurality of switching elements are set to be substantially equal to each other, the arrangement efficiency of the switching elements can be increased and the device can be downsized. .

[0017]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A schematic configuration in which the present invention is applied to a single-phase neutral point-clamped inverter (a multilevel switching inverter having three levels) will be described below with reference to FIGS.

In FIG. 1, a DC power supply 1 has first and second capacitors 2, 3 connected in series and having the same capacity.
Are connected in parallel, and the DC voltage of the DC power supply 1 is divided by the capacitors 2 and 3 so that three levels of potentials (positive potential of DC voltage + V, neutral point potential which is an intermediate voltage of DC voltage) 0V and a negative potential -V of a DC voltage) are generated and output from the potential lines L1, L2, L3, respectively.

On the other hand, a self-extinguishing first to fourth switching element S is provided between potential line L1 and potential line L3.
1 to S4 are connected in series. Also, first to fourth flywheel diodes D1 to D4 are connected in anti-parallel to each of the switching elements S1 to S4. In this case, the upper arm 4 is formed by the first and second switching elements S1, S2, and the third and fourth switching elements S1
The lower arm 5 is formed by 3 and S4. The output terminal 6 is connected to a connection point between the upper arm 4 and the lower arm 5.

Here, a connection point (neutral point) of the capacitors 2 and 3 is connected to a connection point of the first and second switching elements S1 and S2 via a first clamping diode D5 having the illustrated polarity, and The third and fourth switching elements S3 and S4 are connected via a second clamping diode D6 having the illustrated polarity.
Connected to the connection point. A load (not shown) is connected between the output terminal 6 and a connection point (neutral point) between the capacitors 2 and 3.

Now, the inventor of the present application has determined that the switching elements S1 to S4 of the neutral point clamp type inverter shown in FIG.
, And the switching waveforms of the diodes D1 to D6 were analyzed, and the conditions under which the loss was minimized for each of the switching elements S1 to S4 and each of the diodes D1 to D6 were examined. Here, FIG. 2 schematically shows switching waveforms of the switching elements S1 to S4 and the diodes D1 to D6 under typical conditions.

A broken line in FIG. 2A indicates an output voltage command applied to an inverter (not shown), a solid line indicates a current flowing through the load, and a pulse waveform indicates a pulse potential applied to the output terminal 6. I have. Also, FIG.
The broken line in (k) indicates the voltage between the terminals of each element, and the solid line indicates the flowing current.

FIG. 3 shows the switching elements S1 to S4 and the diodes D1 to D6 when the switching elements S1 to S4 and the diodes D1 to D6 are identical under typical operating conditions as shown in FIG. The calculation results of the losses are shown respectively.

As a result of the calculation, the first and fourth switching elements S1 and S4 become the second and third switching elements S2 and S4.
3, while the switching loss of the second and third switching elements S2 and S3 was larger than that of the first and fourth switching elements S1 and S4. . From this, it is necessary to use one having a small switching loss as the first and fourth switching elements S1 and S4, and use one having a small steady-state loss as the second and third switching elements. It is found to be effective in reducing the total loss.

FIGS. 4 to 9 show a first embodiment of the present invention. First to fourth switching elements S1 to S1 are shown in FIGS.
This is a specific example in which the first to fourth IGBTs 7 to 10 are used as S4, and the ON voltages of the second and third IGBTs 8 and 9 are set to be lower than those of the first and fourth IGBTs 7 and 10. It is characterized by. The first to fourth flywheel diodes D1 to D4 are shown corresponding to the reference numerals 11 to 14, and the first and second clamping diodes D5, D
6 corresponds to reference numerals 15 and 16, and the potential line L1
To L3 are shown corresponding to the reference numerals 101 to 103.

Here, the first to fourth switching elements S
The IGBT is used as 1 to S4 because the MOSF
The on-voltage of a unipolar element represented by ET is determined by the required breakdown voltage and current, while the IGB
This is because the ON voltage of T can be arbitrarily selected by minority carrier lifetime control such as electron beam irradiation.
In this case, as shown in FIG. 5, a bipolar element such as an IGBT can reduce the on-voltage as more minority carriers flow, but has a trade-off characteristic in that the switching speed decreases accordingly.

Accordingly, as a result of setting the ON voltage of the second and third IGBTs 8 and 9 lower than the ON voltage of the first and fourth IGBTs 7 and 10, the second and third IGBTs 8 and 9 are The switching speed is lower than that of the fourth IGBTs 7 and 10.

FIG. 6 shows the first to fourth IGBTs 7.
10 shows the relationship between the switching time of 10 and the total loss of each element. From FIG. 6, the first and fourth IGBTs 7,
By adopting a high-speed type with a switching time of about 0.2 μs for 10 and a low-loss type with a switching time of about 1.0 μs as the second and third IGBTs 8 and 9, the total loss can be minimized. I understand.

Next, the operation of the above configuration will be described with reference to each of the IGBTs 7 to 1.
A description will be given with reference to FIG. When the output voltage command value (shown by a dashed line in (a) of the figure) is positive ... The inverter control device (not shown)
A predetermined IGBT is turned on and off in order to output a pulse potential that changes between the positive potential + V and the neutral point potential 0 V (periods indicated by A and D in FIG. 2). That is, at the timing when the positive potential + V is output to the output terminal 6, the first and second IGs are output.
At the timing when the BTs 7 and 8 are turned on and the neutral point potential 0 V is output to the output terminal 6, the second and third IGBTs 8 and 9 are turned on.

In this case, at the timing when the first and second IGBTs 7 and 8 are turned on during the period indicated by A in FIG.
A current flows to the load through the first and second IGBTs 7 and 8 (see FIG. 7A), and at the timing when the second and third IGBTs 8 and 9 are turned on, the first clamping diode 1 is turned on.
5, a current flows to the load through the second IGBT 8 (see FIG. 8A).

In typical operating characteristics, the phase of the voltage applied to the load and the phase of the current flowing through the load are shifted from each other. (Period indicated by D in FIG. 2).

In this case, when the first and second IGBTs 7 and 8 are turned on, a current flows from the load through the first and second freewheel diodes 11 and 12 (see FIG. 7B). Also, the second and third IGBTs 8,
9 is turned on, the third IGBT 9 and the second IGBT 9
A current flows from the load through the clamping diode 16 (see FIG. 8B).

According to the above operation, during the period when the output voltage command value is positive, a potential displacing between the positive potential + V and the neutral potential 0V is applied to the output terminal 6, and the current is applied to the load accordingly. Flows.

When the output voltage command is negative: an inverter control device (not shown) outputs a predetermined IGBT to the output terminal 6 in order to output a pulse potential displacing between the negative potential -V and the neutral potential 0V. It is turned on and off (periods indicated by B and C in FIG. 2). That is, the third and fourth IGBTs 9 and 10 are turned on at the timing of outputting the negative potential −V to the output terminal 6, and the second and third IGBTs are output at the timing of outputting the neutral point potential 0 V to the output terminal 6. The IGBTs 8 and 9 are turned on.

In this case, at the timing when the third and fourth IGBTs 8 and 9 are turned on during the period indicated by C in FIG.
A current flows from the load through the third and fourth IGBTs (see FIG. 9B), and when the second and third IGBTs 8 and 9 are turned on, the current flows through the third IGBT 9 and the second clamping diode 16. Current flows from the load (see FIG. 8B).

In a typical operating characteristic, the phase of the voltage applied to the load is out of phase with the phase of the current flowing through the load. (The period indicated by B in FIG. 2).

In this case, the third and fourth IGBTs 9, 10
At the timing when is turned on, a current flows to the load through the third and fourth freewheel diodes 13 and 14 (see FIG. 9A). Also, the second and third IGBTs 8,
At the timing when 9 turns on, a current flows through the load through the first clamping diode 15 and the second IGBT 8 (see FIG. 8A).

By the above operation, during the period when the output voltage command value is negative, the potential which changes between the negative potential -V and the neutral point potential 0 V is applied to the output terminal 6, and the load is accordingly applied to the load. Electric current flows.

In this embodiment, the second and third IGBs
Since the on-voltage is set to be lower than that of the first and fourth IGBTs 7 and 10 by minority carrier lifetime control at T8 and T9, the steady-state loss of the second and third IGBTs 8 and 9 is reduced to reduce the on-voltage. The total loss of the arms 4 and 5 can be reduced. In this case, even if the switching speed of the IGBTs 8 and 9 decreases with a decrease in the ON voltage of the second and third IGBTs 8 and 9 due to the trade-off relationship shown in FIG. 9
Of the second and third I
The upper arm 4 and the lower arm 5 irrespective of an increase in switching loss due to a decrease in the switching speed of the GBTs 8 and 9.
Loss can be reduced.

According to the above configuration, in the neutral point clamp type inverter mainly composed of the first to fourth IGBTs 7 to 10 connected in series, the second and third IGBTs 8 and 9 having a small number of switching times are provided. Since 9 is set to have a characteristic in which the on-state voltage is lower than that of the first and fourth IGBTs 7 and 10, the steady-state loss of the second and third IGBTs 8 and 9 can be reduced. Therefore, unlike the conventional example in which the same element is used as the first to fourth switching elements, the loss of each of the arms 4 and 5 can be significantly reduced.

FIGS. 10 to 14 show a second embodiment of the present invention, in which the present invention is applied to a four-level multilevel inverter. That is, the first to third capacitors 22 connected in series and having the same capacity are connected to the DC power supply 21.
To 24 are connected in parallel.
24, the DC voltage of the DC power supply 21 is divided to generate four levels of potentials (positive potential V4 of DC voltage, intermediate potentials V3 and V2 of DC voltage, and negative potential V1 of DC voltage). To 107 are output.

First to sixth IGBTs 25 to 30 are connected in series between the potential line 104 and the potential line 107. In addition, first to sixth flywheel diodes 31 to 36 are reversely connected to the first to sixth IGBTs 25 to 30, respectively. In this case, the first to third IGBTs 2
An upper arm 37 is formed by 5 to 27, and a lower arm 38 is formed by fourth to sixth IGBTs 28 to 30. An output terminal 40 is connected to a connection point between the upper arm 37 and the lower arm 38.

Here, the potential line 104 is connected to the first IGB
The connection point of the first and second capacitors 22 and 23 is connected to the collector of the second IGBT 26 via the first clamping diode 40 of the illustrated polarity and the third connection point of the illustrated IGBT 26 is connected to the collector of the T25. Clamping diode 42
Is connected to the emitter of the fourth IGBT 28, and the connection point between the second and third capacitors 23 and 24 is connected to the third IGB via a second clamping diode 41 having the illustrated polarity.
The collector of T27 is connected to the collector of the sixth IGBT 30 via the fourth clamping diode 43 of the polarity shown, and the potential line 107 is connected to the sixth I
It is connected to the emitter of the GBT 30.

In the second embodiment, similarly to the first embodiment, the characteristics are controlled so that the total loss of each switching element is minimized, and the first and sixth IGBs are controlled.
T25 and 30 are high-speed types with high switching speed, and third and fourth IGBTs 27 and 28 are low-loss types with low on-voltage.

An inverter control device (not shown)
By turning on and off a predetermined IGBT, V4 and V3
A pulse potential displacing between V3 and V2, a pulse potential displacing between V3 and V2, and a pulse potential displacing between V2 and V1 are output to an output terminal 39.

Outputting the voltage V4 to the output terminal ...
The first to third IGBTs 25 to 27 are turned on. In this case, according to the difference between the phase of the voltage applied to the load and the phase of the current, a state in which current flows through the load through the first to third IGBTs 25 to 27 (see FIG. 11A), ~
A state in which current flows from the load through the third flywheel diodes 31 to 33 occurs (see FIG. 3B).

Outputting the voltage V3 to the output terminal ...
The second to fourth IGBTs 26 to 28 are turned on. In this case, according to the difference between the phase of the voltage applied to the load and the phase of the current, a state in which a current flows to the load through the first clamping diode 29 and the second and third IGBTs 26 and 28 (FIG. 12). (A)), the fourth IGBT 28, the third IGBT 28
And a state in which a current flows from the load through the clamping diode 42 (see FIG. 3B).

When outputting the voltage V2 to the output terminal ...
The third to fifth IGBTs 27 to 29 are turned on. In this case, the second clamping diode and the third IGBT are used in accordance with the difference between the phase of the voltage applied to the load and the phase of the current.
13 (a)) and a state where current flows from the load through the fourth and fifth IGBTs 28 and 29 and the fourth clamping diode 43 (FIG. 13 (b)). reference).

When outputting the voltage V1 to the output terminal ...
The fourth to sixth IGBTs 28 to 30 are turned on. In this case, the fourth to sixth flywheel diodes 34 to 34 are set in accordance with the difference between the phase of the voltage applied to the load and the phase of the current.
The state in which a current flows to the load through 36 (FIG. 14A)
), And a current flows from the load through the fourth to sixth IGBTs 28 to 30 (see FIG. 3B).

According to the second embodiment, the first and sixth IGBTs 25, 3 having a larger switching loss than the others.
Since a high-speed type is used as 0 and a low on-voltage is used as the third and fourth IGBTs 27 and 28 having a larger steady-state loss than the others, the total loss of each arm 37 and 38 is reduced. be able to.

FIGS. 15 to 20 show a third embodiment of the present invention, in which the present invention is applied to a five-level inverter. That is, the first to fourth capacitors 52 to 55 having the same capacity and connected in series are connected in parallel to the DC power supply 51, and the DC voltage of the DC power supply 51 is divided by the capacitors 52 to 55. 5 level potentials (DC voltage positive potential V5, DC voltage intermediate potentials V4, V3, V
2. The negative potential V1) of the DC voltage is generated, and each potential line is output from 108 to 112 respectively.

The first to eighth IGBTs 56 to 63 are connected in series between the potential line 108 and the potential line 112. Further, the first to eighth flywheel diodes 64 to 71 are reversely connected to the first to eighth IGBTs 56 to 63, respectively. In this case, the first to fourth IGBTs 5
The upper arm 72 is formed by 6 to 59, and the lower arm 73 is formed by the fifth to eighth IGBTs 60 to 63. An output terminal 74 is connected to a connection point between the upper arm 72 and the lower arm 73.

Here, the potential line 108 is connected to the first IGB
The connection point of the first and second capacitors 52 and 53 is connected to the collector of the second IGBT 57 via a first clamping diode 75 of the illustrated polarity, and the fourth connection point of the illustrated IGBT 57 is connected to the collector of the T56. Clamping diode 78
Is connected to the collector of the sixth IGBT 61, and the connection point of the second and third capacitors 53 and 54 is connected to the third IGBT 61 through the second clamping diode 76 having the illustrated polarity.
The collector is connected to the collector of T58 and to the collector of the seventh IGBT 62 via the fifth clamping diode 79 of the illustrated polarity, and the third and fourth capacitors 54 and 55 are connected.
Is connected to a third clamping diode 77 having the illustrated polarity.
And the collector of the eighth IGBT 63 via the sixth clamping diode 80 of the illustrated polarity, and the potential line 112 is connected to the emitter of the eighth IGBT 63. Have been.

In the third embodiment, as in the first embodiment, the switching element farther from the output terminal 74 is of a high-speed type so that the total loss of each switching element is minimized, and the switching element closer to the output terminal 74 is on. A low-loss type with a low voltage was used.

Now, an inverter control device (not shown)
By turning on and off a predetermined IGBT, V5 and V4
Pulse potential (see FIGS. 16 and 17),
4 and V3 (see FIGS. 17 and 18), and a pulse potential that varies between V3 and V2 (see FIG. 1).
8 and 19), and a pulse potential (see FIGS. 19 and 20) displaced between V2 and V1 is output to the output terminal 74. Note that a detailed description of the operation is omitted for simplification of the description.

According to the third embodiment, each arm 62,
In 63, a switching element farther from the output terminal 74 is of a high-speed type, and a switching element closer to the output terminal 74 is of a low-loss type having a lower on-voltage, so that the loss of each of the arms 62 and 63 can be reduced. it can.

FIG. 21 shows a fourth embodiment of the present invention. The same parts as those in the first embodiment are denoted by the same reference numerals and description thereof is omitted. This fourth embodiment is characterized in that MOSFETs are used as the first and fourth switching elements S1 and S4. That is, the first arm 4 is connected to the MOSFET 81
And the IGBT 8 are connected in series, and the second arm 5 is formed by connecting the IGBT 9 and the MOSFET 82 in series.

In the case of the fourth embodiment, the MOSFET 8
Since IGBTs 1, 82 are unipolar elements, IGBTs
Since the switching time is shorter than 8 and 9, the total loss of each arm 4 and 5 can be reduced by using a MOSFET as a switching element having a large number of switching times to reduce switching loss.

The present invention is not limited to the above embodiment, but can be modified or expanded as follows. The ON voltage may be set low by making the effective area of the switching element larger than the other switching elements. When a MOSFET is used as the switching element, a built-in (parasitic) diode of the MOSFET may be used as the freewheel diode.
In this case, there is no need to connect a fly-hole diode to the MOSFET, so that the overall configuration can be reduced in size.

The characteristics or types of the switching elements may be set so that the heat values of the switching elements are substantially equal. In this case, since the amount of heat generated by each switching element can be made uniform, the arrangement efficiency of the self-extinguishing element can be increased, and the device can be downsized.

A plurality of DC power supplies may be connected in series, and a predetermined potential may be applied to the switching element from a common connection point via a clamping diode. The present invention may be applied to an electric vehicle inverter, a general-purpose inverter, an uninterruptible power supply, and an AC servo controller.

[Brief description of the drawings]

FIG. 1 is an electric circuit diagram schematically showing the configuration of the present invention.

FIG. 2 is a timing chart showing an output voltage and a voltage between terminals of each element.

FIG. 3 is a diagram showing a relationship between each element and loss.

FIG. 4 is a view corresponding to FIG. 1, showing a first embodiment of the present invention;

FIG. 5 is a diagram showing a relationship between a switching time and an on-voltage.

FIG. 6 is a diagram showing a relationship between switching time and total loss of each element.

FIG. 7 is a diagram corresponding to FIG. 4, showing an energized state when a positive potential is applied to a load;

FIG. 8 is a diagram corresponding to FIG. 4, showing an energized state when 0 V is applied to a load;

FIG. 9 is a diagram corresponding to FIG. 4, showing an energized state when a negative potential is applied to a load.

FIG. 10 is a view corresponding to FIG. 4, showing a second embodiment of the present invention;

FIG. 11 is a view corresponding to FIG. 10 and shows an energized state when a predetermined potential is applied to a load.

FIG. 12 is a diagram corresponding to FIG. 10 showing an energized state when a different predetermined potential is applied to a load;

FIG. 13 is a diagram corresponding to FIG. 10, showing a current-carrying state when a different predetermined potential is applied to a load;

FIG. 14 is a diagram corresponding to FIG. 10 showing an energized state when a different predetermined potential is applied to a load;

FIG. 15 is a view corresponding to FIG. 4, showing a third embodiment of the present invention.

FIG. 16 is a diagram corresponding to FIG. 15, showing an energized state when a predetermined potential is applied to a load.

FIG. 17 is a diagram corresponding to FIG. 15, showing an energized state when different predetermined potentials are applied to a load;

FIG. 18 is a diagram corresponding to FIG. 15, showing an energized state when a different predetermined potential is applied to a load.

FIG. 19 is a diagram corresponding to FIG. 15, showing an energized state when different predetermined potentials are applied to a load.

FIG. 20 is a diagram corresponding to FIG. 15, showing an energized state when different predetermined potentials are applied to a load;

FIG. 21 is a view corresponding to FIG. 1, showing a fourth embodiment of the present invention.

FIG. 22 shows a conventional example and is equivalent to FIG.

[Explanation of symbols]

1 is a DC power supply, 2 and 3 are capacitors, 4 is an upper arm, 5
Is a lower arm, 6 is an output terminal, 7 to 10 are IGBTs (self-extinguishing switching elements), 11 to 14 are freewheel diodes, 15 and 16 are clamping diodes, 2
1 is a DC power supply, 22 to 24 are capacitors, 25 to 30 are IGBTs (self-turn-off type switching elements), 31 to 36
Is a freewheel diode, 37 is an upper arm, 38 is a lower arm, 39 is an output terminal, 40 to 43 are clamping diodes, 51 is a DC power supply, 52 to 55 are capacitors,
56 to 63 are IGBTs (self-turn-off switching elements), 64 to 71 are freewheel diodes, 72 is an upper arm, 73 is a lower arm, 74 is an output terminal, and 75 to 8
0 is a clamping diode, 81 and 82 are MOSFETs
(Self-extinguishing type switching element) 101 to 112 are potential lines.

Claims (8)

[Claims] 1. A plurality of potential lines to which a plurality of potentials are respectively applied, and one end connected to the highest potential line among the plurality of potential lines, and a plurality of self-extinguishing type switching elements connected in series. An upper arm having an inter-element connection point between each switching element, one end connected to the lowest potential line of the plurality of potential lines, and the other end connected to the upper arm and a connection point. A lower arm having the same number of self-extinguishing type switching elements as the switching elements of the upper arm connected in series, and having a connection point between elements between the switching elements; and the potential in the upper arm. In order from the inter-element connection point on the side closer to the line, to the inter-element connection point on the side farther from the potential line in the lower arm at the inter-arm connection point in order, the inter-arm connection point A clamping diode connected to a potential line of a predetermined potential between the highest potential and the lowest potential of the plurality of potential lines, and the plurality of switching elements of the upper arm and the lower arm A multi-level switching power converter that outputs one of the plurality of potentials to a connection point between the upper arm and the lower arm by operating a predetermined switching element, wherein each arm is configured. At least one switching element of the plurality of switching elements is set to a different type or a different characteristic from other switching elements in the same arm so that the switching loss and the steady loss in each arm are both reduced. A multilevel switching power converter. 2. The on-voltage of the plurality of switching elements constituting each arm is set to be smaller as the switching element is closer to a connection point between the upper arm and the lower arm. 2. The multi-level switching power converter according to claim 1. 3. The switching speed of the plurality of switching elements forming each arm is set to be higher as the switching element is farther from a connection point between the upper arm and the lower arm. 3. The multi-level switching power converter according to 1 or 2. 4. The multi-level switching device according to claim 1, wherein said plurality of switching elements forming each arm are formed of IGBTs, and said IGBTs are controlled in on-voltage by lifetime control. Power converter. 5. The multi-level switching power converter according to claim 1, wherein said plurality of switching elements constituting each arm include an IGBT and a MOSFET. 6. An effective area, which is an area through which a current flows, is set to be larger as the plurality of switching elements forming each arm are closer to a connection point between the upper arm and the lower arm. The multi-level switching power converter according to any one of claims 1 to 5, wherein 7. The multi-level switching according to claim 1, wherein said plurality of switching elements constituting each arm are composed of a MOSFET, and have a freewheel diode composed of a parasitic diode of the MOSFET. Power converter. 8. The multi-level switching power converter according to claim 1, wherein the plurality of switching elements are set so that heat generation amounts are substantially equal to each other.