JPH0993940A - Power supply circuit and switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve the power conversion efficiency of a wide range power supply circuit which is especially decreased when it is applied to an AC 100V system and, further, simplify a circuit structure and realize the reduction of size/weight. SOLUTION: When an AC voltage not higher than 150V is doubled and rectified, a circuit in which a rectified current flows through the parallel circuit of rectifying diodes Da-Dc and Db-Dd is composed. When an AC voltage not lower than 150V is full-wave rectified, rush current limiting resistors RiA and RiB are connected to the AC line in parallel and, when the AC voltage is not higher than 150V, the circuit is so switched as to eliminate the rush current limiting resistors RiA and RiB. The switching operations of the rectifier and the rush current limiting resistors are interlocked with each other and performed by a solenoid relay RL-11 having 2-contact switches S11 and S12 .

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a so-called wide range power supply circuit and a switching power supply circuit which are compatible with AC input voltage of AC100V system and AC200V system, for example.

[0002]

2. Description of the Related Art In recent years, by developing a switching element capable of withstanding a relatively large current and voltage of high frequency as a power supply circuit, most of the power supply devices for rectifying a commercial power supply to obtain a desired DC voltage are switching. It is a power supply device of the method. The switching power supply circuit is used as a power supply for various electronic devices as a high-power DC-DC converter, as well as reducing the size of transformers and other devices by increasing the switching frequency.

Further, as a switching power supply circuit, for example, a so-called wide range power supply adapted to correspond to an AC input voltage range of AC80V to 288V so as to correspond to an AC input voltage AC100V system region and an AC200V system region. The circuit is known. As a power supply circuit compatible with such a wide range, for example, the operation of the rectifying / smoothing circuit unit is switched between the case of AC100V system and the case of AC200V system, and the voltage doubler rectification operation is performed in the case of AC100V system.
In the case of an AC200V system, by performing a normal rectification operation such as full-wave rectification, a nearly constant DC voltage (rectification smoothed voltage) can be obtained regardless of the level of the AC input voltage, and a switching converter in the subsequent stage can be provided. Those configured to supply are known.

The circuit diagram of FIG. 12 shows an example of a wide-range switching power supply circuit capable of switching the operation of the rectifying / smoothing circuit section as described above. In the switching power supply circuit shown in this figure, a common mode choke coil CMC and an across capacitor C _L are provided as a noise filter for removing common mode noise with respect to the commercial AC power supply AC. AC
A main switch MS for turning on / off the power is provided in the line, and similarly, an inrush current limiting resistor Ri is inserted in the AC line to suppress the inrush current flowing in the smoothing capacitor when the power is turned on. ing. A bridge rectifier circuit D ₁ is provided for the commercial AC power supply AC. The bridge rectifier circuit D ₁ is, for example, four rectifier diodes Da and D in a broken line forming bridge rectification.
b, Dc, and Dd are stacked and configured as one component.

In this power supply circuit, two smoothing capacitors Ci _A and Ci _B are connected in series, and are connected between the positive output terminal of the bridge rectifier circuit D ₁ (connection point of the rectifier diodes Da and Db) and the primary side ground. It is provided to be inserted. In this case, the connection point of the smoothing capacitors Ci _A and Ci _B is connected to the negative input terminal (connection point of the rectification diodes Dc and Dd) of the bridge rectification circuit D ₁ via the switch circuit 4 described later. Then, the rectified and smoothed voltage Ei obtained across the smoothing capacitors Ci _A and Ci _B connected in series is input to the switching converter SC in the subsequent stage. The switching converter SC performs a switching operation based on the input rectified and smoothed voltage Ei, and outputs a stabilized secondary side DC output voltage E _O.

The switch circuit 4 shown by a broken line is for switching between the voltage doubler rectifying and smoothing operation and the normal rectifying and smoothing operation. The switch circuit 4 is a hybrid IC including, for example, a bidirectional thyristor such as a triac TA as a switch element as shown in the figure, and an on / off control circuit for the triac TA. It In this case, the resistor R ₁₁ , the capacitors C _{11 to} C ₁₄ , and the diode D ₁₁ are connected to the switch circuit 4 as external parts as shown in the figure. Then, in these external parts, a detection voltage obtained by converting the AC input voltage into a direct current is obtained by the half-wave rectifier circuit including the capacitor C ₁₄ and the diode D ₁₁ . Then, in the switch circuit 4, based on this detected voltage, for example, when AC 150 V or less is supplied as a commercial AC power source, the switch circuit 4 is turned on and is turned on, and when AC 150 V or more is supplied, it is turned off. Controlled to be off. Note that the other external parts, the resistor R ₁₁ and the capacitors C _{11 to} C ₁₃ are the triac TA in the switch circuit 4.
A protection circuit is provided to prevent a surge current when turning on / off the device and noise mixed in the gate signal of the triac TA.

Therefore, in the circuit configuration shown in FIG. 12, an operation will be described when an _AC 100V system ( _AC 150V or less) is supplied as the AC input voltage V _AC supplied to the commercial AC power supply AC. In this case, as described above, the triac TA of the switch circuit 4 is turned on and becomes conductive. And the AC input voltage V _AC
The rectified current during the positive period is commercial AC power supply AC → main switch MS → common mode choke coil CMC winding Na → rectifier diode Db → smoothing capacitor Ci _A → triac TA → inrush current limiting resistance Ri → common mode choke coil The current flows from the winding Nb of the CMC to the commercial AC power supply AC. On the other hand, the AC input voltage V _AC is negative period, the rectification current winding Nb → inrush current limiting resistor R of the commercial AC power supply AC → common mode choke coil CMC
i → triac TA → smoothing capacitor Ci _B → rectifier diode Da → common mode choke coil CMC winding Na → main switch MS → commercial AC power source AC.

That is, at this time, the smoothing capacitor Ci
By charging _A and Ci _{B in} the positive period and the negative period, respectively, a voltage doubler rectifying and smoothing operation is obtained in which the rectifying and smoothing voltage Ei is a 200V system voltage that is almost double the level of the _AC input voltage VAC.

On the other hand, when the AC input voltage V _AC of AC200V-based (or 150V) is supplied to the triac TA of the switch circuit 4 is turned off. In this case, the rectified current during the positive period of the _AC input voltage VAC is as follows: commercial AC power supply AC → main switch MS → common mode choke coil CMC winding Na → rectifier diode Db → smoothing capacitor Ci _A −smoothing capacitor C
i _B → rectifying diode Dc → inrush current limiting resistance Ri → winding Nb of common mode choke coil CMC → commercial AC power supply AC. In the period when the AC input voltage is negative, the current is commercial AC power supply AC → winding Nb of the common mode choke coil CMC → inrush current limiting resistance Ri → rectifying diode Dd → smoothing capacitor Ci _A → smoothing capacitor Ci _B → rectifying diode. Da → common mode choke coil CMC winding Na → main switch M
It flows in the path of S → commercial AC power supply AC. In other words, the bridge rectifier circuit full-wave rectified rectifier output to charge the series connected smoothing capacitors Ci _A - Ci _B to obtain the rectified smoothed voltage full-wave rectification operation is performed input voltage by D ₁ in this case A corresponding DC voltage Ei of 200 V system is obtained.

By switching on / off of the switch circuit 4 in this way, when the AC power input is 100V system, the voltage doubler rectifying and smoothing operation is performed, while the AC power input is 20V.
In the case of 0V system, a normal full-wave rectifying and smoothing operation is performed to configure a power supply circuit corresponding to a wide range AC input voltage.

In the switching power supply circuit,
Generally, when the commercial power supply is rectified, the current flowing through the smoothing circuit has a distorted waveform, which causes a problem that the power factor indicating the utilization efficiency of the power supply is impaired. Further, there is a need for a measure for suppressing harmonics generated due to the distorted current waveform.

Therefore, various kinds of switching power supply circuits for which the power factor has been improved have been proposed by the present applicant.
FIG. 13 is a circuit diagram showing an example of a switching power supply circuit configured based on the invention previously applied by the present applicant. The power supply circuit shown in this figure is similar to the power supply circuit in FIG.
In addition to the wide-range compatible configuration in which the voltage doubler rectification smoothing operation and full-wave rectification smoothing operation are switched between the system, a separately-excited current resonance type switching converter and a power factor correction circuit for improving the power factor are provided. It is provided with a configuration. The same parts as those in FIG. 12 are designated by the same reference numerals and the description thereof will be omitted.

In the switching power supply circuit shown in FIG. 13, a power factor correction rectifier circuit 20 is provided for the commercial AC power supply AC as shown in the figure. This power factor correction rectifier circuit 2
0 is an AC input voltage AC100V system and AC200V system, and is provided with a rectifier circuit capable of switching between a double voltage rectifying and smoothing operation and a full wave rectifying and smoothing operation, and a circuit configuration for improving a power factor as described later. Is formed. In the power factor correction rectifier circuit 20, a filter choke coil L _N is inserted in series in the positive electrode line of the commercial AC power supply AC, and in the normal mode together with the filter capacitor C _N connected in parallel to the commercial AC power supply AC. A low pass filter can be formed to prevent harmonic currents from flowing to the commercial AC power supply AC. In this case, four rectifying diodes D _F1 forming the bridge rectifying circuit D _1A ,
As described below, D _F2 , D _F3 , and D _F4 are of the high-speed recovery type in response to the high-frequency current of the switching period flowing in the rectifying current path.

A primary winding N ₁ of an insulating transformer PIT of a switching converter, which will be described later, is connected to the connection point of the rectifying diodes D _F1 and D _F2 of the bridge rectifying circuit D _1A via a series resonance capacitor C _1. Thus, the switching output of the switching converter obtained at the primary winding N ₁ is fed back to the rectified current path via the capacitive coupling of the series resonance capacitor C ₁ . Also,
Resonance capacitors C ₂ and C ₂ are connected in parallel to the rectifier diodes D _F1 and D _F2 , respectively.

In the case of this power factor correction rectifier circuit 20, FIG.
The switch circuit 4 corresponding to the triac TA provided in the circuit of FIG.
It is supposed to be ₁ . The switch section S ₁ is provided so as to pass through a connection point between the smoothing capacitors Ci _A and Ci _{B and} a connection point between the negative electrode input terminals (rectifier diodes D _F3 and D _F4 ) of the bridge rectifier circuit D _1A .

The electromagnetic relay RL-1 is driven by a relay drive circuit 30 provided in the power factor correction rectifier circuit 20. The relay drive circuit 30 is provided with a half-wave rectification circuit including a rectification diode D _{6 for} rectifying the commercial AC power supply AC in a half-wave and a smoothing capacitor C _5, and between the output of the half-wave rectification circuit and the primary side ground. Resistance R
₁ and R ₂ are connected in series. A zener diode ZD is inserted between the voltage dividing points of the resistors R ₁ and R ₂ and the base of the transistor Q ₃ . In this case, when the AC input voltage V _AC supplied to the commercial AC power source AC is 150 V or more, the above-mentioned components are arranged so that the Zener diode ZD becomes conductive by the voltage value divided by the resistors R ₁ and R _2. It is considered to have been selected. That is, a voltage detection circuit for detecting whether or not the AC input voltage level is equal to or higher than AC 150 V is formed by each of the above components. Transistor Q ₃ is to drive the electromagnetic relay RL-1. This transistor Q ₃
A resistor R ₃ and a capacitor C ₆ are connected between the base and the primary side ground. Also, the transistor Q
The collector of ₃ is grounded to the primary side ground. In addition, the emitter is via the relay drive unit R _D1 of the electromagnetic relay RL-1,
It is connected to the line of the low voltage DC voltage E _P obtained by the tertiary winding N ₃ , the rectifying diode D ₄ and the smoothing capacitor C ₄ of the insulation transformer PIT which will be described later. A protective diode D ₅ for flowing a reverse current is connected in parallel to the relay drive unit R _D1 .

The switching of the rectifying operation of the power factor correction rectifier circuit 20 thus constructed is performed as follows.
For example, when an AC input voltage V _AC of AC 150 V or less is supplied as the _AC 100 V system, the relay drive circuit 3
Since the Zener diode ZD of 0 does not conduct, the base current of the transistor Q ₃ is made to flow through the resistor R ₃ and turned on. As a result, the emitter current is conducted to the relay drive unit R _D1 of the electromagnetic relay RL-1. The switch S ₁ is turned on by the exciting action of the relay driver R _D1 . As a result, in the same manner as described with reference to FIG. 12, the AC input voltage V _AC
Commercial AC power source AC to charge the smoothing capacitor Ci _A, a commercial AC power source AC to the smoothing capacitor Ci ac input voltage V _AC is the negative feedback rectified by the rectifier diode D _F1 There is a positive feedback which is rectified by the rectifier diode D _{F1 The} voltage doubler rectifying / smoothing operation is performed to charge _B , and the 200V rectifying / smoothing voltage Ei, which is almost twice that of the AC100V system, is obtained.

On the other hand, the AC200V system is AC
When the AC input voltage V _AC of 150 V or more is supplied, the Zener diode ZD of the relay drive circuit 30 becomes conductive, so that the base potential of the transistor Q ₃ is pulled up to a predetermined level or higher so that the base current does not flow. Then, the transistor Q ₃ is turned off. Therefore, the emitter current of the transistor Q ₃ stops flowing through the relay driver R _D1 , and the switch S ₁ is turned off. In this case, the full-wave rectification in which the commercial AC power source AC is full-wave rectified by the bridge rectifier circuit D _1A and the smoothing capacitors Ci _A -Ci _B are connected in series as in the case described with reference to FIG. Smoothing operation and AC input voltage V _AC
Thus, the rectified and smoothed voltage Ei of the AC200V system corresponding to is obtained. The power factor improving operation of the power factor improving rectifier circuit 20 will be described later.

The switching converter shown in FIG.
It is a current resonance type converter by excitation type. In this case
Is, for example, a switching element Q of two stones₁₁, Q₁₂Equipped with
Switching element Q₁₁Rectified smoothing voltage E
connected to the line of i, and the switching element Q₁₁Source of
Switching element Q₁₂Connect the drain of the switch
Element Q₁₂Connect the source to the primary side ground, Iwayu
Connected by a half-bridge coupling. These
Switching element Q₁₁, Q₁₂Is the oscillation drive circuit 2.
Switch so that the on / off operation is repeated alternately.
Driven by switching, and intermittently switches the rectified and smoothed voltage Ei.
Teaching output. Note that the switching element Q₁₁, Q₁₂
For example, a MOS-FET is used. In addition, each
Switching element Q ₁₁, Q₁₂Between the drain and source of
Connected in the direction shown in the figure_D , D_D Switchon
Element Q₁₁, Q₁₂Shape the path of the current returned when the
It is used as a clamp diode.

The source-drain connection point of the switching elements Q ₁₁ and Q ₁₂ is a switching output point, and one end of the primary winding N ₁ of the insulating transformer PIT is connected to this switching output point, and this primary winding is connected. It is adapted to provide a switching output for line N ₁ . The primary winding N ₁ of the insulation transformer PIT has a series resonance capacitor C ₁
Isolation transformer PIT which is connected in series with and which includes the capacitance of the series resonance capacitor C ₁ and the primary winding N _1.
The inductance component of forms a series resonance circuit for making the switching power supply circuit a current resonance type.
In the present specification, this series resonance circuit is particularly referred to as "primary side series resonance circuit".

The isolation transformer PIT transmits the alternating voltage obtained by the switching output supplied to the primary winding N ₁ to the secondary side. In the case of this power supply circuit, isolation transformer P
On the secondary side of IT, a rectifier diode D _{3A is connected} to the secondary winding N ₂ whose center tap is grounded to the secondary side ground.
A double wave rectifier circuit is provided by D _3B and the smoothing capacitor C ₃ . As a result, the alternating voltage excited from the primary winding N ₁ to the secondary winding N ₂ is converted into a DC voltage by the double-wave rectifying circuit to obtain a DC output voltage E ₀ .

Further, in this power supply circuit, the control circuit 1 controls the oscillation drive circuit 2 based on the fluctuation of the DC output voltage E _O , and supplies it from the oscillation drive circuit 2 to the gates of the switching elements Q ₁₁ and Q _12. The constant voltage control of the DC output voltage E _O is performed by changing the switching drive signal (for example, varying the pulse width of the drive signal).

The starting circuit 3 is provided to detect the voltage or current obtained in the rectifying and smoothing line and to start the oscillation drive circuit 2 immediately after the power is turned on when the main switch MS is turned on. The starting circuit 3 is supplied with the low-voltage DC voltage E _P supplied by the tertiary winding N ₃ provided in the insulating transformer PIT, the rectifying diode D ₄ , and the smoothing capacitor C ₄ . Since the field effect type switching element used in this embodiment is driven by voltage and self-excited oscillation becomes difficult, it is preferable to provide the oscillation drive circuit 2 and the starting circuit 3 as shown in this figure.

Next, the power factor improving operation of the power factor improving rectifier circuit 20 will be described. As described above, the primary side series resonance circuit is connected to the positive input terminal (the connection point of the rectification diodes D _F1 and D _F2 ) of the bridge rectification circuit D _1A of the power factor correction rectification circuit 20. Is an isolation transformer P
The switching output obtained from the primary winding N ₁ of IT is fed back to the rectified current path via the capacitive coupling of the series resonance capacitor C ₁ . The switching output fed back in this way is an alternating voltage (switching voltage) of a switching cycle with respect to the rectified output voltage passing through the inductance of the filter choke coil L _N.
, And the rectifying diodes D _F1 and D _F2 operate so as to switch the rectified current in a switching cycle by the superimposed switching voltage. Since the rectifier diodes D _F1 and D _F2 are in the path of the rectified current in both the case of voltage doubler rectification and the case of full wave rectification, the above operation is performed in both the case of voltage doubler rectification and the case of full wave rectification. Will also be performed.

By this operation, for example, in the voltage doubler rectifying operation, the rectified output voltage is charged in the smoothing capacitors Ci _A and Ci _B with the switching voltage superposed, and the superposition of the switching voltage causes The voltage across each of the smoothing capacitors Ci _A and Ci _B is lowered in the switching cycle. For this reason, the smoothing capacitors Ci _A and C are also used during the period in which the rectified output voltage level is lower than the voltage across each of the smoothing capacitors Ci _A and Ci _B.
The charging current to i _B is caused to flow. Further, at the time of the full-wave rectification operation, the rectified output voltage is to perform smoothing capacitor Ci _A - Ci _B charging connected in series with the rectified output voltage switching voltage is superimposed, the superimposed portion of the switching voltage, the series Voltage across the connected smoothing capacitors Ci _A -Ci _B (rectified smoothed voltage Ei)
Will be lowered in the switching cycle. For this reason, the charging current is allowed to flow even in a period in which the rectified output voltage level is lower than the voltage across the smoothing capacitors Ci _A -Ci _B connected in series. As a result, in either case of the double voltage rectification operation or the full-wave rectification operation, the average waveform of the _AC input current I _AC is made to approach the waveform of the _AC input voltage VAC, and the AC input current I _AC The conduction angle will be enlarged. In this way, in the power supply circuit shown in this figure, the power factor is improved in both the double voltage rectification operation and the full-wave rectification operation.

Further, in this power supply circuit, the resonance capacitors C ₂ and C ₂ are provided in parallel with the rectifying diodes D _F1 and D _F2 of the bridge rectifying circuit D _1A as described above. In this case, the resonance capacitors C ₂ and C ₂ form a parallel resonance circuit together with the filter choke coil L _N and the like. The resonance impedance of the parallel resonance circuit is changed in response to load fluctuations, and the switching output fed back to the rectified current path is suppressed when the load of the switching power supply circuit becomes light. This makes it possible to suppress an increase in the terminal voltage (rectification smoothing voltage) of the smoothing capacitor when the load is light.

The power supply circuit also includes an electromagnetic relay RL-
Two are provided. The relay drive unit R _D2 of the electromagnetic relay RL-2 includes a line of the low-voltage DC voltage E _P obtained by the tertiary winding N ₃ , the rectifying diode D ₄ , and the smoothing capacitor C ₄ provided on the insulation transformer PIT and the primary side. The switch section S ₁₂ is provided so as to be inserted between the grounds, and is provided in parallel with the inrush current limiting resistor Ri inserted in the commercial AC power supply AC line. In addition, the relay drive unit R _D2
A protection diode D ₅ is connected in parallel with the above. For example, immediately after the main switch MS is turned on and the commercial AC power supply AC is turned on to this power supply circuit, no voltage has yet been generated in the line of the low voltage DC power supply E _P ,
Switch part S ₁₂ of the electromagnetic relay RL-2 is in the off by the rush current limiting resistor Ri, the AC voltage AC
The inrush current that flows into the smoothing capacitor from is limited.
Then, after a lapse of about 200 ms to 300 ms, a voltage of a required level rises in the line of the low voltage DC power supply E _P , which causes the relay drive unit R _D2 of the electromagnetic relay RL-2 to conduct and turn on the switch unit S ₁₂ . Switch to the state,
A path for passing the current limiting resistor Ri is formed. At this time, the inrush current has already been sufficiently attenuated, and thereafter, the circuit becomes an equivalent circuit in which the inrush current limiting resistor Ri is omitted from the line of the commercial AC power supply AC.

If the inrush current limiting resistor Ri is inserted even after the power is turned on, for example, AC200 system P
This is not a problem when an AC input voltage is input, but in the case of an AC100V system, power loss becomes significant. Therefore, with the configuration described above, the inrush current immediately after the power is turned on can be suppressed, and thereafter, the inrush current limiting resistor Ri can be short-circuited to reduce the power loss accordingly. A large cement resistor or the like is used as the inrush current limiting resistor Ri, and its resistance value is selected so that the inrush current is suppressed within the maximum current capacity standard of the rectifier diode.

[0029]

According to the size and cost of the equipment, the switching power supply circuit is reduced in size and weight by reducing the number of parts as much as possible and using small and inexpensive parts. It is preferable to reduce the cost and cost. In addition, it is preferable to improve the electric characteristics such as power conversion efficiency.
In particular, in the power supply circuit shown in FIGS.
Since the AC input current of a level higher than that of the AC200V system in the 0V system flows through the rectifying diode and the triac TA, the power loss in the AC100V system is large.

[0030]

In order to solve the above-mentioned problems, the present invention is directed to a detection circuit for detecting the level of an AC input voltage supplied to a commercial power source and an AC input voltage detected by this detection circuit. Based on the level, the rectifier circuit for rectifying the commercial power source can be switched between a full-wave rectifier circuit for full-wave rectifying the commercial power source by the bridge rectifier circuit and a double-voltage rectifier circuit for double-voltage rectifying the commercial power source. In addition, when the voltage doubler rectifier circuit is used, the power source circuit is provided with a rectifier circuit switching unit configured to form a circuit for rectifying the commercial power source by connecting two sets of rectifier elements in parallel. did.

A switching power supply circuit is provided with a current resonance type switching converter in addition to the above configuration, and with a power factor correction circuit for feeding back the switching output of the switching converter circuit to improve the power factor. To be configured.

Further, in the above configuration of the power supply circuit and the switching power supply circuit, when switching to the voltage doubler rectifier circuit, two inrush current limiting resistors for suppressing the inrush current with a predetermined resistance value are commercialized in parallel. A circuit that can be formed so that it is connected to the power supply line and, when switched to a full-wave rectifier circuit, can be formed so that the inrush current limiting resistor is not connected to the commercial power supply line. When the switching unit is provided or when the voltage is switched to the voltage doubler rectifier circuit, the first inrush current limiting resistor and the second inrush current limiting resistor that suppress the inrush current with a predetermined resistance value are connected in series to the commercial power source. A circuit is formed so that it is connected to the line, and when the circuit is switched to a full-wave rectification circuit, the first inrush current limiting resistor is inserted into the commercial power supply line. Formed it was able to provide a circuit switching unit which can be.

Further, according to the above configuration, for example, in the voltage doubler rectifying operation when the AC input voltage is AC100V system, it is branched into two rectifying elements connected in parallel during the positive and negative periods of the commercial power source. An alternating input current will flow. Further, according to the present invention, a simple circuit provided with an electromagnetic relay is used to switch between the voltage doubler rectification operation and the full-wave rectification operation, and the rush current is applied during the AC100 system in which the AC input current is higher than that during the AC200 system. Omit the current limiting resistor from the commercial power line, or AC
It is possible to switch to a circuit configuration in which the resistance value of the inrush current limiting resistor is lower than that of the 200 system.

[0034]

FIG. 1 is a circuit diagram showing an embodiment of a switching power supply circuit according to the present invention. The same parts as those shown in FIGS. 12 and 13 shown as the conventional example are designated by the same reference numerals and the description thereof will be omitted. In this power supply circuit, two inrush current limiting resistors Ri _A and Ri _B are provided. For example, the inrush current limiting resistors Ri _A and Ri
The resistance value of _B is set to Ri _A = Ri _B = 2Ri with respect to the inrush current limiting resistor Ri shown in FIG. 12, and each resistance value of the inrush current limiting resistors Ri _A and Ri _B is 2 inrush current limiting resistor Ri. It is selected to be equal to double the resistance value.

Further, the power supply circuit shown in this figure is provided with an electromagnetic relay RL-11 and a relay drive circuit 30 for driving the electromagnetic relay RL-11. In this case, the so-called two-circuit, two-contact type is used as the electromagnetic relay RL-11. In other words, the relay drive unit R _D11 is provided with two switch units S ₁₁ and S ₁₂ to form two circuits, and both of the switch units S ₁₁ and S ₁₂ are connected to the terminal T ₁ by the exciting action of the relay drive unit R _D11. There are two contacts that are selectively switched in conjunction with the terminal T ₂ or the terminal T ₃ .
Further, relay drive circuit 30 has the same configuration as relay drive circuit 30 shown in FIG. 13, and in this case as well, the level of _AC input voltage V _AC is 150 V _AC or less (100 V AC).
In the case of the system), the transistor Q ₃ is turned on, the current is conducted to the relay drive unit R _D11, and AC 150 V or more (AC
In the 200 V system), the transistor Q ₃ is turned off so that no current flows in the relay drive unit R _D11 .

In the relay drive circuit 30 in this case, the drive power source for driving the relay drive unit R _D11 is the low-voltage DC voltage E _P of, for example, 12 V drawn from the switching converter SC side. The detection by adjusting the voltage dividing resistors R _1, resistor R ₂ of R ₂ is is used such as a variable resistor or a semi-fixed resistor, the resistance value of the resistor R ₂ for commercial power level of detection accuracy So that it can be raised.

In the electromagnetic relay RL-11, the relay drive unit R
When a current is flowing through _D11 switch section S _11, S ₁₂
Both switch to the terminal T ₃ side, and switch to the terminal T ₂ side when current is not conducted. Therefore,
In the present embodiment, when the AC input voltage V _AC is 150 V _AC or less, both terminals T _{1 and} T ₃ of the switch units S ₁₁ and S ₁₂ are connected.
When the AC voltage is 150 V or more, the terminal T ₁ is controlled to be connected to the terminal T ₂ .

In the power supply circuit shown in FIG. 1, the switch section S ₁₁ has its terminal T ₁ connected to the negative line of the commercial AC power supply AC, and its terminal T ₂ via the inrush current limiting resistor Ri _{B. 1} is connected to the negative input terminal (connection point of rectifying diodes Dc and Dd), and terminal T ₃ is connected to the connection point of smoothing capacitors Ci _A and Ci _B. The terminal T _{1 of} the switch section S ₁₂ is connected to the negative input terminal of the bridge rectifier circuit D ₁ , and the terminal T ₂ is connected to the negative line of the commercial AC power supply AC via the inrush current limiting resistor Ri _A. T ₃ is connected to the positive line of the commercial AC power supply AC.

As the operation of the switching power supply circuit configured as above, first, AC 150 V or more (AC2
When the AC input voltage V _AC of (00V system) is supplied, the operation is as follows. In this case, since the relay drive circuit R _D11 of the electromagnetic relay RL-11 does not conduct current as described above, both the switch units S ₁₁ and S ₁₂ are switched to the terminal T ₂ side. In this state, commercial AC power supply AC
Is full-wave rectified by the bridge rectifier circuit D ₁ to form a circuit for performing the full-wave rectifying and smoothing operation for charging the smoothing capacitors Ci _A -Ci _B connected in series. Further, between the negative line of the commercial AC power supply AC and the negative input terminal of the bridge rectifier circuit D ₁ , inrush current limiting resistors Ri _A and Ri _{B are provided.}
Will be connected in parallel. As described above, the inrush current limiting resistors Ri _A and Ri _B have a resistance value that is twice as large as that of the inrush current limiting resistor Ri inserted in the power supply circuit of FIG. The circuit is equivalent to the case where the resistance value of the inrush current limiting resistor Ri shown in FIG. 13 is inserted in series with respect to the power supply AC, and the inrush current when the power is turned on can be suppressed. In this case, the inrush current limiting resistors Ri _A , Ri _A and
The parallel connection of _B will be inserted in the line of the commercial AC power supply AC, but in the case of AC200V system, AC10
Since the level of the _AC input current I _AC is smaller than that in the 0 V system, the power loss due to the inrush current limiting resistors Ri _A and Ri _B is not particularly a problem. In addition, specifically, for the inrush current limiting resistors Ri _A and Ri _B , for example, both of 6.8Ω are used. Therefore, when connected in parallel, the resistance of 6.8Ω / 2 = 3.4Ω. Is equivalent to being inserted in the AC line.

In addition, AC150V or less (AC100V
When the AC input voltage V _AC of the system) is supplied, it is as follows. First, since the AC input voltage V _AC is 150 V _AC or less in the normal operation after the power supply is started in this case, the emitter current of the transistor Q ₃ is applied to the relay drive circuit R _D11 of the electromagnetic relay RL-11 as described above. Is conducted, and both the switch sections S ₁₁ and S ₁₂ are switched to the terminal T ₃ side. As a result, the smoothing capacitor Ci
The connection point of _A and Ci _B is connected to the negative line of the commercial AC power supply AC. Further, in this case, the terminal T ₁ and the terminal T ₃ of the switch unit S ₁₂ are connected to each other, thereby forming a path in which the positive input terminal and the negative input terminal of the bridge rectifier circuit D ₁ are short-circuited. In addition, inrush current limiting resistors Ri _A and Ri _B
Is equivalent to being deleted from a circuit perspective because both ends are open.

According to the circuit configuration thus formed, for example, the commercial AC power supply AC is rectified by branching to the rectifying diodes Db and Dd during a period when the _AC input voltage V _AC is positive,
The operation of charging the smoothing capacitor Ci _A is obtained, and
During a period in which the _AC input voltage V _AC is negative, the commercial AC power supply AC is branched and rectified by the rectifying diodes Da and Dc, and the smoothing capacitor Ci _B is charged. That is, in this case, the circuit configuration for the voltage doubler rectification operation that generates the rectified and smoothed voltage Ei corresponding to the voltage approximately double the AC input voltage level is obtained, but in the present embodiment, The rectified current at this time is configured to flow through the parallel connection of the rectifier diodes Db-Dd and the parallel connection of the rectifier diodes Da and Dc. The operational effects and the like of this configuration will be described later.

For example, when the AC input voltage is 100V _AC , the level of the _AC input current I _AC is higher than that in the AC 200V system as described above. However, in the circuit configuration formed as described above, the inrush current limiting resistor Ri _{A is used.} , Ri _B are omitted from the line of the commercial AC power supply AC,
The problem of increase in power loss due to the insertion of the inrush current limiting resistor in the C100 system will be solved.

By the way, the AC input voltage V _AC is AC100.
In the case of the V system, for example, immediately after the power is turned on when the main switch MS is switched from the off state to the on state, it takes about 12 seconds until the switching converter SC starts and the low voltage DC voltage E _{P of} 12 V rises. 200 ms ~
It will take 300 ms. Thus, in about 200ms~300ms immediately after power-on, even if even if the AC input voltage is a AC100V system, since the relay driver R _D11 of the electromagnetic relay RL-11 is in a non-conducting state, the switch section S ₁₁ , S ₁₂ are switched to the terminal T ₂ side. Therefore, at this time, the circuit form is the same as that of the AC 200 V system, and the inrush current is suppressed by the inrush current limiting resistors Ri _A and Ri _B inserted in parallel to the line of the commercial AC power supply AC. Then, after about 200 ms to 300 ms has elapsed, the inrush current is attenuated at this time, the low-voltage DC voltage E _P rises, the relay drive unit R _D11 becomes conductive, and both the switch units S ₁₁ and S ₁₂ are connected to the terminal T. Switch to ₃ . Then, as described above, the circuit is switched to a corresponding circuit form when the AC100V system is used.

2 and 3 show operation waveforms of the _AC input current I _{AC with} respect to the AC input voltage V _AC in the power supply circuit of this embodiment. For example, when the AC input voltage V _AC = 100V is supplied as the _AC 100V system as shown in FIG. 2A, the AC input current I _AC has a waveform as shown in FIG. _It flows only when the _AC level is higher than the voltage across the smoothing capacitors Ci _A and Ci _B. In addition, as shown in FIG.
When the AC input voltage V _AC = 230 V is supplied as the 00 V system, the AC input current I _AC becomes as shown in FIG. 3B, and the level of the _AC input voltage V _AC is connected in series to the smoothing capacitor. The current flows during a period higher than the voltage across Ci _A -Ci _B. 2 (b) and 3
As can be seen by comparing (b), the AC input current I _AC has different levels corresponding to the _AC input voltage V _AC = 100 V and 230 V, and as described above, the AC input voltage V _AC = The AC input current I _AC level is higher at 100 V.

Further, FIG. 4 compares the power conversion efficiency characteristics with respect to an AC input voltage at a load power of 230 W with the power supply circuit of this embodiment shown in FIG. 1 and the switching power supply circuit shown as a conventional example in FIG. Is shown. As can be seen from this figure, the power supply circuit of FIG. 1 and the power supply circuit of FIG. 12 have the same power conversion efficiency characteristics during the full-wave rectification operation in the so-called AC200V system in which the _AC input voltage V _AC is 150 V or more. AC input voltage V _AC is 150V or less (A
During the voltage doubler rectification operation (in the C100V system), the power conversion efficiency characteristic of the power supply circuit of FIG. 12 becomes remarkable as the _AC input voltage V _AC rises.

For example, in the case of the power supply circuits of FIGS. 1 and 12, during the voltage doubler rectifying operation when the _AC input voltage V _AC is 100 V system (AC 100 system), the AC input current I _AC is the rectifying diode D ₁ , D ₂ and smoothing capacitors Ci _A , C
flows through the i _B and the triac TA. For example, the AC input voltage V _AC = 100 V Sometimes, the AC input current I having about twice the peak at the time of the alternating input voltage V _AC = 220V as shown in FIG. 2 _AC flows in. Therefore, in the power supply circuit of FIG. 12, the forward voltage drop voltage of the rectifier diodes D ₁ and D ₂ rises, and the forward voltage drop voltage of the triac TA (for example, forward voltage drop voltage V _F = 1.5).
In addition to V), the power loss due to the inrush current limiting resistor Ri is superimposed, so that the power loss increases and the power conversion efficiency characteristic deteriorates especially in the AC100 system.

Because of this, components such as the stacked bridge rectifier circuit D ₁ and the IC of the switch circuit 4 generate heat, so that it is necessary to provide a heat sink for these components. When the load power is about 230 W, for example, the inrush current limiting resistor Ri has 1Ω / 1.
Connect 3 sets of 0W cement resistors in series, or 1
Since a large cement resistance of 3.3 Ω / 30 W was inserted to deal with it, it was difficult to directly mount it on a printed circuit board together with other components.

On the other hand, in the power supply circuit shown in FIG.
Electromagnetic relay RL-2 for passing through switch circuit 4 of IC including triac TA and inrush current limiting resistor Ri
Etc. are omitted, and an electromagnetic relay RL-11 is provided instead to switch between the full-wave rectifier circuit and the voltage doubler rectifier circuit.
i _A and Ri _B are omitted. Further, during the voltage doubler rectification, as described above, the rectification current is the parallel connection of the rectification diodes Db-Dd and the rectification diodes Da-Dc.
The forward voltage drop voltage of the rectifier diode is suppressed from increasing by shunting the current through the parallel connection of the. As a result, the power loss in the AC100V system is reduced by that much, and as shown in FIG. 4, the power efficiency is improved to the extent that the power efficiency equivalent to that in the AC200V system is obtained. For example, specifically, an AC input voltage V _AC = with a load power of 230 W
At 100 V, the power conversion efficiency of the power supply circuit of FIG. 12 was 85%, while the power conversion efficiency of the power supply circuit of FIG.
It was improved to 7%, and the input power was reduced by about 8.3 W as compared with the power supply circuit of FIG.

Along with this, the bridge rectifier circuit D ₁ and the switch circuit 4 required in the power supply circuit of FIG.
The heat radiation plate for the IC and other parts is unnecessary, and the cost can be reduced accordingly. Further, in the present embodiment, for example, inrush current limiting resistors Ri _A and Ri _B are
It is also possible to mount it on a printed circuit board together with other components by using a medium-sized winding resistance of 6.8Ω / 15W, respectively. Is possible.

FIG. 5 is a circuit diagram showing the configuration of a switching power supply circuit according to another embodiment of the present invention.
2 and the same parts as in FIG. In this case, two inrush current limiting resistors Ri _C , R
i _D is provided, for example, the inrush current limiting resistor Ri _C is 1Ω
/ 15W, inrush current limiting resistance Ri _D is 2.2Ω / 20W
Therefore, a medium-sized winding resistor is used for both. The inrush current limiting resistor Ri _D is inserted between the terminal T ₂ of the switch S ₁₁ of the electromagnetic relay RL-11 and the negative line of the bridge rectifier circuit D ₁ . The inrush current limiting resistor Ri _C is inserted between the terminal T ₁ of the switch S ₁₁ of the electromagnetic relay RL-11 and the negative line of the commercial AC power supply AC. In this case, the terminal T ₂ of the switch section S ₁₂ is open.

In the power supply circuit having such a configuration, when the AC input voltage V _AC is a 200 V system of _AC 150 V or more, the relay drive circuit 30 is used as in the power supply circuit of FIG.
By this operation, both switch sections S ₁₁ and S ₁₂ are switched to the terminal T ₂ side. In this state, the bridge rectifier circuit D ₁ performs full-wave rectification on the commercial AC power supply AC to form a circuit for full-wave rectification smoothing operation for charging the smoothing capacitors Ci _A and Ci _B connected in series. It Further, at this time, a series connection of the inrush current limiting resistors Ri _C and Ri _D is inserted in the negative electrode line of the commercial AC power supply AC. In this case, inrush current limiting resistors Ri _C , R
The resistance value obtained by combining i _D is Ri _C + Ri _D = 1Ω + 2.2Ω = 3.3Ω, and is obtained, for example, by connecting the inrush current limiting resistors Ri _A and Ri _B inserted in the circuit of FIG. 1 in parallel. The resistance value is 6.8Ω / 2 = 3.4Ω, which is almost the same.

Further, when the AC input voltage V _AC is a 100 V system of _AC 150 V or less, the relay drive circuit 30
By this, both switch sections S ₁₁ and S ₁₂ are switched to the terminal T ₃ side. Thereby, as in the case of FIG. 1, during the period when the AC input voltage is positive, the smoothing capacitor Ci _A is charged with the rectifying current through the parallel connection of the rectifying diodes Db-Dd,
In the negative period, a voltage doubler rectifier circuit is formed which charges the smoothing capacitor Ci _B with the rectified current through the parallel connection of the rectifier diodes Da-Dc. In the present embodiment, the path of the rectified current (in this case, the smoothing capacitor Ci
Inrush current limiting resistor Ri _C (1Ω / 15W) at the connection point between _A and Ci _B and the negative line of the commercial AC power supply AC
Only the insert will be made.

By thus inserting the inrush current limiting resistor Ri _C even in the AC 100 V system, for example, as described in FIG. 1, 200 ms to 300 ms after the power is turned on, the full-wave rectifier circuit doubles the voltage doubler rectifier circuit. After switching to
Even if the inrush current is not sufficiently attenuated for some reason, the inrush current limiting resistor Ri _C can suppress the inrush current to protect the circuit. The inrush current limiting resistor Ri _C is continuously inserted into the circuit when the AC 100 V system is used, but its resistance value is set to 1 Ω, so that, for example, no problematic power loss occurs. .

Further, in the relay drive circuit 30 of this figure, a circuit portion (30A) surrounded by a broken line, that is, a voltage detection circuit portion composed of resistors R ₁ and R ₂ for voltage division and a Zener diode ZD, and a transistor Q _{3 are shown.} With respect to a circuit portion corresponding to a relay driver portion including a peripheral component and a peripheral component, for example, component parts are mounted as a chip component and a thick film printed resistor on a ceramic substrate or the like, and an AC input voltage V _AC = 150.
If the module circuit unit 30A is formed by trimming the resistor R ₂ so as to switch the rectifying operation within the error range of V ± 1V, the number of components constituting the power supply circuit of the present embodiment can be reduced. At the same time, it is possible to eliminate adjustment. In addition, such a module circuit unit 3
0A can be applied to the power supply circuit of FIG. 1 described above, and can also be applied to the case where the relay drive circuit 30 is provided in the switching power supply circuit of the embodiments described below. It is said that

FIG. 6 is a circuit diagram showing an embodiment in which the present invention is applied to a switching power supply circuit including a current resonance type switching converter. In this case, the switching converter is of the separately excited type in which two switching elements are half-bridge coupled, and is the same part as the power supply circuit of each of the above-described embodiments shown in FIGS. 1 and 5 and the conventional example shown in FIG. Are denoted by the same reference numerals and description thereof will be omitted.

In the power factor correction rectifier circuit 10 shown in the power circuit of this figure, the switch section S of the electromagnetic relay RL-11 is used.
The terminal T ₁ of ₁₁ is connected to the negative line of the commercial AC power supply AC, the terminal T ₂ is opened, and the terminal T ₃ is connected to the connection point of the smoothing capacitors Ci _A and Ci _B. The terminal T ₁ of the switch S ₁₂ is connected to the negative input terminal of the bridge rectifier circuit D _1A , the terminal T ₂ is connected to the negative line of the commercial AC power supply AC, and the terminal T ₃ is the positive electrode of the bridge rectifier circuit D _1A . Connected to the input terminal.

In this embodiment, the inrush current limiting resistance Ri is passed by the electromagnetic relay RL-2 in the same manner as in the power supply circuit of FIG. 13 when 200 ms to 300 ms elapses after the power is turned on. .

According to such a connection form, the AC200
In the V system (AC input voltage V _AC = 150 V or more), a full-wave rectifier circuit is formed by the bridge rectifier circuit D _1A to full-wave rectify the commercial AC power supply AC and smoothing capacitors Ci _A -Ci.
_The circuit configuration is such that the _B series connection is charged. Then, by the same operation as described in the power supply circuit of FIG. 13, the conduction angle of the _AC input current I _AC is expanded and the power factor is improved.

When the AC100 system (AC input voltage V
_{At AC} = 150V or less), as in the power supply circuits of FIGS. 1 and 5, the rectifying diode D _F2 is operated during the period when the AC input voltage is positive.
-Smoothing capacitor C for rectifying current via parallel connection of D _F4
i _A is charged, and in the negative period, the rectifier diodes D _F1 −D _F3
A voltage doubler rectifier circuit for charging the smoothing capacitor Ci _B with the rectified current through the parallel connection of In this case,
By the switching output superimposed on the rectification current path via the primary side series resonance circuit, the rectification diodes D _{F2 to} D _F4 perform the operation of intermittently switching the rectification current in the switching period during the period when the AC input voltage is positive, and during the negative period. In this case, the rectifying diodes D _{F1 to} D _F3 are used to perform the operation of connecting and disconnecting the rectified current. In this case as well, the power factor is improved by the same operation as described with reference to FIG.

Here, FIGS. 7 and 8 show operation waveforms of the AC input current with respect to the AC input voltage of the switching power supply circuit shown in FIG. For example, when the AC input voltage V _AC = 100V is supplied as shown in FIG. 7A and the voltage doubler rectification operation is performed, the AC input current I _AC becomes as shown in FIG. 7B. The τ period that is supposed to flow into the smoothing capacitor side is extended, that is, a waveform with an enlarged conduction angle is obtained, and power factor is improved. In this case, if the half cycle of the _AC input voltage V _AC is 10 ms, the τ period is 5 m.
A power factor of about 0.8 can be obtained by selecting constants for the required parts for improving the power factor so as to obtain s. Further, as shown in FIG. 8A, the AC input voltage V _AC = 220
Even when V is supplied and the full-wave rectification operation is performed, similarly, as shown in FIG. 8B, the conduction angle of the _AC input current I _AC flowing in the τ period is expanded to improve the power factor. To be Also in this case, a power factor of about 0.8 can be obtained by selecting constants for the required parts for improving the power factor so that the τ period becomes 5 ms.

Further, FIG. 9 shows power conversion efficiency characteristics (at load power of 230 W) with respect to an AC input voltage in comparison with the power supply circuit of FIG. 6 and the power supply circuit of FIG. Also in this case, in the AC100V system in which the AC input voltage is AC150V or less, the power conversion efficiency of the power supply circuit of FIG. 6 is higher than that of the power supply circuit of FIG. This is the same as described in the power supply circuit of FIGS. 1 and 5, when the voltage doubler rectifier circuit is formed, the AC input current I _AC is the parallel connection of the rectifier diodes D _{F2 to} D _F4 , or the rectifier diode. D _F1 -D _F3
By branching through the parallel connection to the flow path to the rectification path, the level of the forward voltage drop of the rectification diode is prevented from increasing, and the power loss is reduced. Specifically, when the load power is 230 W and the AC input voltage V _AC = 100 VB, the power conversion efficiency of the power supply circuit of FIG. 13 was 86%, whereas the power conversion efficiency of the power supply circuit of FIG. 6 was improved to 87.2%. Has been done. Further, the AC input power is reduced by about 3.7 W in the power supply circuit of FIG. 6 than in the power supply circuit of FIG. As a result, for example, in the power supply circuit of FIG. 13, it is necessary to provide a radiator plate for at least the rectifying diodes Da and Db through which the AC input current flows during the voltage doubler rectifying operation. In the power supply circuit of No. 2, a heat sink is not required, and the size of the board can be reduced and the cost can be reduced accordingly.

FIG. 10 is a circuit diagram showing the configuration of another embodiment in which the present invention is applied to a switching power supply circuit in which a current resonance type is used in a switching converter as in FIG. 1, shown in FIG.
The same parts as those in FIG. 6 are designated by the same reference numerals and the description thereof will be omitted. By the way, since the switching converter in the power supply circuit of the present embodiment is a self-excited current resonance type converter in which two switching elements are half-bridge coupled, a configuration of the switching converter will be described first.

The switching converter of the power supply circuit of this embodiment is provided with _two switching elements Q ₁ and Q _{2 which} are half-bridge coupled as shown in the figure, and is connected between the connection point on the positive electrode side of the smoothing capacitor Ci and the ground. On the other hand, they are connected via their respective collectors and emitters. Starting resistors R _S and R _S are inserted between the collectors and bases of the switching elements Q ₁ and Q ₂ , respectively, and the resistors R
_B, and adjust the switching elements Q _1, Q ₂ of the base current (drive current) by R _B. Further, the bases of the switching elements Q _1, Q ₂ - each damper diode between the emitter D _D, D _D is inserted. The resonance capacitors C _B and C _B form a series resonance circuit for self-excited oscillation together with the drive windings N _B and N _{B of} the drive transformer PRT described next.

Drive Transformer PRT (Power Regulati
ng Transformer) variably controls the switching frequency of the switching elements Q ₁ and Q ₂ , and in the case of this figure, the drive windings N _B and N _B and the resonance current detection winding N _D are wound. control winding N _C is the orthogonal saturable reactor is wound in a direction orthogonal to the windings. _One end of the drive winding N _B on the switching element Q ₁ side of the drive transformer PRT is connected to the base of the switching element Q ₁ via a resistor R _B resonance capacitor C _B , and the other end is the emitter of the switching element Q ₁ . Connected to. Also, the drive winding N _B on the switching element Q ₂ side
One end of which is grounded and the other end of which is a resistor R _B ,
It is connected via the resonance capacitor C _B to the base of the switching element Q ₂ . The drive winding N _B on the switching element Q ₂ side outputs a voltage having a polarity opposite to that of the drive winding N _B on the switching element Q ₁ side.

Isolation transformer PIT (Power Isolation Tr)
The ansformer transmits the switching outputs of the switching elements Q ₁ and Q _{2 to} the secondary side. One end of the primary winding N ₁ of the insulating transformer PIT is connected to the emitter of the switching element Q ₁ and the switching element Q via the resonance current detection winding N _D.
The switching output can be obtained by connecting to the contact of the ₂ collector. In addition, the primary winding N ₁
Is connected to the connection point of the fast recovery type rectifier diodes D _F1 and D _{F2 in} the power factor correction rectifier circuit 11 (the positive input terminal of the bridge rectifier circuit D _1A ) via the series resonance capacitor C ₁ . The switching output is fed back to the rectified current path by being connected. The inductance component of the insulating transformer PIT including the series resonance capacitor C ₁ and the primary winding N ₁ forms a primary side series resonance circuit for making the switching power supply circuit a current resonance type.

In this case, the control circuit 1 compares the DC voltage output E _O on the secondary side with the reference voltage, for example, and sets the DC current corresponding to the error as the control current to the drive transformer PRT.
Of the control winding N _C of the error amplifier.

In the switching operation of the switching converter having the above configuration, when the commercial AC power source AC is first turned on, the base current is supplied to the bases of the switching elements Q ₁ and Q ₂ via the starting resistors R _S and R _S , for example. However, if the switching element Q ₁ is turned on _first , the switching element Q ₂ is controlled to be turned off. Then, as an output of the switching element Q ₁ , a resonance current flows through the resonance current detection winding N _D → the primary winding N ₁ → the series resonance capacitor C ₁ , but the switching element Q ₂ is turned on in the vicinity where the resonance current becomes zero. , The switching element Q ₁ is controlled to be turned off. And
A resonance current in the opposite direction flows through the switching element Q ₂ . Thereafter, a self-excited switching operation in which the switching elements Q ₁ and Q ₂ are turned on alternately is started. In this way, the switching element Q ₁ and Q ₂ are alternately opened and closed by using the terminal voltage of the smoothing capacitor Ci as the operating power supply, and the primary winding N _{1 of the} insulation transformer PIT is thus repeated.
Is supplied with a drive current close to the resonance current waveform to obtain an alternating output to the secondary winding N ₂ .

When the DC output voltage E _O on the secondary side decreases, the control circuit 1 controls the current flowing through the control winding N _C so that the switching frequency becomes low (close to the resonance frequency). Is controlled so that the drive current flowing through the primary winding N ₁ is increased to achieve a constant voltage. Hereinafter, such a constant voltage control method will be referred to as a switching frequency control method.

Next, the power factor correction rectifier circuit 11 shown in the power supply circuit of this embodiment will be described. First, in the power factor correction rectifier circuit 11, inrush current limiting resistors Ri _A and Ri _B are provided and switch portions S ₁₁ and S of the electromagnetic relay RL-11 are provided in the same manner as described in the power supply circuit of FIG. ₁₂
Since the circuit switching of the voltage doubler rectifier circuit and the full-wave rectifier circuit and the connection mode of the inrush current limiting resistors Ri _A and Ri _B are switched accordingly, the description thereof is omitted here.

Then, the power factor correction circuit 11 according to the present embodiment.
Then, the winding Li of the choke coil CH is inserted in series between the filter choke coil L _N and the positive input terminal of the bridge rectifier circuit D _1A . In this case, the switching output voltage fed back from the primary side series resonance circuit to the rectified current path is superimposed on the rectified input voltage via the inductance of the winding Li with the winding Li of the choke coil CH as a load. To be That is, in the present embodiment, the switching output obtained in the primary side series resonance circuit is configured to be fed back to the rectified current path via the magnetic coupling of the choke coil CH. Then, due to the superposition of the switching output voltage, a switching operation for interrupting the rectification current is performed by the high-speed recovery type rectification diode forming the bridge rectification circuit D _1A in both the voltage doubler rectification operation and the full-wave rectification operation. After that, the conduction angle of the AC input current is expanded by the action similar to the method of feeding back the switching output via the capacitive coupling of the series resonant capacitor C ₁ shown in FIGS. 13 and 6. The power factor will be improved.

The circuit diagram of FIG. 11 shows another embodiment of the present invention. The same parts as those of FIGS. 5, 10 and 13 are designated by the same reference numerals and the description thereof will be omitted. In the case of this power supply circuit, the switching converter is of the self-exciting type current resonance type as in FIG. In addition, the electromagnetic relay RL
Switching of the voltage doubler rectifier circuit / full-wave rectifier circuit according to the AC input voltage level by -11 is similar to that of the power supply circuit of FIG.

In the power factor correction rectifier circuit 12 of the power supply circuit shown in this embodiment, a magnetic coupling transformer MCT is provided. Magnetic coupling transformer MCT shown in this figure
Is constituted by magnetically tightly coupling the primary winding L _P and the secondary winding Li corresponding to the winding of the choke coil CH shown in FIG. In this case, the primary winding L _P of the magnetic coupling transformer MCT is connected in series with the primary side series resonance circuit as shown in the figure, and the secondary winding Li is the filter choke coil L _N.
And a positive input terminal of the bridge rectifier circuit D _1A . Further, in such a case, the resonance capacitor C ₂ is one and is connected in parallel with the secondary winding Li to form a parallel resonance circuit as shown in the figure. This is similar to the case where the power supply circuits of FIGS. 6, 10 and 13 are provided in the rectifier diodes D _F1 and D _F2 , and the phenomenon that the rectified smoothed voltage level increases as the AC input voltage level decreases Will be suppressed.

In the power factor correction rectifier circuit 12 thus provided with the magnetic coupling transformer MCT, the switching output obtained in the primary side series resonance circuit is the magnetic coupling transformer MC.
It is supplied to the primary winding L _P of T. In the magnetic coupling transformer MCT, the switching output voltage supplied to the primary winding L _P is excited in the secondary winding Li via the magnetic coupling, and the switching voltage is superimposed on the rectified current path. Accordingly, thereafter, in the same manner, in both the double voltage rectifying operation and the full-wave rectifying operation, the operation of intermittently rectifying the rectified current by the fast recovery type rectifying diode forming the bridge rectifier circuit D _1A is obtained. By the action based on this operation, the conduction angle of the AC input voltage is expanded and the power factor is improved.

Further, in the power supply circuit shown in this figure, the drive transformer CDT (Converter Drive Transformer) is constructed so that the control winding N _C is not wound, and therefore the switching frequency is fixed. In this case, the insulating converter transformer is a PRT, and the primary winding N ₁ and the secondary winding N ₂ are provided with the control winding N _C so that their winding directions are orthogonal to each other. . In such a configuration, the DC current of a level which is varied according to the variation of the DC output voltage E _O is supplied from the control circuit 1 to the control winding N _C as the control current. In the transformer PRT, the leakage magnetic flux is changed to change the inductance of the primary winding N ₁ . Due to this inductance change, the resonance frequency of the primary side series resonance circuit is variably controlled with respect to the switching frequency, which makes it possible to make the secondary side DC output voltage E _O a constant voltage (series resonance frequency control method). ).

The present invention is not limited to the configuration shown in each of the above-described embodiments, but various changes can be made. For example, a double rectifier circuit / full-wave rectifier circuit switching and It goes without saying that the combination of the connection configuration switching configuration of the inrush current limiting resistor and the circuit configuration for improving the power factor is not limited to the combination pattern shown as the embodiment in each of the above drawings. .
Similarly, the combination pattern of the power factor correction rectifier circuit and the current resonance type switching converter connected to the subsequent stage is also a self-excited oscillation type / externally excited oscillation type, switching frequency control method / series resonance frequency control method, switching element Half-bridge coupling type / Full-bridge coupling type (formed by combining four stone switching elements)
Various combinations of various methods and types are conceivable.

[0076]

As described above, according to the present invention, when the AC input voltage is 100V AC system and the AC input voltage is AC corresponding to a wide range.
In a power supply circuit and a switching power supply circuit configured to switch to a voltage doubler rectifier circuit and a full-wave rectifier circuit at the time of 200 V system, when a voltage doubler rectifier circuit corresponding to the AC100 V system is formed, in the rectifier path. The rectifying diodes are connected in parallel, and the inrush current limiting resistor for suppressing the inrush current is omitted from the AC line, or the circuit is switched so that the resistance is lower than that in the AC200V system. That is, especially A
It has an effect that remarkable power loss is significantly reduced at the time of C100V system and power conversion efficiency can be improved. Further, along with this, it becomes possible to eliminate the need for a heat dissipation plate to be provided in the rectifying diode. Also,
By switching the switching of the rectifying circuit and the switching of the inrush current limiting resistance insertion mode by interlocking with, for example, a circuit configuration including an electromagnetic relay with two circuits and two contacts, an IC including, for example, a triac Circuit and this I
The heat sink provided in the C circuit and the dedicated relay circuit that passes the inrush current limiting resistance can be omitted, and the circuit components can be reduced accordingly, and the power supply circuit can be made smaller and lighter. It will be possible.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a switching power supply circuit as an embodiment of the present invention.

FIG. 2 is a waveform diagram showing an operation of an AC input current in a commercial AC power supply cycle (at AC 100 V) of the switching power supply circuit of the present embodiment.

FIG. 3 is a waveform diagram showing an operation of an AC input current in a commercial AC power supply cycle (at 220V AC) of the switching power supply circuit of the present embodiment.

FIG. 4 is an explanatory diagram showing power conversion efficiency characteristics of the switching power supply circuit of the present embodiment in comparison with a conventional example.

FIG. 5 is a circuit diagram showing a switching power supply circuit according to another embodiment.

FIG. 6 is a circuit diagram showing a switching power supply circuit as still another embodiment.

FIG. 7 is a waveform diagram showing an operation of an AC input current in a commercial AC power supply cycle (at AC 100 V) of the switching power supply circuit of the embodiment shown in FIG.

8 is a waveform diagram showing an operation of an AC input current in a commercial AC power supply cycle (at 220V AC) of the switching power supply circuit of the embodiment shown in FIG.

9 is an explanatory diagram showing power conversion efficiency characteristics of the switching power supply circuit of the embodiment shown in FIG. 6 in comparison with a conventional example.

FIG. 10 is a circuit diagram showing a switching power supply circuit as still another embodiment.

FIG. 11 is a circuit diagram showing a switching power supply circuit as still another embodiment.

FIG. 12 is a circuit diagram showing a switching power supply circuit as a conventional example.

FIG. 13 is a circuit diagram showing a switching power supply circuit as a conventional example.

[Explanation of symbols]

1 Control circuit 2 Oscillation drive circuit 3 Starter circuit 10, 11, 12 Power factor improvement rectifier circuit 30 Relay drive circuit L _N filter choke coil C _N filter capacitor D ₁ , D _1A Bridge rectifier circuit Ci _A , Ci _B smoothing capacitor CH choke Coil MCT Magnetic coupling transformer PIT (PRT) Isolation transformer CDT (PRT) Drive transformer Q ₁ , Q ₂ , Q ₁₁ , Q ₁₂ Switching element C ₁ Series resonance capacitor N ₁ Primary winding C ₂ Resonance capacitor

Claims (20)

[Claims] 1. A detection means for detecting an AC input voltage level supplied to a commercial power supply, and a rectification circuit for rectifying the commercial power supply based on the AC input voltage level detected by the detection means, by a bridge rectification circuit. It is possible to switch between a full-wave rectifier circuit for full-wave rectifying a commercial power source and a double-voltage rectifier circuit for double-voltage rectifying a commercial power source. In the case of a double-voltage rectifier circuit, two sets of rectifying elements are provided. And a rectifier circuit switching means configured to form a circuit for rectifying a commercial power source by parallel connection of the power source circuit. 2. The power supply circuit according to claim 1, wherein the rectifying element in the voltage doubler rectifying circuit is a rectifying element forming the bridge rectifying circuit. 3. A circuit in which two inrush current limiting resistors for suppressing an inrush current with a predetermined resistance value are connected in parallel to a commercial power supply line when switched to the voltage doubler rectifier circuit. And a circuit switching means capable of forming a circuit in which the inrush current limiting resistor is not connected to the commercial power supply line when switched to the full-wave rectifier circuit. The power supply circuit according to claim 1 or claim 2. 4. A first resistor that suppresses an inrush current with a predetermined resistance value when switched to the voltage doubler rectifier circuit.
Forming a circuit in which the inrush current limiting resistor and the second inrush current limiting resistor are connected in series to the commercial power supply line, and switching to the full-wave rectification circuit, the first inrush current limiting resistor 3. The power supply circuit according to claim 1, further comprising a circuit switching means capable of forming a circuit in which only the resistor is inserted into the commercial power supply line. 5. The rectifier circuit switching means and the circuit switching means are provided with an electromagnetic relay and an electromagnetic relay driving means for driving the electromagnetic relay based on a detection output of the detecting means, thereby operating in an interlocking manner. The power supply circuit according to any one of claims 1 to 4, wherein the power supply circuit is configured to: 6. The circuit component forming at least one of the detecting means and / or the electromagnetic relay driving means is at least partially modularized.
Alternatively, the power supply circuit according to claim 4. 7. A bridge rectifying circuit for detecting a means for detecting an AC input voltage level supplied to a commercial power source and a rectifying circuit for rectifying the commercial power source based on the AC input voltage level detected by the detecting means. It is possible to switch between a full-wave rectifier circuit for full-wave rectifying a commercial power source and a double-voltage rectifier circuit for double-voltage rectifying a commercial power source, and in the case of a double-voltage rectifier circuit, two sets of rectifying elements Rectifying circuit switching means configured to form a circuit for rectifying a commercial power supply by parallel connection of, and a primary side series resonance circuit formed by series connection of a primary side winding of an insulating transformer and a series resonance capacitor. The smoothed DC voltage obtained based on the rectified output of the rectifier circuit is input for switching operation, and the DC output voltage is output from the secondary side of the insulation converter transformer. A current resonance type switching converter means and a power factor improving means for feeding back the switching output of the switching converter means to the rectified current path of the rectifying circuit to improve the power factor. A switching power supply circuit characterized by being provided. 8. The switching power supply circuit according to claim 7, wherein the rectifying element in the voltage doubler rectifying circuit is configured using a rectifying element forming the bridge rectifying circuit. 9. A circuit in which two inrush current limiting resistors for suppressing an inrush current with a predetermined resistance value are connected in parallel to a commercial power supply line when switched to the voltage doubler rectifier circuit. And a circuit switching means capable of forming a circuit in which the inrush current limiting resistor is not connected to the commercial power supply line when switched to the full-wave rectifier circuit. 7. The switching power supply circuit according to claim 7 or 8. 10. A commercial power supply line in which a first inrush current limiting resistor and a second inrush current limiting resistor that suppress an inrush current with a predetermined resistance value when switched to the voltage doubler rectifier circuit are connected in series. Is formed so as to be connected to the full-wave rectification circuit, and the circuit is formed so that the first inrush current limiting resistor is inserted into the commercial power supply line when switched to the full-wave rectification circuit. 9. The switching power supply circuit according to claim 7 or 8, further comprising a circuit switching means capable of performing the above. 11. The rectifier circuit switching means and the circuit switching means are provided with an electromagnetic relay and an electromagnetic relay driving means for driving the electromagnetic relay based on a detection output of the detecting means, thereby operating in an interlocking manner. The switching power supply circuit according to claim 9, wherein the switching power supply circuit is configured to: 12. The method according to claim 7, wherein at least a part of circuit components forming the detection means and / or the electromagnetic relay drive means is modularized. The switching power supply circuit described. 13. The power factor improving means includes a filter choke coil inserted in series in a rectifying path of the rectifying circuit, and a filter capacitor provided so as to form a low pass filter together with the filter choke coil, 13. The switching power supply circuit according to claim 7, wherein the primary side series resonance circuit is connected so that a switching output is applied to a rectifying element that forms a rectifying circuit. 14. The power factor improving means is provided so as to form a low-pass filter together with a filter choke coil, a choke coil winding, and the filter choke coil which are inserted in series in a rectification path of the rectification circuit. 13. The primary side series resonance circuit is connected so that a switching output is applied to a rectification element forming the rectification circuit, the filter being provided with a filter capacitor. The switching power supply circuit described in. 15. The switching power supply circuit according to claim 13, wherein a resonance capacitor is provided in parallel with the rectifying element forming the rectifying circuit. 16. The power factor improving means includes a magnetic coupling transformer formed by magnetically coupling a first winding and a second winding, and the filter choke coil and the first winding are connected to each other. A filter capacitor that is inserted in series in a rectification path of a rectification circuit, the second winding is connected in series to the primary side series resonance circuit, and is provided so as to form a low-pass filter together with the filter choke coil. The switching power supply circuit according to any one of claims 7 to 12, wherein the switching power supply circuit comprises: 17. The switching power supply circuit according to claim 16, wherein a resonance capacitor is provided in parallel with the first winding. 18. The switching converter means comprises:
A constant voltage control is performed by varying a switching frequency of a switching element of the switching converter means based on a DC output voltage obtained on the secondary side of the isolation transformer. The switching power supply circuit according to any one of claims 7 to 17. 19. The switching converter means comprises:
8. The constant voltage control is performed by changing the magnetic flux of the insulation transformer based on the DC output voltage obtained on the secondary side of the insulation transformer.
A switching power supply circuit according to claim 17. 20. The switching converter means is a separately-excited current resonance type converter, and constant voltage control is performed by varying a switching drive signal based on a DC output voltage obtained on the secondary side of the insulation transformer. 18. The switching power supply circuit according to claim 7, wherein the switching power supply circuit is configured as described above.