JPH09140143A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce the size, weight and cost of a switching power supply circuit. SOLUTION: A power factor improving circuit 11 is constructed so as to have the switching output of a switching circuit (Q3 and Q4 ) which is connected in parallel to the switching devices of a current resonance type converter which are coupled with each other by half-bridge coupling supplied to the connection point of rectifying diodes DF1 and DF2 . With this constitution, the peak level of a switching current applied to a switching device (Q1 , Q2 , Q3 or Q4 ) is reduced to approximately a half of the peak level of the conventional constitution wherein a switching circuit is not provided. Further, a rectification circuit system with which the rectification operation mode can be switched between the double- voltage rectification operation and the full-wave rectification operation for an AC 100V system and an AC 200V system respectively is provided to keep the equivalent power factor characteristics for both the AC 100V system and the AC 200V system.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current resonance type switching power supply circuit for improving power factor, for example.

[0002]

2. Description of the Related Art In recent years, a switching power supply circuit has been widely used. Generally, in a switching power supply circuit, when a commercial power supply is rectified, a current flowing through a smoothing circuit has a distorted waveform, so that a power factor indicating the utilization efficiency of the power supply is obtained. The problem of being damaged occurs. Further, there is a need for a measure for suppressing harmonics generated due to the distorted current waveform.

Therefore, a switching power supply circuit with improved power factor was previously proposed by the present applicant, but the power factor was improved based on the invention of the applicant previously filed. An example of the switching power supply circuit is shown in the circuit diagram of FIG. This switching power supply circuit is provided with a voltage doubler rectifier circuit as will be described later, so that it is possible to cope with a relatively heavy load condition.

In the switching power supply circuit shown in this figure, a common mode choke coil CMC and an across capacitor C _L are provided as a noise filter for removing common mode noise with respect to the commercial AC power supply AC. The power factor correction rectifier circuit 20 receives the commercial AC power supply AC, performs double voltage rectification as described later, and generates a rectified and smoothed voltage corresponding to almost twice the AC input voltage level, and a switching converter in the subsequent stage. Is supplied as an operating power source, and the conduction angle of the AC input current is expanded to improve the power factor. In this power factor correction rectifier circuit 20, a filter choke coil L _N is inserted in series to the positive line of the commercial AC power supply AC, and the filter capacitor C _N is connected in parallel to both pole lines of the commercial AC power supply AC as shown in the figure. It is inserted to form a normal mode low-pass filter together with the filter choke coil L _N. The rectifier diodes D ₁₁ and D ₁₂ are supposed to form a voltage doubler rectifier circuit,
The cathode of the rectifier diode D ₁₁ is connected to the positive line of the commercial AC power supply AC via the filter choke coil L _N , and the anode is grounded to the primary side ground. The anode of the rectifier diode D ₁₂ is connected to the cathode of the rectifier diode D ₁₁ , and its anode is the smoothing capacitor Ci.
Connected to the positive terminal of _A. The rectifying diodes D ₁₁ and D ₁₂ are of a high-speed recovery type in response to the flow of a rectified current that is intermittent in a switching cycle as described later. Further, the rectifier diodes D ₁₁ and D ₁₂ have
Resonant capacitors C ₂ and C ₂ are respectively connected in parallel, and the function thereof will be described later.

The choke coil CH ₁ and the second series resonance capacitor C _1A are connected in series to the power factor correction rectifier circuit 20, and the inductance L _O and capacitance form a series resonance circuit. . In addition, in the present specification, this series resonance circuit will be described in "Second
"Series resonance circuit" of the primary side series resonance circuit described later. In this case, choke coil CH ₁
Is connected to the switching output point of the switching converter described later, and the end on the side of the second series resonance capacitor C _1A is connected to the connection point of the rectifying diodes D ₁₁ and D ₁₂ . As a result, the switching output can be fed back to the rectification path via the second series resonance circuit to improve the power factor, which will be described later.

As a smoothing circuit of this switching power supply circuit, in order to form a voltage doubler rectifying circuit, two smoothing capacitors Ci _A and Ci _B are connected in series, and between the rectifying and smoothing line and the primary side ground. It is provided so as to be inserted. This smoothing capacitor Ci _A -C connected in series
The connection point of i _B is connected to the negative line of the commercial AC power supply AC.

The power factor correction rectifier circuit 20 is formed by the above-mentioned connection configuration, and its voltage doubler rectification operation is as follows. For example, during a period in which the AC input voltage supplied to the commercial power supply AC is positive, the rectified current is “commercial AC power supply AC →
Common mode choke coil CMC winding Na → filter choke coil L _N → rectifier diode D ₁₂ → smoothing capacitor Ci _A → common mode choke coil CMC winding Nb → commercial AC power supply AC ”, and a rectifier diode D ₁₂ The smoothing capacitor Ci _A is charged by the rectified output of. Therefore, as shown in the figure, a rectified and smoothed voltage of Ei level corresponding to the AC input voltage level is obtained across the smoothing capacitor Ci _A.

On the other hand, while the AC input voltage V _AC is negative, the rectified current is commercial AC power supply AC → winding Nb → smoothing capacitor Ci _B → rectifying diode D ₁₁ → filter choke coil L _N → winding Na → commercial AC. It flows in the path of the power supply AC, and the smoothing capacitor Ci _B is charged with the rectified output of the rectifying diode D ₁₁ , and E is applied to both ends of the smoothing capacitor Ci _B.
A rectified smoothed voltage of the level i is obtained. As a result of performing the charging operation on the smoothing capacitors Ci _A and Ci _{B in} the positive period and the negative period, respectively, the total rectified and smoothed voltage obtained across the smoothing capacitors Ci _A -Ci _B connected in series is Ei + Ei. The voltage doubler rectifying operation is performed to obtain = 2Ei. Therefore, if the input AC input voltage level is 100V system, a rectified and smoothed voltage of 200V system corresponding to almost double the peak level of the AC input voltage can be obtained.

Further, the power factor correction rectifier circuit 20 is provided with a separately excited current resonance type converter in which two switching elements are half-bridge coupled as a switching converter using the rectified and smoothed voltage of the level of 2Ei as an operating power source. ing. In this case, for example, two switching elements Q ₁₁ and Q ₁₂ are provided and the switching element Q ₁₁
Drain was connected to the positive terminal of the smoothing capacitor Ci _A of a source connected to the drain of the switching element Q ₁₂ of the switching element Q _11, the source of the switching element Q ₁₂ is connected to a primary side ground, connected by so-called half bridge coupling Has been done. These switching elements Q ₁₁ , Q
₁₂ is switching-driven by the oscillation drive circuit 2 so that the ON / OFF operation is alternately repeated, and the voltage across the smoothing capacitors Ci _A -Ci _B connected in series (rectified and smoothed voltage) is intermittently switched to a switching output. To do.
The switching elements Q ₁₁ and Q ₁₂ are, for example, MOS transistors.
-FET is used. The drain of the switching elements _{Q 11, Q 12 - D D1} , D D2 connected in the direction shown with respect to inter-source switching elements Q _11, Q
It is used as a clamp diode that forms a current path that is fed back when ₁₂ is turned off.

In this switching converter, the source-drain connection point of the switching elements Q ₁₁ and Q ₁₂ is used as the switching output point. To this switching output point, one end of the primary winding N ₁ of the insulating transformer PIT is connected via the series resonance capacitor C ₁ and the primary winding N ₁ is connected.
_The switching output is supplied to ₁ .
In this case, the other end of the primary winding N ₁ is grounded to the primary side ground. In such a connection form, the primary winding N ₁ of the isolation transformer PIT is connected in series with the series resonant capacitor C ₁ , but the insulation of the primary resonant circuit C _{1 and} the capacitance of the series resonant capacitor C ₁ is included. Trans PIT
The inductance component of forms a primary side series resonance circuit for making the switching power supply circuit a current resonance type.

In the case of this power supply circuit, the second series resonant circuit consisting of the choke coil CH ₁ and the second series resonant capacitor C _1A in the power factor correction rectifier circuit 20 is also connected to the switching output point. To be done.

In this power supply circuit, as described above,
The series resonance circuit composed of the choke coil CH ₁ and the second series resonance capacitor C _1A in the power factor correction rectifier circuit 20 is referred to as a second series resonance circuit, and the primary winding N ₁ and the series resonance capacitor of the insulating transformer PIT are connected. The primary side series resonance circuit composed of C ₁ will be referred to as a first series resonance circuit, and will be distinguished from each other.

The isolation transformer PIT transmits the alternating voltage obtained by the switching output supplied to the primary winding N ₁ to the secondary side. In the case of this power supply circuit, isolation transformer P
Two sets of secondary windings N ₂ and N ₂ are provided on the secondary side of IT, and the center taps are grounded to the secondary side ground. Then, for each secondary winding N _2, the rectifier diode D _OA, wave rectification circuit by D _OB and a smoothing capacitor C _O is connected and is excited in the secondary winding N _2, N ₂ DC output voltage E ₁ , E ₂ from alternating voltage respectively
Is obtained.

In this power supply circuit, the control circuit 1 controls the oscillation drive circuit 2 based on the fluctuation of the DC output voltage E ₁ , and supplies the oscillation drive circuit 2 to the gates of the switching elements Q ₁₁ and Q _12. The constant voltage control of the DC output voltage E ₁ is performed by changing the switching drive signal (for example, varying the pulse width of the drive signal).

The starting circuit 3 is provided to detect the voltage or current obtained in the rectifying and smoothing line immediately after the power is turned on, and start the oscillation drive circuit 2. The starting circuit 3 includes an insulating transformer PIT. the tertiary winding N ₃ provided in the rectifying diode D _3, and a smoothing capacitor C ₃
The low voltage DC voltage supplied by Since the field effect type switching element as shown in this figure is driven by voltage and self-excited oscillation becomes difficult, it is preferable to provide the oscillation drive circuit 2 and the starting circuit 3 as shown in this figure.

Next, the power factor improving operation of this switching power supply circuit will be described. For example, when the switching converter performs the switching operation as described above, the switching output thereof is supplied to the insulation transformer PIT side via the first series resonance circuit, and at the same time branches and branches in the power factor correction rectification circuit 20. It will also be supplied to the second series resonant circuit. Then, in the second series resonance circuit, the switching output corresponding to the series resonance current I _OA flowing in the inductance L _O of the choke coil CH ₁ is rectified by the rectification diode D via the second series resonance capacitor C _1A.
It is applied to the connection point of ₁₁ and D ₁₂ , that is, the switching output is fed back to the path of the rectified current. The switching output fed back in this manner is converted into a rectified output via the inductance of the filter choke coil L _N , for example, as a voltage of a switching cycle (switching voltage).
, And the fast recovery type diode D ₂ operates so that the rectified current is intermittently switched in the switching cycle due to the superposition of the switching voltage. By this operation, in the power factor correction rectifier circuit 20, the smoothing capacitor Ci in the state where the switching output is superimposed on the rectified output voltage is used.
Then, the voltage across the smoothing capacitor Ci is lowered in the switching cycle by the superposition of the switching voltage. For this reason, the charging current is made to flow to the smoothing capacitor Ci also for the feedback that the rectified output voltage level is lower than the voltage across the smoothing capacitor. As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased, thereby improving the power factor.

Further, resonance capacitors C ₂ and C ₂ connected in parallel with the rectifier diodes D ₁₁ and D ₁₂ , respectively.
Form a parallel resonant circuit together with the filter choke coil L _N and the filter capacitor C _N , for example. This parallel resonant circuit is designed so that its resonant impedance changes in response to load changes, and when the load of this switching power supply circuit becomes lighter, it suppresses the switching voltage fed back to the rectification path. There is. As a result, an increase in the rectified and smoothed voltage when the load is light is suppressed.

By the way, as a power supply circuit with improved power factor, the switching output is fed back to the rectified current path by the first series resonance circuit without providing the second series resonance circuit described above, and the power supply circuit of FIG. The present invention has previously proposed an invention configured to improve the power factor by the same action as that of the power supply circuit, but in this case, an appropriate power factor setting is, for example, about 0.8, and it is assumed that the power factor is 0. It is known that if the power factor is set to about 0.95, the ripple component superimposed on the primary winding of the insulation transformer increases, and the AC ripple component on the secondary side increases. On the other hand, like the power supply circuit shown in FIG. 8, the switching output is fed back through the second series resonance circuit, thereby improving the power factor to about 0.9, for example. Even if the feedback amount of the switching output is set so that the ripple component of the commercial power supply cycle superimposed on the first series resonance circuit can be suppressed.

[0019]

However, in the power supply circuit of FIG. 8, since the series resonance currents I _O and I _OA flowing in the first and second series resonance circuits are combined and flow into the switching element Q ₁₁ , The switching current amount is about twice as much as that of the power supply circuit having the configuration in which the two series resonance circuits are not provided. That is, in the case of the power supply circuit of FIG. 8, the resonance current I _{O is} maintained during the period when the switching element Q ₁₁ is on and Q ₁₂ is off.
Flows in a path of “smoothing capacitor Ci → switching element Q ₁₁ (drain → source) → series resonance capacitor C ₁ → primary winding N ₁ ”. Further, the resonance current I _OA is “smoothing capacitor Ci → switching element Q ₁₁ (drain → source) → choke coil CH ₁ → second series resonance capacitor C _1A → fast recovery type diode D ₂ / resonance capacitor C ₂ ”. It is made to flow through the route. On the other hand, the resonance current I _O during the period when the switching element Q ₁₁ is off and Q ₁₂ is on is “primary winding N ₁ → series resonance capacitor C ₁
→ Switching element Q ₁₂ (drain → source) → smoothing capacitor Ci '. Further, the resonance current I _OA is
“Smoothing capacitor Ci → resonance capacitor C ₂ → second series resonance capacitor C _1A → choke coil CH ₁ → switching element Q ₁₂ (drain → source).

Therefore, specifically, the load power P _O = 250 W
In the condition that the AC input voltage is AC100V, the inductance L _O of the choke coil, the second series resonance capacitor C _1A , and the resonance are set so that the series resonance current I _O = I _OA = 10 Ap-p (peak to peak). Assuming that selects the capacitor C _2, the drain of the switching element Q _11, Q ₁₂ - switching current I _C11, I _C12 flowing between the source 1 for each period when the switching element Q _11, Q ₁₂ is turned on
The current waveform that flows depends on the level of 0 Ap (peak),
When the AC input voltage V _AC is about 80 V, both the switching currents I _C11 and I _C12 increase to about ₁₂ Ap.

Generally, since the heat generation amount of the switching element is said to increase in proportion to the square of the current, the heat generation of the switching element in this case is considerable, so that the switching element has a low on-resistance. It is necessary to adopt one, for example, 15A / 400V if the switching element is a MOS-FET like the power supply circuit of FIG.
It is necessary to reduce the power loss by selecting the TO-3P type with low on-resistance, which increases the cost and size. Further, in such a switching element, since the package is a non-insulating type, it is necessary to attach the heat radiating plate through an expensive silicon rubber, which also causes a cost disadvantage and a high thermal resistance. Therefore, the heat dissipation efficiency also decreases.

[0022]

In order to solve the above problems, the present invention doubles the voltage of a commercial power source to generate a rectified smoothed voltage corresponding to approximately twice the input AC input voltage level. Rectifier circuit system, or voltage doubler rectifier circuit that generates a rectified smoothed voltage corresponding to approximately twice the AC input voltage based on the AC input voltage level, and full wave that generates a rectified smoothed voltage corresponding to the AC input voltage level A rectifier circuit system capable of switching to the rectifier circuit is provided. A rectifying smoothed voltage output from the rectifying circuit system is input to perform a switching operation, including the rectifying circuit system and a primary side series resonant circuit formed by series connection of a primary winding of an insulating transformer and a series resonant capacitor. The power factor is improved based on the current resonance type switching converter that outputs a DC output voltage from the secondary side of the isolation transformer and the switching output fed back from the current resonance type switching converter to the rectification current path. The power factor correction circuit is provided in parallel with the switching element of the current resonance type switching converter, and the switching drive is performed using the switching drive power of the current resonance type switching converter, and the switching output is supplied to the power factor correction circuit. And a switching circuit provided to It was possible to configure the switching power supply circuit combined.

According to the above configuration, for example, a switching circuit in which two switching elements are half-bridge coupled is provided in parallel with a set of switching elements of the half-bridge coupled current resonance converter, and this switching is performed. It is possible to feed the switching output back to the rectification path by the output of the circuit to improve the power factor, and thereby the switching current is branched and flows in parallel to the two switching elements. Therefore, for example, the level of the switching current flowing through the switching element is reduced accordingly.

[0024]

FIG. 1 is a circuit diagram showing an embodiment of a switching power supply circuit according to the present invention. In the power supply circuit shown in this figure, the current resonance type converter is of the separately excited type in which two switching elements are half-bridge coupled in the same manner as the power supply circuit of FIG. 8, and the same parts as those in FIG. And the description is omitted.

In the power factor correction rectifier circuit 10 of this embodiment, two switching elements Q ₁₃ and Q ₁₄ are provided. These switching elements Q ₁₃ and Q ₁₄ are, for example, M
A switching element Q ₁₁ , Q used in the current resonance type converter shown in this figure, which uses an OS-FET or the like.
_It may be the same type as ₁₂ .

The switching element Q ₁₃ has its drain connected to the positive electrode (rectified and smoothed voltage line) of the smoothing capacitor Ci, and its source connected to the drain of the switching element Q ₁₄ . The source of the switching element Q ₁₄ is grounded to the primary side ground. Then, from the oscillation drive circuit 2, switching drive signal lines are connected in parallel to the switching elements Q ₁₁ and Q ₁₃ , and similarly, switching drive signal lines are connected in parallel to the switching elements Q ₁₂ and Q ₁₄ . Are connected. Clamp diodes D _D3 and D _D4 are connected in parallel between the drains and sources of the switching elements Q ₁₃ and Q ₁₄ , respectively, in the direction shown in the figure. That is, the switching elements Q ₁₃ and Q ₁₄ form a half-bridge-coupled switching circuit between the rectifying and smoothing voltage line and the primary side ground, as well as the switching elements Q ₁₁ and Q _12.
11 and Q ₁₂ are provided in parallel with the switching circuit formed by the half-bridge coupling. Therefore,
The switching operation includes switching elements Q ₁₁ / Q ₁₃
When the pair is ON, the pair of switching elements Q ₁₂ / Q ₁₄ is turned OFF, so that the ON / OFF operation is repeated in synchronism with alternate timing.

In this case, the switching output points of the half bridge-coupled switching elements Q ₁₃ and Q ₁₄ ,
Source of switching element Q ₁₃ and switching element Q ₁₄
Is connected to the end of the choke coil CH ₁ forming the second series resonance circuit. For example, in the power supply circuit shown in FIG. 8 described above as the prior art, the end of the choke coil CH ₁ of the power factor correction rectifier circuit 20 is the switching element Q _{11 of} the switching converter,
Directly connected to the switching output point of Q _12, the switching output of the switching element Q _11, Q ₁₂ has been configured to be supplied to the second series resonant circuit. On the other hand, in the present embodiment, the switching outputs of the switching circuits of the half bridge-coupled switching elements Q ₁₃ and Q ₁₄ are supplied to the second series resonance circuit. .

Therefore, in the power factor correction rectifier circuit 10 of the present embodiment, the switching outputs of the switching elements Q ₁₃ and Q ₁₄ are connected to the connection point of the rectifier diodes D ₁₁ and D ₁₂ via the second series resonance circuit. The switching voltage applied to the rectification path and superimposed on the rectification path on the basis of this switching output causes the rectification diodes D ₁₁ and D ₁₂ to intermittently operate the rectification current in the switching cycle. Then, the conduction angle of the AC input current is expanded by the same operation as described with reference to FIG. 8 to improve the power factor.

FIG. 2 is a waveform diagram showing the operation of the main part of the power supply circuit of the present embodiment shown in FIG. 1 by the switching cycle in comparison with the power supply circuit of FIG. 8 which is the prior art. In this case, the power supply circuits of FIG. 1 and FIG.
10 Ap-p at a switching cycle of s (100 KHz)
It is assumed that the operation is shown in the case where the required elements are selected so as to obtain the series resonance current of the level.
For example, in the power supply circuits of FIGS. 1 and 8, the series resonance currents I _o and I _o flowing through the first and second series resonance circuits are
_In both _OA , as shown in FIG. 2A, a sine wave having the same waveform is obtained, which corresponds to the current resonance type operation.

The switching current I _Q11 flowing in the switching element Q ₁₁ in the power supply circuit of FIG. 8 is the series resonance circuit I of the first and second series resonance circuits as described above.
_{Since O} and I _OA are combined to flow, as shown in FIG. 2C, a waveform of 10 Ap flows in a half cycle of the switching cycle in which the switching element Q ₁₁ is turned on. The switching current I _Q12 flowing through the switching element Q ₁₂
Although not shown, the same waveform as in FIG. 2C is phase-shifted by a half cycle of the switching cycle.

On the other hand, in the present embodiment, as described above, the pair of switching elements Q ₁₁ / Q _{13 and} the pair of switching elements Q ₁₂ / Q ₁₄ are turned on / off in synchronization with each other at alternate timings. Perform switching. That is, the series resonance current I _O flows through the switching elements Q ₁₁ and Q ₁₂ on the switching converter side, and the series resonance current I _OA flows through the switching elements Q ₁₃ and Q ₁₄ that supply the switching output to the power factor correction rectifier circuit 10. In addition, they are configured independently of each other. Therefore, switching elements Q ₁₁ and Q ₁₃
The switching currents I _Q1 and I _Q3 flowing in the
As shown in (b), a current waveform with a level of 5 Ap is obtained during the period when each switching element is turned on. Although not shown, the switching current I _Q2, I _Q4 flowing through the switching element Q ₁₂ and Q ₁₄ is made to that the waveform, respectively, in FIG 2 (b) half cycle phase of the switching period.

When the present embodiment is compared with the power supply circuit shown in FIG. 8 as the prior art, the switching current level flowing in the switching element in the power supply circuit of FIG. 8 is as shown in FIG. 2 (c). The power supply circuit shown in FIG. 1 according to the present embodiment is different from that shown in FIG.
As shown in (b), the level will be reduced to almost half, that is, 5 Ap. Therefore, in the power supply circuit of FIG. 1, the switching elements Q ₁₁ , Q ₁₂ , Q ₁₃ , and Q ₁₄ are 8
It is possible to select an inexpensive general-purpose product with an A / 400V, TO-220 type resin-molded insulating package, and the number of switching elements is increased by 2 compared to Fig. 8, but the overall cost is lower. Can be configured. Further, since the switching loss in the switching element is also reduced by reducing the switching current amount, for example, when the load power is within about 240 W, the heat dissipation plate becomes unnecessary.

In the power supply circuit of FIG. 8, the maximum load power that can be handled is limited to about 250W. This means that the current level flowing into the switching element increases as the load power increases, but in the power supply circuit of FIG. 8, it is costly to select a switching element having a capacity corresponding to the condition of the load power of 250 W or more. Because it was considered difficult. On the other hand, in the present embodiment, the switching current is about 1/2 of that of the power supply circuit of FIG.
300W even with the above-mentioned general-purpose switching element
It is possible to handle the maximum load power up to a certain degree.

FIG. 3 is a circuit diagram showing another embodiment of the present invention. The same parts as those in FIGS. 1 and 8 are designated by the same reference numerals and the description thereof will be omitted. In the present embodiment, as described below, switching control is performed so as to perform a double voltage rectification operation in an AC100V system and a full-wave rectification operation in an AC200V system.
As a result, a rectified and smoothed voltage of almost the same level can be obtained between the AC100V system and the AC200V system, and this is applied to a so-called wide range power factor correction converter circuit that can be shared by the AC100V system and the AC200V system. Will be suitable. Therefore, a configuration for switching the rectifying operation in the present embodiment will be described.

In the power factor correction rectifier circuit 11 shown in this figure, a normal mode low-pass filter (L _N , C) is used.
_The bridge rectifier circuit D _1A is connected to the latter stage of _N ). The bridge rectifier circuit D _1A is formed by four fast recovery type diodes D _F1 , D _F2 , D _F3 and D _F4 . In this case, resonance capacitors C ₂ and C ₂ are connected in parallel to the fast recovery type diodes D _F1 and D _F2 , respectively.

The positive output terminal of the bridge rectifier circuit D _1A (connection point of the fast recovery type diodes D _F2 and D _F4 ) is connected to the positive terminal of the smoothing capacitor Ci _A. The positive input terminal (the connection point of the fast recovery type diodes D _F1 and D _F2 ) is connected to the second series resonant circuit including the second series resonant capacitor C _1A and the choke coil CH ₁ . The negative input terminal is connected to a connection point of the smoothing capacitors Ci _A and Ci _B via a switch S of an electromagnetic relay RL which will be described later.

In the electromagnetic relay RL, the opening / closing of the switch S is controlled according to the energized state of the drive unit R _D. The energization control of the drive unit R _D of the electromagnetic relay RL is performed by the voltage detection circuit 11.

In the voltage detection circuit 11, a half-wave rectification circuit composed of a rectification diode D ₅ and a smoothing capacitor C ₄ is connected to the commercial AC power supply AC, and a DC voltage corresponding to the input AC input voltage level is generated. Is generated. Between the DC voltage line and the primary side ground, a voltage dividing resistor R ₁ ,
R ₂ is connected in series, and the anode of the Zener diode ZD is connected to the voltage dividing point. In the case of the present embodiment, for example, when an AC input voltage of AC 150 V or more is input, the DC voltage of the half-wave rectifier circuit (D ₅ , C ₄ ) is divided by the voltage dividing resistors R ₁ and R ₂ . The Zener diode ZD ₂ is set to be conductive depending on the voltage value. The cathode of the Zener diode ZD is connected to the base of the transistor Q ₅ . Further, a capacitor C ₅ is provided between the base of the transistor Q ₅ and the primary side ground.
And resistor R ₃ are connected in parallel. Insulation transformer P
The DC low voltage generated by the half-wave rectifier circuit composed of the tertiary winding N _{3 of} IT, the rectifying diode D ₃ and the smoothing capacitor C ₃ is applied to the emitter of the transistor Q ₅ via the drive unit R _D of the electromagnetic relay RL. On the other hand, it is supplied as an operating power source for driving the electromagnetic relay. A diode D ₄ for supplying a reverse current is connected in parallel to the drive unit R _D.

The switching of the rectifying operation of the voltage detection circuit 11 having the above configuration is performed as follows. For example, AC1
When an AC input voltage of AC 150 V or less is supplied as the 00 V system, the Zener diode ZD does not conduct, so that the base current of the transistor Q ₅ is the resistance R ₃
And is turned on. As a result, the emitter current is conducted to the relay drive unit R _D of the electromagnetic relay RL. Then, the switch S is turned on by the exciting action of the relay driver R _D.

When the switch S is turned on, there is conduction between the connection point of the smoothing capacitors Ci _A -Ci _B and the negative input terminal of the bridge rectifier circuit D _1A (connection point of the fast recovery diodes D _F3 and D _F4 ). However, as a result, a voltage doubler rectifier circuit equivalent to the rectifier circuit system shown in FIG. 1 is formed. That is, while the AC input voltage V _AC is positive, the commercial power source AC rectified by the rectifying diode D _F1 is used as the smoothing capacitor C.
i _A is charged to generate a rectified and smoothed voltage of Ei level across the smoothing capacitor Ci _A , and a commercial power supply AC rectified by a rectifier diode D _F1 while the AC input voltage V _AC is negative.
To the smoothing capacitor Ci _B ,
_A rectified and smoothed voltage of Ei level is generated across i _A.
As a result, the smoothing capacitors Ci _A -C connected in series are connected.
A rectified and smoothed voltage of 2Ei level is obtained at both ends of i _B , and in actuality, it is about 2 times that of the AC100V system.
A rectified and smoothed voltage of 00V system can be obtained.

On the other hand, the AC200V system is AC
When an AC input voltage of 150 V or more is supplied, the Zener diode ZD ₂ of the voltage detection circuit 11 is turned on to raise the base potential of the transistor Q _{5 to} a predetermined level or more so that the base current does not flow. , The transistor Q ₅ is turned off. Therefore, the emitter current of the transistor Q ₅ stops flowing through the drive unit R _D , the switch S is turned off, and the smoothing capacitor C
The connection point between i _{A and} Ci _B and the negative input terminal of the bridge rectifier circuit D _1A are open. In this case, commercial power supply A
Four high speed recovery type diode of C a bridge rectifier D _1A performs full-wave rectification by _{(D F1, D F2, D} F3, D F4), charging the series connection of the smoothing capacitor Ci _A - Ci _B Full-wave rectifying and smoothing operation. Therefore, the rectified smoothed voltage of 200V system corresponding to the AC input voltage is obtained across the series connected smoothing capacitors Ci _A -Ci _B.

As described above, the AC 150 is used in this embodiment.
Below V (100V AC system), double voltage rectification operation is performed,
Switching control is performed so that full-wave rectification operation is performed at AC 150 V or higher (AC 200 V system), so that the input AC input voltage level is 100 V system and 200 V system.
A rectified and smoothed voltage of 200 V system can be obtained in both cases.

The rectified and smoothed voltage of the 200 V system is supplied as an operating power source of the switching converter in the subsequent stage, but in the present embodiment, a self-excited half bridge coupling type current resonance type converter is used.

The switching converter of this power supply circuit is provided with switching elements Q ₁ and Q _{2 formed} by two bipolar transistors half-bridge coupled as shown in the figure, and connected between the connection point on the positive side of the smoothing capacitor Ci and the primary side ground. To the respective collectors and emitters. This switching element Q ₁ , Q ₂
Between the collector and the base of the starting resistance R _S1 ,
R _S2 is inserted and the base currents (drive currents) of the switching elements Q ₁ and Q ₂ are adjusted by the resistors R _B1 and R _B2 . Further, clamp diodes D _D1 and D _D2 are respectively inserted between the bases and emitters of the switching elements Q ₁ and Q ₂ . The resonance capacitors C _B1 and C _B2 are the drive windings N _B1 and N _B of the drive transformer PRT described below.
Together with _B2 , it forms a series resonant circuit for self-excited oscillation.

Drive transformer PRT (Power Regulati
ng Transformer) drives the switching elements Q ₁ and Q ₂ and variably controls the switching frequency.
In the case of this drawing, the drive windings N _B1 , N _B2 and the resonance current detection winding N _D are wound, and the control winding N _C is wound in a direction orthogonal to each of these windings. It is said to be a type saturable reactor. One end of the drive winding N _B1 of the drive transformer PRT has a resistor R _B1 and a resonance capacitor C _B1.
Is connected to the base of the switching element Q ₁ through, and the other end is connected to the emitter of the switching element Q ₁ . Further, one end of the drive winding N _B2 is grounded and
The other end is connected to the base of the switching element Q ₂ via a resistor R _B2 and a resonance capacitor C _{B2 so} that a voltage having a polarity opposite to that of the drive winding N _B1 on the switching element Q ₁ side is output. There is.

In the first series resonance circuit (C ₁ , N ₁ ) of the present embodiment, one end of the primary winding N ₁ of the insulating transformer PIT is connected to the switching element via the resonance current detection winding N _D. to Q ₁ emitter and the switching element Q ₂ collector contacts, and are connected to the contacts of the collector of the emitter and the switching element Q ₃ of the switching element Q _3, it is adapted switching output is obtained.

The control circuit 1 compares the DC voltage output E ₁ on the secondary side with a reference voltage, and supplies a DC current corresponding to the error to the control winding N _C of the drive transformer PRT as a control current I _C. Error amplifier.

As a basic switching operation of the switching power supply having the above configuration, when a commercial AC power supply is first turned on, a base current is applied to the bases of the switching elements Q ₁ and Q ₂ through, for example, the starting resistors R _S1 and R _S2. However, if the switching element Q ₁ is turned on _first , the switching element Q ₂ is controlled to be turned off. The output of the switching element Q ₁ is the second series resonance circuit and the resonance current detection winding N _D.
A resonance current flows by branching to the first series resonance circuit via the control circuit so that the switching element Q ₂ is turned on and the switching element Q ₁ is turned off in the vicinity where the resonance current becomes zero. The reverse resonant current flows from the preceding through the switching element Q _2. Thereafter, a self-excited switching operation in which the switching elements Q ₁ and Q ₂ are turned on alternately is started. As described above, the switching elements Q ₁ , Q ₂
By alternately opening and closing the insulation transformer P
A drive current close to the resonance current waveform flows in the primary winding N ₁ of IT, and an alternating output is obtained in the secondary winding N ₂ on the secondary side. Then, based on this alternating voltage, the secondary side DC output voltages E ₁ and E ₂ are obtained as in FIG.

When the DC output voltage E ₁ on the secondary side drops, the control circuit 1 controls the current flowing through the control winding N _C so that the switching frequency becomes low (close to the resonance frequency). Is controlled to increase the drive current flowing through the primary winding N ₁ to achieve a constant voltage (switching frequency control method).

Next, the power factor correction rectifier circuit 1 of the present embodiment
The power factor improving operation by 1 will be described. The power factor correction rectifier circuit 11 is provided with a switching circuit including switching elements Q ₃ and Q ₄ that are half-bridge coupled, and a half-bridge coupled switching provided on the switching converter side as in the case of FIG. 1. It is connected in parallel with the set of elements Q ₁ and Q ₂ . These switching elements Q ₃ and Q ₄ include switching elements Q ₁ and
A bipolar transistor of the same type as Q ₂ may be selected. For these switching elements Q ₃ and Q ₄ , the resonance capacitors C _B3 and C _B4 , the damping resistors R _B3 and R _B4 , the clamp diodes D _B3 and D are connected by the same connection configuration as the switching elements Q ₁ and Q _{2. B4} is connected, and the drive windings N _B1 and N _B2 are connected to the switching element Q.
₁ and Q ₂ are commonly connected, and these elements form a self-excited oscillation circuit for switching-driving the switching elements Q ₃ and Q ₄ by self-excitation. Therefore, the switching element Q ₃ by the same timing as the switching element Q _1, a switching element Q ₄ are at the same timing as the switching element Q _2, so that switching to repeat the on / off alternately performed.

[0051] Then, the switching output point of the switching elements Q _3, Q ₄ is connected to the second series resonant circuit consisting of the choke coil CH ₁ and second series resonant capacitor C _1A, the switching element Q _3, Q The switching output of ₄ is supplied to the choke coil CH ₁ .

In the power factor correction circuit 11 configured as above, the switching outputs of the switching elements Q ₃ and Q ₄ obtained by the inductance of the choke coil CH ₁ are the electrostatic output of the second series resonance capacitor C _1A . The voltage is _{fed back} through the capacitive coupling to the connection point of the rectifying diodes D _F1 and D _F2 .

Therefore, the AC100 according to the present embodiment.
During the voltage doubler rectification operation in the V system, the power factor correction rectification circuit 11
The connection form of the rectifier circuit system is equivalent to the case of the power factor correction rectifier circuit 10 shown in FIG. 1, and the power factor is improved by the same operation as described in FIG.

Further, during the full-wave rectifying and smoothing operation corresponding to the AC200V system, the switching voltage is superimposed on the rectified output through the fast recovery diode D _F2 during the period when the AC input voltage is positive, and the fast recovery is performed during the period when the AC input voltage is negative. The switching voltage is superimposed on the rectified output via the type diode D _F1 , and the rectified current operates so as to be intermittent in the switching cycle. Thus, the same action as described in FIG. 1, also to the rectification current flows through the series connected smoothing capacitors Ci _A - Ci ac input voltage level is low period AC200V system than the voltage across the _B, As a result, the conduction angle of the AC input current is expanded and the power factor is improved.

In this embodiment, the resonance capacitors C ₂ and C ₂ are connected in parallel to the fast recovery type diodes D _F1 and D _F2 , respectively. ₂ and C ₂ have the same operation as in the previous embodiment, and suppress the voltage across the smoothing capacitors Ci _A -Ci _B connected in series at a light load.

FIG. 4 shows a power factor characteristic of the switching power supply circuit having the configuration shown in FIG. 3 with respect to an AC input voltage. In the present embodiment, as described above, the AC15
Since the rectification circuit system is switched with 0 V as the boundary, the power factor characteristic is switched stepwise between AC 150 V and above and below AC 150 V as shown in this figure. And
In this case, when the AC input voltage V _AC = 100V, the power factor P
F = 0.9 is obtained, and when the AC input voltage V _AC = 230 V, the power factor PF = 0.90 is obtained so that the inductance L _O of the choke coil CH ₁ and the series resonance capacitor C
The capacitances of ₁ and C _1A are selected so that an appropriate power factor can be obtained in the AC100V system and the AC200 system.

Further, in the present embodiment, when the AC input voltage V _AC is supplied at the cycle shown in the waveform diagram of FIG. 5A, the full-wave rectification operation in which the _AC input voltage V _AC is 230 V _AC The AC input current I _O at the time and at the time of voltage doubler rectification at AC 100 V are both shown in FIG. In this case, the AC23
AC input current level at 0V is almost 1 at 100V AC.
The level is about / 2. In addition, in practice, the conduction angle is enlarged to the extent that the above power factor is obtained.

When the power factor correction rectifier circuit 11 is configured as described above, it is possible to obtain a more suitable switching power supply circuit for sharing the AC100V system and the AC200V system, and each switching element. Since the level of the switching current flowing through the device is reduced, it is possible to obtain the same effect as that of the previous embodiment. In this embodiment, a semiconductor switch such as a triac may be used instead of the electromagnetic relay RL.

FIG. 6 is a circuit diagram showing the configuration of a switching power supply circuit according to still another embodiment of the present invention.
The same parts as those in FIGS. 2 and 7 are designated by the same reference numerals and the description thereof will be omitted. In the power factor correction rectifier circuit 12 shown in this figure,
As in the embodiment of FIG. 1, the voltage doubler rectifier circuit is provided. Further, in the present embodiment, the filter choke coil L _N and the rectifying diode D ₁₁ (cathode),
Choke coil C between the connection points of D ₁₂ (anode)
H ₂ is inserted in series and acts as a load for the switching output supplied from the second series resonant circuit. In this case, the resonance capacitor C ₂ is the choke coil CH ₂
Are connected in parallel with respect to each other to form a parallel resonance circuit together with the inductance L _S of the choke coil CH ₂ , but the operation thereof is the same as that of each of the above-described embodiments. Further, the filter capacitor C _N is provided in parallel with the commercial AC power supply AC so that one end thereof is connected between the connection points of the filter choke coil L _N and the choke coil CH ₂ , but such a connection is provided. Depending on the form, a normal mode low-pass filter is formed together with the filter choke coil.

[0060] In the power factor improving rectification circuit 12 of the above configuration, the switching output of the switching circuit of the second series resonant circuit (CH _1, C _1A) is supplied to the switching element Q _3, Q ₄ are, choke coil CH ₁ And the choke coil CH ₂ with magnetic coupling interposed therebetween, the rectified output voltage obtained by the inductance L _S of the choke coil CH ₂ is superimposed as a switching voltage. _{In 11} and D ₁₂ , the rectified current operates so as to be intermittent in the switching cycle. Therefore, thereafter, the conduction angle of the AC input current is expanded by the same operation as described with reference to FIG. 1 to improve the power factor.

In this embodiment, a half-bridge coupling type self-exciting current resonance type converter is provided as in the above embodiments, but the constant voltage control method is different. In the case of the power supply circuit shown in this figure, the drive transformer CDT has a configuration in which the control winding N _C is not wound, and therefore the switching frequency is fixed to drive the switching elements Q ₁ and Q ₂ . In this case, in the insulating transformer PRT, the control winding N _C is provided so that the winding directions of the primary winding N ₁ and the secondary winding N ₂ are orthogonal to each other. In this configuration, the control circuit 1 supplies the control current to the control winding N _C as the control current, which is a DC current having a level that is varied according to the variation of the DC output voltage E ₁ on the secondary side. As a result, the leakage magnetic flux is changed in the insulating transformer PRT to change the inductance of the primary winding N ₁ . Then, due to this inductance change, the resonance frequency of the primary side series resonance circuit is variably controlled with respect to the switching frequency, which makes it possible to make the DC output voltage on the secondary side a constant voltage (series resonance frequency control method). With such a configuration as well, the switching current level is reduced, and the same effects as those of the above-described respective embodiments are obtained.

FIG. 7 is a circuit diagram showing the configuration of a switching power supply circuit according to still another embodiment of the present invention.
The same parts as those in FIG. 1, FIG. 3 and FIG. In the power factor correction rectifier circuit 13 shown in this figure, the filter choke coil L _N and the rectifier diode D
_The secondary winding Ni of the magnetic coupling transformer MCT is inserted in series between the connection points of ₁₁ (cathode) and D ₁₂ (anode). Further, the resonance capacitor C ₂ is connected in parallel to the secondary winding Ni of the magnetic coupling transformer MCT to form a parallel resonant circuit together with the inductance Li of the secondary winding Ni, and the above-described respective embodiments. Similarly, the rise of the rectified and smoothed voltage is suppressed when the load becomes light.

[0063] The magnetic coupling transformer MCT, for example the core formed by a ferrite material or the like, the secondary winding Ni and the primary winding N _O 1: with magnetically tightly coupled by a turns ratio It is formed. In this case, the primary winding N _O corresponds to the winding of the choke coil CH ₁ shown in each of the above embodiments and has an inductance L _O. Then, one end thereof is connected to the second series resonant capacitor C _1A to form a second series resonant circuit. The other end is grounded to the primary side ground.

In the power factor correction circuit 13 thus configured, the magnetic coupling transformer MCT is generated based on the switching output supplied from the switching elements Q ₁₃ and Q _14.
A switching voltage is obtained at the inductance L _O of the primary winding N _O. In the magnetic coupling transformer MCT, the switching voltage obtained in the primary winding N _O is transmitted to the secondary winding Ni via the magnetic coupling. Since the secondary winding Ni of the magnetic coupling transformer MCT is inserted in the rectification path, the switching voltage excited in the secondary winding Ni causes the switching voltage to be superimposed on the rectified output voltage of the rectification path. It As a result, similarly to each of the above-described embodiments, as a result of prompting the intermittent operation of the rectification current by the rectification diodes D ₁₁ and D ₁₂ , the conduction angle of the AC input current is expanded and the power factor is improved. And
Also in this embodiment, since the switching current flowing through each switching element is reduced, it is possible to obtain the same effect as that of the power supply circuit of the previous embodiment shown in FIG.

The switching power supply circuit of the present invention is
The combination of various types of the rectifier circuit system, the power factor correction circuit, and the current resonance type converter presented as the present invention is not limited to the configuration shown in each of the above embodiments, and for example, as shown in FIG. In contrast to the configuration of the embodiment shown in FIG. 7 and the like, it is naturally possible to provide a rectifying circuit system capable of switching a voltage doubler / full-wave rectification operation and configure the switching power supply circuit of the present invention. . Further, the circuit configurations of the respective parts shown as the above-described respective embodiments can be changed according to actual usage conditions and the like. Moreover,
Although not shown, it may be possible to adopt a full bridge coupling type (formed by four switching elements) as the current resonance converter. Further, even in the details of the circuit configurations shown in the respective drawings as the above-mentioned respective embodiments, the connection form may be appropriately changed according to the actual use conditions and the like.

[0066]

As described above, according to the present invention, as a switching power supply circuit having a voltage doubler rectifying circuit and capable of supporting a relatively heavy load, for example, a current resonance type switching converter of a half bridge coupling type is used. Then, the switching output of the half-bridge coupling type switching circuit that performs switching using the switching drive power is fed back to the rectification path via the second series resonance circuit to improve the power factor. The peak level of the switching current flowing through the switching element should be reduced to about half of that in the case where the switching circuit is not provided, while suppressing the AC ripple component at the time of low AC input voltage. become. As a result, a general-purpose product of an inexpensive insulating package can be used as the switching element, and the heatsink can be attached by a simple method without using silicon rubber or the like, so that the cost of the entire power supply circuit can be reduced. It has the effect that it becomes possible. Further, since the switching loss is reduced, the heat radiation plate for the switching element is not required especially when a light load is supported, and further cost reduction and circuit miniaturization are promoted. In addition, along with this, it is possible to expand the maximum load power that can be handled while selecting a general-purpose switching element.

The AC input voltage of the rectifier circuit system is AC1.
When the double voltage rectification / full-wave rectification circuit can be switched between the 00V system and the AC200V system, the AC100V system and the A
With C200V, almost the same power factor characteristics can be obtained. Therefore, it is possible to provide a more suitable power supply circuit compatible with a so-called wide range, which can be commonly used for both AC100V system and AC200V.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit as an embodiment of the present invention.

FIG. 2 is a waveform diagram showing an operation of the switching power supply circuit of the present embodiment in a switching cycle.

FIG. 3 is a circuit diagram showing a configuration of a switching power supply circuit according to another embodiment.

FIG. 4 is a diagram showing a power factor characteristic of the switching power supply circuit of the embodiment shown in FIG.

5 is a waveform diagram showing the operation of the switching power supply circuit of the embodiment shown in FIG. 3 in commercial power supply cycles.

FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit as still another embodiment.

FIG. 7 is a circuit diagram showing a configuration of a switching power supply circuit according to still another embodiment.

FIG. 8 is a circuit diagram showing a configuration of a switching power supply circuit as a conventional example.

[Explanation of symbols]

1 Control circuit 2 Oscillation drive circuit 3 Start-up circuit 10, 11, 12, 13 Power factor correction rectifier circuit 11A Voltage detection circuit D _1A Bridge rectifier circuit D ₁₁ , D ₁₂ , DF ₁ , DF ₂ , DF ₃ , DF ₄ rectifier diode (High-speed recovery type) Ci _A , Ci _B Smoothing capacitor PIT (PRT) Isolation transformer CDT (PRT) Drive transformer Q ₁ , Q ₂ , Q ₃ , Q ₄ , Q ₁₁ , Q ₁₂ , Q ₁₃ , Q ₁₄ Switching element C ₁ series resonance capacitor N ₁ primary winding MCT magnetic coupling transformer C _1A second series resonance capacitor CH ₁ , CH ₂ choke coil L _N filter choke coil C _N filter capacitor

Claims (11)

[Claims] 1. A rectifying means for rectifying a rectified and smoothed voltage corresponding to approximately twice an AC input voltage level which is obtained by rectifying a voltage of a commercial power source, and a series connection of a primary winding of an insulating transformer and a series resonant capacitor. A current resonance type switching circuit which comprises a primary side series resonance circuit formed by the above, performs a switching operation by inputting the rectified smoothed voltage output from the rectification means, and outputs a DC output voltage from the secondary side of the insulation transformer. Converter means, power factor improving means adapted to improve the power factor based on the switching output fed back from the current resonance type switching converter means to the rectified current path, and the switching element of the current resonance type switching converter means. It is provided in parallel and uses the switching drive power of the current resonance type switching converter means. And a switching circuit provided so as to supply the switching output to the power factor improving means, and a switching power supply circuit. 2. A voltage doubler rectifier circuit for generating a rectified and smoothed voltage corresponding to approximately twice the AC input voltage level based on the AC input voltage level input to the commercial power supply, and a rectifier corresponding to the AC input voltage level. A rectifying means that can be switched to a full-wave rectifying circuit that generates a smooth voltage, and a primary-side series resonant circuit formed by a series connection of a primary winding of an insulating transformer and a series resonant capacitor. A current resonance type switching converter means for inputting the output rectified and smoothed voltage to perform a switching operation and outputting a DC output voltage from the secondary side of the insulation transformer, and a rectification current path from the current resonance type switching converter means. A power factor improving means adapted to improve the power factor based on the switching output fed back, and the current resonance type switching converter A switching circuit that is provided in parallel with the switching element of the switching means, is driven to perform switching using the switching drive power of the current resonance type switching converter means, and supplies the switching output to the power factor improving means. A switching power supply circuit comprising: 3. The switching power supply according to claim 1, wherein the current resonance type switching converter means and the switching circuit are configured by half bridge coupling two switching elements. circuit. 4. The power factor improving means uses a rectifying element forming the rectifying means as a high-speed recovery type, and a normal mode low-pass filter including a filter choke coil and a filter capacitor provided for a commercial power supply, A choke coil connected to the output of the switching circuit; and a coupling capacitor provided so as to form a series resonance circuit together with the choke coil and apply the switching output supplied to the choke coil to the rectified current path. The switching power supply circuit according to claim 1, 2 or 3, wherein the switching power supply circuit is configured. 5. The switching power supply circuit according to claim 4, wherein a resonance capacitor is provided in parallel with the rectifying element forming the rectifying means. 6. The power factor improving means uses a rectifying element forming the rectifying means as a high-speed recovery type, and a normal mode low-pass filter including a filter choke coil and a filter capacitor provided for a commercial power supply, A first choke coil that is inserted in series in the rectified current path, a second choke coil that is connected to the output of the switching circuit, and a second choke coil that forms a series resonance circuit, and the second choke coil is formed. The switching power supply according to claim 1, 2 or 3, further comprising: a coupling capacitor provided so as to apply the switching output supplied to the coil to the rectified current path. circuit. 7. The switching power supply circuit according to claim 6, wherein a resonance capacitor is provided in parallel with the first choke coil. 8. The power factor improving means uses a rectifying element forming the rectifying means as a high speed recovery type, and a normal mode low pass filter including a filter choke coil and a filter capacitor provided for a commercial power source, A series resonance winding connected to the output of the switching circuit, a coupling capacitor provided so as to form a series resonance circuit together with the second choke coil, the series resonance winding, and the series resonance winding inserted in series in a rectification current path. The magnetic coupling transformer formed by magnetically tightly coupling the superposed windings, and the switching according to claim 1 or 2 or 3. Power supply circuit. 9. The switching power supply circuit according to claim 8, wherein a resonance capacitor is provided in parallel with the superposed winding. 10. The switching device according to claim 1, wherein the current resonance type switching converter means is of a self-excited type having a self-excited oscillation circuit and performing a switching operation. Power supply circuit. 11. The switching according to claim 1, wherein the current resonance type switching converter means is of a separately excited type for performing a switching operation by including a separately excited oscillation circuit. Power supply circuit.