JPH09247934A - Current resonance type converter circuit and power factor improving converter circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To eliminate variations in characteristics of a self-excited current resonance type converter such as the storage time of a bipolar switching device, the reverse recovery time of a clamping diode, etc., and improve the design efficiency and the reliability of the circuit. SOLUTION: A clamping diode DB1 and a Zener diode DZ1 are connected in series between the base and emitter of a switching device Q1 . In the same way, a clamping diode DB2 and Zener diode DZ2 are connected in series between the base and emitter of a switching device Q2 . A parallel resonance circuit (NB1 /Cd1 and NB2 /Cd2 ) is used as a self-excited oscillation circuit which drives the switching devices Q1 and Q2 to obtain a high level driving voltage. With this constitution, a base current IB is not shunted into the clamping diode to keep a driving voltage and a minus base current IB at the turning-off time to reduce the storage time Ts of the switching device.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a current resonance type converter circuit used as a switching power supply circuit and a power factor improving converter circuit using a current resonance type converter provided to improve the power factor of the switching power supply circuit. is there.

[0002]

2. Description of the Related Art In recent years, as a power supply device for rectifying a commercial power supply to obtain a desired DC voltage by developing a switching element capable of withstanding a relatively large current and voltage of a high frequency, a switching type power supply is mostly used. It is a device. The switching power supply is used as a power supply for various electronic devices as a high power DC-DC converter while increasing the switching frequency to reduce the size of a transformer and other devices.

In general, when a commercial power supply is rectified, the current flowing through the smoothing circuit has a distorted waveform, which causes a problem that the power factor indicating the efficiency of use of the power supply is impaired. Further, there is a need for a measure for suppressing harmonics generated due to the distorted current waveform.

Therefore, as a power factor improving means for improving a power factor in a switching power supply circuit, there is known a method of providing a PWM control type boost converter in a rectifying circuit system and providing a so-called active filter for bringing the power factor close to one. ing. However, since such an active filter has many factors such as an increase in circuit scale and a high cost due to an increase in the number of components and an increase in size for high EMI countermeasures, it is necessary to reduce the size and cost of the power supply circuit. It is disadvantageous from a viewpoint. It is also known that power loss in an active filter is relatively large.

[0005] Therefore, the applicant has previously switched the rectified output using a self-excited current resonance type converter,
There have been proposed various power factor improving converter circuits configured to increase the conduction angle of the AC input current to thereby improve the power factor. In a power factor improving converter circuit using such a current resonance type converter, since the switching operation of the converter circuit is a current resonance type, low noise is realized and the circuit scale is small as compared with an active filter. Thus, the cost can be reduced accordingly. In addition, power loss is greatly reduced, and power conversion efficiency is improved.

FIG. 9 is a circuit diagram showing an example of a power supply circuit including a power factor correction converter circuit which can be constructed based on the invention previously filed by the present applicant. In the power supply circuit shown in this figure, the commercial AC power supply A
A common mode choke coil CMC and an across capacitor C _L are provided as a noise filter for removing common mode noise with respect to C. Commercial AC power supply A
C is full-wave rectified by the bridge rectifier circuit D ₁ . In this case, the rectification output line of the bridge rectification circuit D ₁
A power factor correction converter unit 20 is provided between the smoothing capacitors Ci, which are smoothing circuits, to improve the power factor as described later.

The switching power supply 1 performs a switching operation by inputting a rectified and smoothed voltage Ei obtained between both ends of a smoothing capacitor Ci, and performs DC output voltages E ₁ and E from a secondary side.
A DC-DC converter that outputs ₂ is provided, and in this case, for example, a switching converter that performs constant voltage control by the PWM method is provided.

In the power factor improving converter section 20 shown in FIG. ₁ , first, a line (rectified output line) between the positive output terminal of the bridge rectifier circuit D1 and the positive terminal of the smoothing capacitor Ci is applied to the filter choke coil L. _N - high speed recovery type diode D ₂ are connected in series. Here, the high speed recovery type diode D ₂ is inserted by the direction in which the anode is a bridge rectifier circuit D ₁ side.
In this case, the connection point of the positive output terminal and said filter choke coil L _N of the bridge rectifier circuit D _1, a filter capacitor C _N is inserted between the positive terminal of the smoothing capacitor Ci, normal mode with filter choke coil L _N Is formed. Further, resonant capacitor C ₂ is connected in parallel for high speed recovery type diode D _2. The resonance capacitor C
The function of ₂ will be described later.

The power factor improving converter section 20 includes a self-excited current resonance type converter using the rectified smoothed voltage Ei as an operating power supply. As the current resonance type converter, switching elements Q ₁ and Q 2 of two bipolar transistors half-bridge-coupled as shown in the figure are used.
_2, and performs a switching operation using the rectified smoothed voltage Ei as an operation power supply. In this case, the switching element Q ₁
The collector is connected to the positive terminal of the smoothing capacitor Ci, the emitter is connected to the collector of the switching element Q _2. The emitter of the switching element Q ₂ is grounded to the primary side ground. The starting resistors R _S1 and R _S1 are connected between the collectors and the bases of the switching elements Q ₁ and Q ₂ , respectively.
_S2 is inserted, and the base current (drive current) of the switching elements Q ₁ and Q ₂ is adjusted by the resistors R _B1 and R _B2 .
Further, the bases of the switching elements Q _1, Q ₂ - each clamp diode between the emitter D _B1, D _B2 is inserted. The reverse recovery time Trr of the clamp diodes D _B1 and D _B2 is long (in this case, 10 μs to 16 μs).
Slow speed diodes are used. And
The resonance capacitors C _B1 and C _B2 , together with the drive windings N _B1 and N _{B2 of} the drive transformer PRT described below, form a series resonance circuit for driving the switching element by self-oscillation. Further, the collectors of the switching elements Q _1, Q ₂ - respectively between the emitters capacitor C _C1, C _C2 are connected in parallel, the switching element Q ₁ which is a rectangular wave, Q
The rise / fall feedback of the switching voltage of ₂ is given a slope to suppress the switching noise.

[0010] Drive transformer PRT (Power Regulati
ng Transformer) drives the switching elements Q ₁ and Q ₂ and variably controls the switching frequency. The drive transformer PRT includes drive windings N _B1 and N _B2 and a winding N _D provided to wind up the drive winding N _B1.
In contrast, a saturable reactor of the orthogonal type provided with a control winding N _C wound so that the winding directions are orthogonal to each other. One end of the drive winding N _B1 has a resistance R _B1 −.
It is connected to the base of switching element Q ₁ via a resonant capacitor C _B1, and the other end is connected to the emitter of the switching element Q _1. One end of the drive winding N _B2 is is grounded to the earth, the other end resistor R _B2 - resonant capacitor C
Is connected to the base of the switching element Q ₂ via the _B2, it is adapted to reverse polarity voltage is outputted to the drive winding N _B1.

The converter transformer CVT has a primary winding N.
It is constructed by winding ₁ and a tertiary winding N ₃ . In this case, the primary winding N ₁ is an inductor for superimposing a switching voltage on the rectification path as described later. One end of the primary winding N ₁ of the converter transformer CVT is connected to the connection point (switching output point) of the emitter of the switching element Q _{1 and} the collector of the switching element Q ₂ via the winding N _D , and the other end is connected. Series resonance capacitor C
_It is connected via ₁ to the connection point between the filter choke coil L _N and the fast recovery diode D ₂ . According to the above connection form, the primary winding N _{1 of the} converter transformer CVT
And the series resonance capacitor C ₁ are connected in series, but a series resonance circuit for making this switching converter a current resonance type by the inductance component of the primary winding N ₁ and the capacitance of the series resonance capacitor C _1. Are formed. A switching output obtained by the switching operation of the switching elements Q ₁ and Q ₂ is supplied to the series resonance circuit, and this switching output is connected to a connection point between the filter choke coil _LN and the high-speed recovery type diode D ₂ . To be applied.

Further, the transistor Q ₁₀ is connected to the control winding N _C
For supplying the control current I _C to In this case, the base of the transistor Q ₁₀ is rectified and smoothed voltage E
i line and is connected to the voltage dividing point of the voltage dividing resistors R _11, R ₁₂ which is inserted in series with between the primary side ground, a DC current corresponding to the level of the rectification smoothed voltage Ei is supplied. Also,
An alternating voltage is generated in the tertiary winding N ₂ of the converter transformer CVT by the switching operation of the switching converter, but a half-wave composed of a rectifying diode D ₃ and a smoothing capacitor C ₃ connected to the tertiary winding N ₃ . A DC voltage of a predetermined level is generated by the constant voltage circuit including the rectifier circuit, the resistor R ₃ and the Zener diode ZD ₁ . This DC voltage is supplied to the collector of the transistor Q ₁₀ via the control winding N _C as an operating power supply of the transistor Q _10. The emitter of the transistor Q ₁₀ is grounded to the primary side ground via a resistor R _13.

The power factor improving converter section 20 is constructed as described above. The switching operation of the current resonance type converter is as follows. First, when the commercial AC power supply is turned on, the base current is supplied to the bases of the switching elements Q ₁ and Q ₂ via the starting resistors R _S1 and R _S2 , for example, but the switching element Q ₁ comes first. If it is turned on, the switching element Q ₂ is controlled to be turned off. And the switching element Q ₁
Output of the resonance current detection winding N _D → primary winding N ₁ →
A resonance current flows through the series resonance capacitor C ₁ , but the switching element Q ₂ is turned on in the vicinity where the resonance current becomes 0,
The switching element Q ₁ is controlled so as to be turned off.
The reverse resonant current flows from the preceding through the switching element Q _2. Hereinafter, switching elements Q ₁ , Q ₂
Are turned on alternately to start a self-excited switching operation. As described above, the switching elements Q ₁ and Q ₂ alternately open and close with the terminal voltage of the smoothing capacitor Ci as the operating power supply, thereby supplying a drive current close to the resonance current waveform to the primary winding N ₁ of the converter transformer CVT. I do. Note that the variable control of the switching frequency by the drive transformer PRT will be described later.

Then, the power factor improving operation in the power factor improving converter unit 20 of the present embodiment is as follows. As described above, when the switching operation of the current resonance type converter is performed, the switching output is supplied to the primary winding N ₁ of the converter transformer CVT. In the converter transformer CVT, the primary winding N ₁
The alternating voltage of the switching cycle generated by the switching output supplied to is applied to the connection point between the filter choke coil L _N and the fast recovery diode D ₂ via the capacitive coupling of the series resonance capacitor C ₁ . The filter choke coil L _N and the high-speed recovery type diode D _2, since that is inserted to the positive output line of the bridge rectifier circuit D _1, the switching voltage applied through the series resonant circuit, the rectifier via the rectifying path The switching voltage is superimposed on the output voltage.
Then, the superposed portion of the switching voltage, a fast-recovery diode D ₂ in the rectified current is inserted in the rectification path that operates intermittently at the switching period is obtained. By this operation, the smoothing capacitor Ci is charged in a state where the switching output is superimposed on the rectified output voltage in the power factor correction converter section 20, and the voltage across the smoothing capacitor Ci is switched by the superimposed amount of the switching voltage. It is made to decrease in a cycle. Therefore, the charging current flows to the smoothing capacitor Ci even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor (rectified smoothed voltage Ei). As a result, the average waveform of the AC input current is made closer to the waveform of the AC input voltage, and the conduction angle of the AC input current is increased, thereby improving the power factor.

Further, resonant capacitor C ₂ connected in parallel to the high speed recovery type diode D _2, for example a filter choke coil L _N and filter capacitor C
Form a parallel resonance circuit with _N. The resonance impedance of the parallel resonance circuit is changed in response to a load change. When the load of the power supply circuit is reduced, the switching voltage fed back to the rectification path is suppressed. As a result, an increase in the terminal voltage of the smoothing capacitor Ci at a light load is suppressed.

Further, in a power factor correction converter configured to feed back a switching output of a current resonance type converter to a rectification path, such as a power factor correction converter circuit as shown in FIG. In this case, the power factor characteristic decreases as the input AC input voltage increases. Therefore, by changing the switching frequency of the current resonance type converter using the drive transformer PRT, the power factor can be controlled to be substantially constant with respect to the input AC input voltage level.

[0017] For example, when the AC input voltage V _AC input to a commercial AC power source of the power supply circuit shown in FIG changes as rose, since also increases rectified smoothed voltage Ei corresponding to, rectification base current supplied to the base of the transistor Q ₁₀ via a voltage dividing resistor R ₁₁ from smoothed voltage Ei is increased. As a result, the transistor Q ₁₀ operates to increase the collector current level, so that the collector current flows as the control current I _S in the control winding N _C connected to the collector. That is, the control current I _S flowing through the control winding N _C is controlled to increase in level as the AC input voltage level increases. In the drive transformer PRT, since the level of the control current I _S becomes large as described above, the drive winding N
_Reduce the inductance of _B1 and _NB2 . This allows
The drive winding N _B1 and the resonance capacitor C _B1 , and the drive winding N _B2
Lowers the resonant frequency of the two sets of self-oscillation driving circuit formed by the resonance capacitor C _B2 and the switching element Q
_11, will be controlled so as to increase the switching frequency of the Q _12. In this case, the switching frequency changes with respect to the resonance frequency of the series resonance circuit, but this changes the feedback amount of the switching output supplied to the series resonance circuit, and in this case, improves the power factor improvement. Will be controlled as follows.

FIG. 10 is a waveform diagram showing the operation of the main part of the power factor correction converter section 20 shown in FIG. 9 in the switching cycle. In this case, the operation related to the switching operation of the switching element Q ₂ side. The waveform is shown. Note that the switching elements Q ₁ and Q ₂ alternately repeat on / off operations, and since the operations are the same in terms of alternating current, the operation waveforms on the switching element Q ₁ side are not shown.

In FIG. 10, FIG. 10A shows the collector-emitter voltage V _CE of the switching element Q ₂ .
FIG. 10B shows the collector current I _C of the switching element Q ₂ . 10C shows the base current I _B of the switching element Q ₂ , and FIG. 10D shows the base-emitter voltage V _BE of the switching element Q ₂ . Further, FIG. 10D shows the drive voltage V _D obtained across the drive winding N _B2 , and FIG. 10F shows the clamp diode D _B2.
The current I _D flowing through In this case, the direction indicated by the arrow in the figure is regarded as the positive direction in which the current I _D flows.

In this case, as can be seen from the fact that the current I _D in FIG. 10 (f) has been at a positive level from the period before the period T11, the drive winding N on the switching element Q ₂ side is seen.
_The drive current (not shown) flowing in _B2 turns to a positive level before the switching element Q ₁ is turned off. However, clamp diode D _B1 from the presence of the reverse recovery time Trr the clamp diode D _B1 as described above, for example, in the period T11, as shown in FIG. 10 (f)
The reverse recovery current flows to the base current I _B so that it does not conduct, and the switching element Q ₂ operates in the period T11.
It is controlled not to turn on until. That is, in the power factor correction converter circuit of the current resonance type, the switching element Q is utilized by utilizing the reverse recovery current of the low speed diode.
It is arranged to avoid the situation where both ₁ and Q ₂ are turned on.

The switching element Q ₂ is a converter transformer C generated when the switching element Q ₁ is off.
The damper current generated by the excitation energy of the primary winding N ₁ of the VT causes it to change from OFF to ON in the period T12. The damper current at this time is as follows: "Primary winding N _1- > series resonance capacitor C _1- > (resonance capacitor C ₂ / fast recovery type diode connected in parallel)-> smoothing capacitor Ci-> primary side ground-> clamp diode D _B2 → base → switching element collector of Q ₂ → winding N _D → primary winding N ₁ of the switching element Q ₂ "
It is made to flow in the loop. Next, in the period T13 in which the base current I _B flows to the emitter, the previous period T1
A reverse recovery current for the damper current at ₂ is generated in the clamp diode D _B2 . Therefore, during this period T13
In, the base current I _B that should flow to the switching element Q ₂ branches and also flows to the clamp diode D _B2 , and its level becomes smaller than the drive current of the drive winding N _B2 . Along with this, the amplitude of the drive voltage V _D of the drive winding N _B2 (the peak level is indicated by V ₁ ) also decreases, but this causes the base current I _B shown in FIG. The level I _{1 of the} negative current (base charge extraction current) also becomes small. Therefore, the storage time Ts of the switching element Q ₂ also becomes long.

[0022]

The switching operation of the power factor correction converter circuit shown in FIG. 9 is, as can be understood from the explanation of the operation waveform diagram of FIG. 10, the switching elements Q ₁ and Q _{2 of} bipolar transistors. Storage time Ts and the reverse recovery time Trr of the low speed type clamp diodes D _B1 and D _B2 . However, the storage time Ts of the switching element and the reverse recovery time Trr
It is known that there is variation and that the difference between the worst condition and the optimum condition is relatively large depending on the value. Therefore, it is necessary to select at least the storage time Ts for the switching element and the reverse recovery time Trr for the clamp diode. Further, while the storage time Ts of the switching element changes exponentially with the temperature rise, the reverse recovery time Trr of the clamp diode changes linearly. For this reason, it is relatively difficult to balance the two and perform good compensation for temperature changes.For example, intermittent oscillation or startup failure due to insufficient drive may occur at low temperatures, and switching may occur at high temperatures. It has been found that the margin for thermal runaway due to both the elements Q ₁ and Q ₂ being turned on / off is reduced.

Further, as explained in FIG. 10 (c),
At the time of turning off the switching element, the amount of negative level current of the base current I _B in the period T13 decreases due to the action of the reverse recovery current on the damper current of the clamp diode generated in the period T12. Therefore, the switching element Q ₂ has a relatively large switching loss due to the fall time Tf. Further, along with this, since the storage time Ts of the switching element also increases, the positive current of the base current I _B is shunted to the clamp diode D _B2 when the switching element is on. For this reason, the base current I _B that should be originally supplied to the base of the switching element Q ₂ decreases, but it is also known that the power loss due to the collector saturation voltage V _{CE (SAT)} increases.

[0024]

In view of the above-mentioned problems, the present invention considers the above-mentioned problems and reduces the variations in the characteristics of the current resonance type converter and the power factor correction converter circuit, expands the temperature guarantee range, and improves the power conversion efficiency. It is an object of the present invention to improve reliability by improving characteristics such as the above, and to eliminate the factor of variation in characteristics of the switching element and the clamp diode as much as possible to improve circuit design efficiency.

For this reason, a bipolar transistor is used as a switching element, and a switching circuit system that includes a self-excited oscillation drive circuit and drives the switching element,
Formed by a series resonant capacitor and a series resonant winding,
In a current resonance type converter including a series resonance circuit to which the switching output of the switching circuit system is supplied, a drive voltage provided so that the drive voltage generated by the self-excited oscillation circuit can be set to a required level. It was decided to have a level setting circuit. The drive voltage level setting circuit is based on the base of the switching element.
It is configured to include a Zener diode element that is inserted between the emitters and is connected in series to the diode element for forming the path of the clamp current and that has the Zener voltage corresponding to the drive voltage level to be set. . As the self-excited oscillation drive circuit, a parallel resonance circuit in which a drive winding and a resonance capacitor are connected in parallel is used. In addition, the circuit portion including the switching element and the drive voltage level setting circuit provided for the switching element is configured as a component of one package.

According to the above construction, for example, the Zener diode element prevents the damper current from flowing in the diode element for forming the path of the clamp current, and the Zener voltage of the Zener diode element drives the switching element. The drive voltage level (amplitude) of the self-excited oscillation drive circuit is determined. However, by setting the drive voltage level to a large value, a negative level base current when the switching element is turned off (base accumulated charge The drawing current) is also increased. As a result, it is possible to reduce the storage time Ts of the switching element which is a bipolar transistor and reduce the influence of the storage time Ts on the operation of the circuit.

[0027]

BEST MODE FOR CARRYING OUT THE INVENTION Embodiments of the present invention will be described below with reference to FIGS. The current resonance type converter circuit of the present invention can be configured as a single switching power supply circuit. However, the embodiments described below are not limited to the current resonance type converter circuit of the present invention. It is assumed that the power factor correction converter is configured by adding a circuit configuration for improvement.

First Embodiment FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit including a power factor correction converter circuit according to a first embodiment of the present invention, which has been shown as a prior art. The same parts as those in FIG. 9 are designated by the same reference numerals and the description thereof will be omitted. The power factor correction converter circuit according to the present embodiment shown in this figure corresponds to the power factor correction converter unit 10. In the power factor correction converter unit 10, as shown in the drawing, a Zener diode D _Z1 and a clamp diode D _B1 are inserted in series between the base and emitter of the switching element Q ₁ . In this case, the anode of the Zener diode D _Z1 is connected to the base of the switching element Q ₁ , and the cathode of the Zener diode D _Z1 is connected to the cathode of the clamp diode D _B1 . The anode of the clamp diode D _B1 is connected to the emitter of the switching element Q _1. In addition, between the base and emitter of the switching element Q ₂ ,
The Zener diode D _Z2 and the clamp diode D _B2 are connected in series and inserted, and the connection form is the same as the Zener diode D _Z1 and the clamp diode D _B1 provided on the switching element Q ₁ side. In the case of the present embodiment, the clamp diodes D _B1 and D _B2 act as reverse current blocking diodes for blocking currents that try to flow in the zener diodes D _Z1 and D _Z2 in the forward direction (anode → cathode), respectively. Have.

Further, in the present embodiment, the drive winding N
_A resonance capacitor C _D1 is provided in parallel with _B1 to form a parallel resonance circuit, and a resonance capacitor C _D2 is connected in parallel with the drive winding N _B2 to form a parallel resonance circuit. To be done. That is, in the present embodiment, the switching elements Q ₁ and Q ₂ have parallel resonances formed by [driving winding N _B1 / resonant capacitor C _D1 ] and [driving winding N _B2 −resonant capacitor C _D2 ], respectively. The circuit is driven by self-excited oscillation, and its switching frequency is also set by these parallel resonant circuits. In this case, the capacitors C _B1 and C _B2 each act as a DC element capacitor.

In the power factor correction converter unit 10 constructed in this way, the circuit configuration for feeding back the switching outputs of the switching elements Q ₁ and Q ₂ to the rectified current path is the same as in the case of FIG. Therefore, the power factor improving operation according to the present embodiment is similar to that described in FIG. Further, the control transformer PRT changes the switching frequency according to the fluctuation of the rectified and smoothed voltage Ei to keep the power factor constant with respect to the change of the AC input voltage level. It will be similar.

FIG. 2 is a waveform diagram showing the operation of the main part in the switching cycle of the power factor correction converter unit 10. In this case, the operation waveform of the main part relating to the switching operation of the switching element Q ₂ side is shown. Has been done. Although not shown in the operation waveform of the switching element Q ₁ side, the switching element Q ₁ is intended to perform the ON / OFF operation alternately with the switching element Q _2, with respect to the operation timing shown in FIG. 2, the switching period In addition, the phases are different at the timing of 1/2 cycle, and the same operation waveform is obtained. Figure 2 (a) ~ (f) are respectively the collector of the switching element Q ₂ - emitter voltage V _CE, the collector current I _C, the base current I _B, the base - emitter voltage V
_BE , the voltage across the drive winding N _B2 (hereinafter referred to as drive voltage) V _D , and the clamp current I _D flowing through the clamp diode D _B2 in the direction of the arrow in FIG.

In FIG. 2, the period T1 is a period during which the switching element Q ₁ shifts from the ON state to the OFF state,
As a result, the electric charge accumulated in the resonance capacitor C ₂ is discharged by the damper current generated in the primary winding N ₁ by the excitation energy of the converter transformer CVT.
At this time, the base-emitter voltage V _BE (see FIG. 10D)
Of the switching element Q
₂ is still off. This prevents both switching elements Q ₁ and Q ₂ from turning on / on. In this case, even after the discharge of the resonance capacitor C ₂ is completed and the period T2 is reached, the damper current continues to flow, and during this period T2, the damper current is “primary winding N ₁ → series resonance capacitor C ₁ → (Resonance capacitor C
₂ / parallel connection of high-speed recovery type diode D ₂ ) → smoothing capacitor Ci → primary side ground → parallel resonant circuit (driving winding N _B2 , resonant capacitor C _D2 ) → switching element Q ₂
Of the base → the collector of the switching element Q ₂ → the resonance current detection winding N _D → the primary winding N ₁ ′. Therefore, the base current I _B has a positive level as shown in FIG. 10C, and the collector current I _C has a positive level.
A waveform flowing in the negative direction is obtained as shown in (b), and thus the switching element Q ₂ is turned on. During this period T2, the damper current flowing through the above path causes the potential of the negative side of the base of the switching element Q ₂ to cause the capacitor C to become negative.
_{B2 is} charged, and as a result, the base potential of the switching element Q ₂ (in this case V _BE ) becomes the drive voltage V _D of the drive winding N _B2 with respect to FIG.
The voltage shifts to the minus side by the voltage of the capacitor C _B2 (FIG. 10 (d)). After that, the damper current decreases to zero, but the switching element Q ₂ maintains the ON state because the accumulation time Ts for the damper current that has been flowing so far occurs.

In the next period T3, as can be seen from the fact that the collector current waveform of FIG. 10B changes to the positive level, the positive level base current I _B of FIG. 10C flows to the emitter. . In the next period T4, an operation of discharging the accumulated charge due to the current flowing into the base of the switching element Q ₂ in the periods T2 and T3 is obtained, whereby a current flows from the emitter to the base. Therefore, the base current I _B changes to a negative level. Then, in the period T5, the switching element Q
The level of the base current I _B of ₂ becomes zero, the switching element Q ₂ shifts to the OFF state, and the damper current generated by the ON operation of the previous switching element Q ₂ is changed to the switching element Q ₁ in the next period T6. Allowed to flow to the side. During this period T6, the clamp current does not flow in the loop through which the drive current on the switching element Q ₂ side flows,
Along with this, the base-emitter voltage V _BE decreases as the drive voltage V _D decreases to the negative side.
Then, in the next period T7, the absolute value of the level of the falling drive voltage V _D becomes [Zener voltage of Zener diode D _Z2 + Forward voltage V of clamp diode D _B2
When it becomes larger than _F ], the Zener diode D _Z2 and the clamp diode D _B2 become conductive, and a clamp current flows as shown in FIG. 10 (f). This clamp current is a parallel resonance circuit (drive winding N _B2 , resonance capacitor C _D2 ) → clamp diode D _B2 → zener diode D _Z1 → capacitor C _B2 → resistance R _B2 → parallel resonance circuit (drive winding N _B2 , resonance). It flows in the loop of the capacitor C _D2 ). After that, as the drive voltage V _D rises, the clamp current level becomes 0 in the period T8.

As can be seen from the description of the operation waveforms thus far, in the present embodiment, since the Zener diode D _Z2 is provided, the damper current does not flow in the clamp diode D _B2 unlike the circuit of FIG. In order to prevent the reverse recovery current from occurring, the current level flowing through the clamp diode D _B2 is set to 0 when the switching element is on. That is, in the period T3 subsequent to the waveform diagram of FIG. 10 is prior art, the base current I _B
Is no longer seen in the clamp diode D _B2 , but this prevents the amplitude of the drive voltage V _D from being reduced. Therefore, in this embodiment, the switching element is inserted in series between the base and the emitter. The clamp diode D _B2 and the zener diode D _{Z2 that} are connected are
It can be seen as determining the amplitude V ₁ of the drive voltage V _D. Also, the parallel resonance circuit (N _B1 / C _D1 parallel connection and N _B2 / C _D2 parallel connection) consisting of the drive winding and the resonance capacitor causes voltage resonance, so that the switching elements Q ₁ and Q ₂ are connected. Generating drive voltage.
In this embodiment, the Zener diode D _Z1 ,
For the Zener voltage of D _Z2 , for example, about 4 to 5 V can be selected.

[0035] Therefore, the amplitude V ₁ of the said drive voltage V _D in the present embodiment (FIG. 2 (e)), the amplitude V of the drive voltage V _D of FIG. 10 is an operation waveform of the prior art (e) _As can be seen by comparing with ₁ , it is set to obtain a larger amplitude. This is because the self-excited oscillation drive circuit according to the present embodiment uses the drive winding N _B2 / resonance capacitor C _D2.
This is because it is a parallel resonance circuit of (drive winding N _B1 / resonance capacitor C _D1 ). On the other hand, in the power factor correction improving converter unit 20 shown in FIG. 9, the drive winding _NB2 -resonant capacitor _CB2 (drive winding _NB1 -resonant capacitor _CB1 ).
It is a series resonance circuit.

Then, when the amplitude of the drive voltage V _D is large as described above, it acts to increase the level (I ₁ ) of the negative level base current I _{B in} the period T4 (FIG. 2 (c)). It will be. For example, even when compared with the level (I ₁ ) of the base current I _B shown in the period T14 of FIG. 10C, it is confirmed that the level is higher in the present embodiment. Since the minus-level base current I _{B in} the period T4 can be regarded as the extraction current of the charge accumulated in the switching element Q ₂ ,
In this embodiment, the time length of the period T4 can be shortened accordingly. That is, in the present embodiment, the ratio of the storage time Ts to one switching cycle can be reduced by setting the Zener voltages of the Zener diodes D _Z1 and D _Z2 , and the influence of the variation in the storage time Ts can be reduced accordingly. It becomes possible.

For example, if the storage time Ts of the switching element becomes long, the start of the operation as the period T2 in response to the drive voltage V _D is delayed, and the base potential rises near the end of the period T1 extended by that amount. Then, as a result of the switching element Q ₂ being turned on at this point, the switching elements Q ₁ and Q ₂ are more likely to be turned on / on at the same time. As described above, since the storage time Ts has a relatively large variation, in order to avoid the above phenomenon, the bipolar transistor Ts which is a switching element is selected. Further, it has been found that the accumulation time Ts is relatively large with respect to the ambient temperature. On the other hand, in the present embodiment, it is possible to reduce the influence of the variation in the accumulation time Ts, and therefore, even when the stability of the switching operation and the margin for the ambient temperature change as described above are obtained, It is possible to further widen the allowable range of variations in Ts. In the case of the present embodiment, it is not necessary to sort at least the switching elements by the storage time Ts.

Further, as in the prior art shown in FIG. 9, a positive level base current I _B branches and flows in the clamp diode D _B2 , which may cause the switching element to continuously cause intermittent oscillation at the time of startup. I know there is. As a countermeasure, drive current limiting resistors R _B1 , R
There is a method of increasing the drive current supplied to the base by setting the value of _B2 small, but in this case, the margin for preventing the ON / ON state of the switching elements Q ₁ and Q ₂ at high temperature is sacrificed. , The selection of the accumulation time Ts was required. On the other hand, in the present embodiment, as shown in FIG.
As shown in (f), since only the clamp current when the switching element is off is made to flow through the clamp diode D _B2 , the problem of intermittent oscillation at startup is solved, and this also eliminates the need for selection by the accumulation time Ts. Will be said.

Further, the power conversion efficiency is also improved in the present embodiment. This is because, as described above, the Zener diodes D _Z1 and D _Z2 are inserted to set the amplitude of the drive voltage V _D large, and the negative level of the base current I _B in the period T4 is increased to extract the electric charge from the base. It is possible to reduce the time required and suppress the power loss when the switching element is off. Further, in the circuit of FIG. 9, the plus level base current is also shunted to the clamp diode and the base current level is suppressed, so that the collector saturation voltage V _{CE (SAT)} rises and the power loss when the switching element is turned on exists. However, in the present embodiment, since the plus level base current is prevented from flowing through the clamp diode, the collector saturation voltage V
_{CE (SAT) is} also reduced and power loss is reduced accordingly. Along with this, it is possible to set a higher switching frequency while maintaining the power conversion efficiency equivalent to that of the circuit of the prior art, and to reduce the size of peripheral components such as the converter transformer CVT and the resonance capacitor by that amount. .

Further, since the on-time of the transistor changes due to variations in the storage time Ts of the switching transistor, temperature changes, etc., if the inductance of the drive winding is constant, the storage time Ts will increase. The switching frequency decreases and the output voltage changes so as to increase. In the case of the present embodiment, the influence of the accumulation time Ts can be seen to be small as described above, so that the fluctuation rate of the DC output voltage (rectified and smoothed voltage Ei) is suppressed accordingly. Become. Along with this, the change in the control state with respect to the change in the rectified and smoothed voltage Ei due to the temperature change also becomes small. Since the characteristic change with respect to the temperature change is stable in this way, it is possible to obtain a current resonance type converter suitable as a power supply circuit of an in-vehicle device used in an environment where the temperature changes drastically. Further, although the imbalance of the accumulation time Ts between the switching Q ₁ and Q ₂ affects, for example, the fluctuation rate of the DC output voltage,
The allowable range of this unbalance margin is also expanded in this embodiment.

Second Embodiment FIG. 3 is a circuit diagram showing the configuration of a switching power supply circuit having a power factor correction converter circuit according to a second embodiment of the present invention. Are denoted by the same reference numerals and description thereof will be omitted. The power factor correction converter unit 1 shown in this figure
1, the magnetic coupling transformer MCT is provided as a circuit system for improving the power factor. Magnetic coupling transformer MCT
Is formed by winding at least the primary winding N ₁ and the secondary winding N _{2 so as} to be magnetically tightly coupled by a ferrite core or the like. In this case, the switching element Q ₁ ,
Drive windings N _B1 and N _B2 for Q ₂ are wound, and in this embodiment, the magnetic coupling transformer MCT also functions as a converter drive transformer for driving the switching elements Q ₁ and Q _2. The composition is adopted.

The secondary winding N ₂ wound around the magnetic coupling transformer MCT is inserted in series between the fast recovery diode D ₂ and the positive terminal of the smoothing capacitor Ci in the positive output line of the bridge rectifier circuit D _1. There is. Further, resonant capacitor C ₂ in this case is provided in parallel with the secondary winding N _2, but is adapted to form a parallel resonant circuit with the inductance component of the secondary winding N _2, the The operation is the same as in the case of FIG. 1, and suppresses the rise of the rectified and smoothed voltage Ei under light load. In this case, the primary winding N
_One end of ₁ is connected to a switching output point via a series resonance capacitor, and the other end is grounded to a primary side ground to supply a switching output.

The power factor improving operation is provided when the magnetic coupling transformer MCT is provided. In this case, the switching element Q is connected to the primary winding N ₁ of the magnetic coupling transformer MCT.
The switching outputs of ₁ and Q ₂ are supplied to obtain the voltage of the switching cycle (switching voltage). The switching voltage obtained in the primary winding N ₁ is transferred to the secondary winding N ₂ via the magnetic coupling of the magnetic coupling transformer MCT.
Is transmitted to the bridge rectifier circuit D ₁ , and the switching voltage is fed back to the rectifier current path of the bridge rectifier circuit D ₁ . In this way, the fast recovery type diode D ₂ operates so as to switch the rectified current at the switching cycle due to the superposition of the switching voltage superposed on the rectified output voltage. The power factor is improved by the same action as.

Further, in the present embodiment, the switching element Q ₁ , the Zener diode D _Z1 and the clamp diode D _B1 inserted between the base and the emitter thereof, the switching element Q ₁ , the Zener diode D _Z2 and the clamp diode D 1. _It is configured as a package component unit 3 in which each element of _B2 is packaged. FIG. 4 is a front view showing an example of the appearance of the package component unit 3.
In this case, the main body has a size of 20 mm × 16.8 mm, and six pin terminals P1 to P6 are provided. FIG. 5 is a circuit diagram showing the internal circuit configuration of the package component unit 3, which includes [switching element Q ₁ , Zener diode D _Z1 , clamp diode D _B1 ] and [switching element Q ₂ , Zener diode D _Z2 , clamp diode D]. _B2 ] is the same as the connection configuration shown in FIG. Then, the pin terminal P1 is connected to the base of the switching element Q _1, the pin terminal P2 is connected to the collector, the pin terminal P3 is connected to the emitter. Similarly, the pin terminal P4 is connected to the base of the switching element Q _2, the pin terminal P5 is connected to the collector, the pin terminal P6 is connected to the emitter.

As in the present embodiment, two sets of switching elements Q ₁ and Q ₂ , clamp diodes D _B1 and D _B2 , and zener diodes D _Z1 and D _Z2 are formed as a single package part 3 by a discrete structure. By configuring, the degree of variation in the characteristics of each element is further reduced, and thereby the design efficiency and the circuit reliability such as the temperature compensation range and the power conversion efficiency are further improved. Further, the board mounting area for each of the above-mentioned elements is reduced, and only one heat radiation plate should be attached to the switching elements Q ₁ and Q ₂ , so that the converter circuit can be made smaller and lighter. become.

Third Embodiment FIG. 6 is a circuit diagram showing the configuration of a switching power supply circuit having a power factor correction converter circuit according to a third embodiment of the present invention. The same parts are designated by the same reference numerals and the description thereof will be omitted. The power factor correction converter circuits of the first and second embodiments described above are preferably applied to the single range condition of the AC input voltage AC100V system or AC200V system. In form,
It is configured by including a voltage doubler rectifier circuit that generates a rectified and smoothed voltage that is approximately twice the AC input voltage, and is suitable for application to the condition of a single range of AC100 system and a relatively heavy load.

The power factor correction rectifier circuit 12 corresponds to the power factor correction converter circuit of the present embodiment shown in this figure. In this power factor correction rectifier circuit 12, a filter choke coil L _N is inserted in series to the positive line of the commercial AC power supply AC, and the filter capacitor C _N is inserted in parallel to both pole lines of the commercial AC power supply AC as shown in the figure. Thus, a normal mode low-pass filter is formed. In this embodiment, as in the embodiment of FIG. 3, a configuration for improving the power factor including a magnetic coupling transformer MCT is adopted. In this case, however, the secondary winding N _{2 of the} magnetic coupling transformer MCT is used. Is a filter choke coil L _N and a rectifying diode D
It is inserted in series with respect to the connection point of ₁₁ and D ₁₂ .

The rectifier diodes D ₁₁ and D ₁₂ form a voltage doubler rectifier circuit, and the cathode of the rectifier diode D ₁₁ is a secondary winding N ₂ -filter choke with respect to the positive line of the commercial AC power supply AC. It is connected via a coil L _N and the anode is grounded to the primary side ground. The anode of the rectifying diode D ₁₂ is connected to the cathode of the rectifying diode D ₁₁ , and its cathode is connected to the positive terminal of the smoothing capacitor Ci _A. In this case, the rectifier diodes D ₁₁ , D
₁₂ is a high-speed recovery type in response to the flow of rectified current that is intermittent in the switching cycle.

In this case, the smoothing circuit provided for the power factor correction rectifier circuit 12 includes smoothing capacitors Ci _A and Ci _B.
Are connected in series, and are inserted between the rectifying and smoothing line and the primary side ground. The series connection point of the smoothing capacitor Ci _A - Ci _B is connected to the negative line of the AC voltage AC.

In the power factor correction rectifier circuit 12 of the present embodiment, the voltage doubler rectifier circuit is formed by the above-mentioned connection form, and the voltage doubler rectifier operation is as follows.
For example, during a period in which the AC input voltage V _AC supplied to the commercial power supply _AC is positive, the rectified current is “commercial AC power supply AC → common mode choke coil CMC winding Na → filter choke coil L _N → secondary winding N. ₂ → Rectifier diode D ₁₂ →
Smoothing capacitor Ci _A → common mode choke coil C
MC winding Nb → commercial AC power supply AC ”, and smoothing capacitor C is generated by the rectified output of rectifier diode D _12.
The operation is to charge i _A. Therefore, the smoothing capacitor C
As shown in the figure, a rectified and smoothed voltage of Ei level corresponding to the AC input voltage level is obtained across i _A.

On the other hand, while the AC input voltage V _AC is negative, the rectified current is "commercial AC power supply AC → winding Nb → smoothing capacitor Ci _B → rectifying diode D ₁₁ → secondary winding N ₂ → filter choke coil L. _N → winding Na → commercial AC power supply AC ”
And the smoothing capacitor Ci _B is charged with the rectified output of the rectifying diode D ₁₁ , and a rectified and smoothed voltage of Ei level is obtained at both ends of the smoothing capacitor Ci _B. As a result of performing the charging operation on the smoothing capacitors Ci _A and Ci _B for the positive period and the negative period, respectively, the total rectified smoothed voltage obtained across the smoothing capacitors Ci _A -Ci _B connected in series is 2Ei. Therefore, when the _AC input voltage VAC is AC100 system, the voltage doubler rectification operation is performed in which a rectified and smoothed voltage of 200V system, which is almost twice the peak level, is obtained. Therefore, in the case of the present embodiment, the switching elements Q ₁ and Q ₂ perform the switching operation using the 200 V rectified and smoothed voltage as the input power source.

In this case, a switching voltage is generated in the secondary winding N ₂ of the magnetic coupling transformer by the operation similar to that described with reference to FIG. 2, whereby the rectifying diodes D ₁₁ and D ₁₂ are generated. The switching voltage is superposed on the rectification current path of the rectification current path, and by the superposition of the switching voltage, the rectification diodes D ₁₁ and D ₁₂ which are high-speed recovery type diodes can operate to intermittently rectify the rectification current in the switching cycle. Will be done. By this operation, in the power factor correction rectifier circuit 12, the smoothing capacitors Ci _A and Ci _B are charged in a state where the switching output is superimposed on the rectified output voltage, and the AC input voltage is positive / negative during each period. Thus, the voltage across the smoothing capacitors Ci _A and Ci _B is lowered in the switching cycle by the superposition of the switching voltage. Therefore, the charging current is made to flow to the smoothing capacitors Ci _A and Ci _B even during the period when the rectified output voltage level is lower than the voltage across the smoothing capacitor. As a result, the average waveform of the AC input current approaches the waveform of the _AC input voltage VAC, the conduction angle of the AC input current is expanded, and the power factor is improved.

As described above, the present embodiment is provided with the voltage doubler rectifier circuit so that a suitable power factor correction converter circuit can be obtained by applying it at the time of a heavy load of AC100V system, and then, in each of the above embodiments. It is made to have the same effect as the form.

Fourth Embodiment The circuit diagram of FIG. 7 shows the configuration of a switching power supply circuit having a power factor correction converter circuit according to a fourth embodiment of the present invention. The same parts as those in FIGS. 1, 3 and 6 are designated by the same reference numerals and the description thereof will be omitted. The power factor correction rectifier circuit 13 shown in this figure includes the voltage doubler rectifier circuit system shown in FIG.
Similarly, it comprises a converter transformer CVT having wound a primary winding N _1, method to be applied to the rectified current path of the switching output obtained at the primary winding N ₁ by capacitive coupling of the series resonance capacitor and Is taken. In this case, the primary winding N ₁ of the converter transformer CVT is connected to the switching output point via the series resonance capacitor C ₁ , and the other end is connected to the connection point of the rectifying diodes D ₁₁ and D ₁₂ for switching. It is configured to return the voltage to the rectified current path. As a result, the rectifying diodes D ₁₁ and D ₁₂ are prompted to intermittently switch the rectifying current in the switching cycle based on the switching output superimposed on the rectifying current path, and thereafter, the same operation as described with reference to FIG. 6 is performed. The rate will be improved. In addition, in the present embodiment, as in the embodiment of FIG. 1, a configuration is provided that includes a control transformer PRT in which the control winding N _C is wound around the drive windings N _B1 and N _B2 . The power factor and the rectified smoothed voltage are stabilized against AC input voltage and load fluctuation. Note that, also in the present embodiment, the switching elements Q ₁ , Q ₂ , the clamp diodes D _Z1 , D _Z2 ,
The Zener diodes D _B1 and D _B2 can be a component of one package.

Fifth Embodiment FIG. 8 is a circuit diagram showing a configuration example of a power supply circuit including a power factor correction converter circuit according to a fifth embodiment of the present invention, and FIG. The same parts as those in FIGS. 3, 6 and 7 are designated by the same reference numerals and the description thereof will be omitted. The power factor correction converter unit 14 shown in this figure is provided with _four switching elements Q ₁ , Q ₂ , Q ₃ and Q _{4, which} are configured by full bridge coupling.

In this case, a full bridge coupling is formed by providing a pair of switching elements Q ₁ and Q _{2 and} a pair of switching elements Q ₃ and Q ₄ which are respectively half-bridge coupled as shown in the figure. . In addition, each element (N is provided as a drive circuit system of the switching elements Q ₃ and Q ₄ ).
_B3 , D _Z3 , D _B3 , C _D3 , R _S3 , C _B3 , C _C3 and N _B4 , D
The connection form of _Z4 , D _B4 , C _D4 , R _S4 , C _B4 , C _C4 ) is the same as that of the elements of the drive circuit system provided in the switching elements Q ₁ and Q ₂ .

In the present embodiment, the package component section 3 including the switching elements Q ₁ and Q ₂ , the clamp diodes D _Z1 and D _Z2 , and the zener diodes D _B1 and D _B2.
Then, two package parts, that is, the package part ₃ including the switching elements Q ₃ , Q ₄ , the clamp diodes D _Z3 , D _Z4 , and the Zener diodes D _B3 , D _B4 are used. In the case of such a full bridge coupling, for example, as a series resonance circuit, one end of the primary winding N ₁ of the magnetic coupling transformer MCT is connected to the switching element Q.
₁ and Q ₂ are connected to the source-drain connection point, and the other end is connected to the source-drain connection point of the switching elements Q ₃ and Q ₄ via a series resonance capacitor C ₁ to provide a switching element. The switching outputs of Q ₁ , Q ₂ , Q ₃ and Q ₄ are supplied.

As a switching operation in the case of the full bridge coupling type as described above, a group of switching elements Q ₁ and Q _{4 and} a group of switching elements Q ₂ and Q ₃ are alternately turned on / off. The switching operation is performed. Therefore, the operation waveform obtained by the switching operation of the switching element Q ₃ is the same as that of the switching element Q ₂ shown in FIG. Further, the operation waveforms obtained by the switching operation of the switching elements Q ₁ and Q ₄ are the same operation waveforms as those in FIG. 2, and the timing is shifted by ½ of the switching cycle.

As the power factor improving operation of this embodiment, the full bridge-coupled switching element Q is used.
Switching output of _{1 to} Q ₄ is magnetic coupling transformer MC
By being supplied to the primary winding N ₁ of T, the operation thereafter will be performed in the same manner as described with reference to FIG.

In this case, the control circuit 2 is provided. The control circuit 2 is, for example, a block diagram of an amplifier circuit including the transistor Q ₁₀ shown in FIG. A direct current corresponding to the fluctuation of the rectified and smoothed voltage is supplied as a control current to the control winding N _C wound around the transformer PRT. By this control current, the switching frequencies of the switching elements Q _{1 to} Q ₄ are changed to stabilize the power factor and the rectified and smoothed voltage with respect to the AC input voltage and the load fluctuation as in the embodiment of FIG. 1. Become.

In the case of the configuration of this embodiment, the same effects as those of the above-described embodiments can be obtained, and by providing the voltage doubler rectification circuit system and adopting the full bridge coupling system. For example, a power factor correction converter circuit corresponding to a heavy load condition can be obtained with an AC input voltage AC100V system.

By the way, the respective embodiments described so far are the power factor correction converter circuits which can be added to the existing switching power supply circuit in order to improve the power factor. The improved converter circuit can of course be used as a single switching power supply circuit itself. For example, if a switching power supply circuit is specifically configured based on the configuration of the power factor correction converter unit 10 illustrated in FIG. 1, although not illustrated, for example, instead of a converter transformer, an insulated converter transformer PIT (Power Isolation Transformer) to provided a primary winding N ₁ is the primary winding, the leakage inductance by winding a separately arranged to loosely-coupled secondary winding for obtaining an alternating voltage on the secondary side Configure. Then, a rectifying and smoothing circuit may be provided on the secondary side so that a secondary DC voltage is obtained by rectifying the alternating voltage obtained on the secondary winding of the insulating converter transformer PIT.

The combination mode of the half-bridge / full-bridge coupling system, the rectification system, and the power factor correction system as the power factor correction converter according to the present invention is not limited to the above embodiments. The power factor correction converter circuit according to the present invention can be configured by combinations other than those shown in FIGS. 1, 3, and 6 to 8. Further, as the switching power supply unit 1 shown in each of the above-mentioned embodiments, in addition to a flyback converter or a forward converter by a PWM system, an RCC
Needless to say, various types of switching converters based on other methods such as (ringing choke converter) may be used, but a switching converter in which the current waveform or voltage waveform of the switching operation is a rectangular wave is connected. It is suitable when applied. Further, the applicant has previously proposed various switching power supply circuits for improving the power factor by feeding back the switching output of the current resonance type converter to the rectified current path, in addition to the circuit configurations shown in the above embodiments. But,
It is also possible to apply the configurations of these inventions as the configuration of the converter circuit of the present invention.

[0064]

As described above, according to the present invention, a clamp diode and a Zener diode are provided in series between a base and an emitter of a bipolar switching element, and a self-excited oscillation circuit for switching drive has a parallel resonance. With the simple configuration of providing the circuit, the influences of the storage time Ts of the switching element and the reverse recovery time Trr of the clamp diode during the switching operation, which have been problems so far, can be extremely reduced. This eliminates the need for part selection management due to variations in the accumulation time Ts, so that part management in the manufacturing process is simplified, and cost is reduced and manufacturing efficiency is improved. In addition, the accumulation time Ts and the reverse recovery time Tr
Since variations in the operating characteristics of the converter itself due to variations in r and the like are also stabilized, the reliability of the product is improved and the compensation range for temperature changes is expanded.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit according to a first embodiment of the present invention.

FIG. 2 is a waveform diagram showing the operation of the power factor correction converter circuit of the embodiment in a switching cycle.

FIG. 3 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit according to a second embodiment.

FIG. 4 is a front view showing an example of the external appearance of the package component part of the present embodiment.

FIG. 5 is a circuit diagram showing an internal configuration of a package component section.

FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit according to a third embodiment.

FIG. 7 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit according to a fourth embodiment.

FIG. 8 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit according to a fifth embodiment.

FIG. 9 is a circuit diagram showing a configuration of a switching power supply circuit including a power factor correction converter circuit as a prior art.

FIG. 10 is a waveform diagram showing the operation of the power factor correction converter circuit of the prior art in switching cycles.

[Explanation of symbols]

1 switching power supply section, 2 control circuit, 3 package parts section, ₁₀ , ₁₁ , ₁₄ power factor improving converter circuit, 12, 13 power factor improving rectifying circuit, D ₁ bridge rectifying circuit, D ₁₁ , D ₁₂ rectifying diode Ci, Ci _A ,
Ci _B smoothing capacitor, CVT converter transformer, CDT converter drive transformer, PRT control transformer, Q ₁ , Q ₂ , Q ₃ , Q ₄ switching element, N _{B1 to} N _B4 drive winding, Z _{D1 to} Z _D4 Zener diode, D _B1 ~ D _B4 Clamp diode, C _D1 ~ C
_D4 resonance capacitor, C ₁ series resonance capacitor, N ₁
Primary winding, MCT magnetic coupling transformer, N ₂ secondary winding, L _N filter choke coil, C _N filter capacitor, C ₂ resonance capacitor

Claims (19)

[Claims] 1. A bipolar transistor is used as a switching element, which is formed by a switching means having a self-excited oscillation drive circuit for driving the switching element, a series resonance capacitor and a series resonance winding, and a switching output of the switching means. As a current resonance type converter circuit having a supplied series resonance circuit, it is provided with a drive voltage level setting means provided so that the drive voltage generated by the self-excited oscillation drive circuit can be set to a required level. A current resonance type converter circuit characterized in that 2. The drive voltage level setting means is connected in series with a diode element for forming a path of a clamp current by being inserted between the base and the emitter of the switching element, and the drive voltage to be set. The Zener diode element having a Zener voltage corresponding to the level is provided, and the current resonance type converter circuit according to claim 1. 3. The current resonance type converter circuit according to claim 1, wherein the self-excited oscillation drive circuit is a parallel resonance circuit in which a drive winding and a resonance capacitor are connected in parallel. 4. A circuit unit including at least one or a plurality of the switching elements and the drive voltage level setting means provided for the switching elements is configured as one package component. The current resonance type converter circuit according to item 1. 5. A bipolar transistor is used as a switching element, is formed by a switching means having a self-excited oscillation drive circuit for driving the switching element, a series resonance capacitor and a series resonance winding, and a switching output of the switching means. A current resonance type converter provided with a series resonance circuit to which power is supplied, and a power factor improving means adapted to improve the power factor based on the switching output fed back from the series resonance circuit to the rectified current path. And a drive voltage level setting means provided so that the drive voltage generated by the self-excited oscillation drive circuit can be set to a required level. 6. The drive voltage level setting means is connected in series to a diode element for forming a clamp current path by being inserted between the base and emitter of the switching element, and the drive voltage to be set. The power factor correction converter circuit according to claim 5, further comprising a Zener diode element having a Zener voltage corresponding to the level. 7. The power factor correction converter circuit according to claim 5, wherein the self-excited oscillation drive circuit is a parallel resonance circuit in which a drive winding and a resonance capacitor are connected in parallel. 8. A circuit unit including at least one or a plurality of the switching elements and the drive voltage level setting means provided for the switching elements is configured as one package component. Item 5. A power factor correction converter circuit according to item 5. 9. The power factor improving means includes a normal mode low-pass filter including a filter choke and a filter capacitor provided for the output of the rectifying circuit, and a high-speed recovery type that is inserted in series in the rectifying current path of the rectifying circuit. The series resonant circuit includes a rectifying element and a converter transformer in which at least the series resonant winding is wound, and the series resonant circuit outputs a switching output obtained in the series resonant winding through capacitive coupling of the series resonant capacitor. 6. The power factor correction converter circuit according to claim 5, wherein the power factor correction converter circuit is provided so as to be applied to the rectified current path. 10. The power factor improving means is a normal mode low-pass filter including a filter choke and a filter capacitor provided for the output of the rectifier circuit, and a high-speed recovery type that is inserted in series in the rectifier current path of the rectifier circuit. And a magnetic coupling transformer formed by magnetically tightly coupling the rectifying element, at least the series resonance winding, and the superposition winding inserted in series in the rectification current path. The power factor correction converter circuit according to claim 5, which is characterized in that. 11. The apparatus according to claim 5, further comprising a power factor control means for controlling the power factor to be substantially constant with respect to a change in input AC input voltage level. The power factor correction converter circuit described. 12. Four switching elements are provided,
The power factor correction converter circuit according to claim 5, wherein these four switching elements are full-bridge coupled. 13. A voltage doubler rectifier circuit for generating a rectified and smoothed voltage corresponding to approximately twice the input AC input voltage level by rectifying a voltage of a commercial power source, and a bipolar transistor is used as a switching element. A series resonance, which is formed by a series resonance capacitor and a series resonance winding, and which is provided with a switching means for performing a switching operation by a self-excited system using a smoothing voltage as an operation power supply and a self-excited oscillation drive circuit, and a switching output of the switching means is supplied. A current resonant converter including a circuit; and a power factor improving means adapted to improve the power factor based on the switching output fed back from the series resonant circuit to the rectified current path. Drive provided so that the drive voltage generated by the excitation oscillation drive circuit can be set to the required level Power factor improving converter circuit, characterized in that it comprises pressure level setting means. 14. The driving voltage level setting means is connected in series to a diode element for forming a path of a clamp current by being inserted between a base and an emitter of the switching element, and a driving voltage to be set. The power factor correction converter circuit according to claim 13, further comprising a Zener diode element having a Zener voltage corresponding to the level. 15. The power factor correction converter circuit according to claim 13, wherein the self-excited oscillation drive circuit is a parallel resonance circuit in which a drive winding and a resonance capacitor are connected in parallel. 16. A circuit unit comprising at least one or a plurality of the switching elements and the drive voltage level setting means provided for the switching elements is one.
The power factor correction converter circuit according to claim 13, wherein the power factor correction converter circuit is configured as one package component. 17. The power factor improving means uses a high-speed recovery type rectifying element forming the voltage doubler rectifying circuit, and a normal mode low-pass filter including a filter choke and a filter capacitor provided for a commercial power source. And at least a magnetic coupling transformer formed by magnetically tightly coupling the series resonance winding and a superposition winding inserted in series in the rectified current path. The power factor correction converter circuit according to claim 13. 18. The power factor improving means uses a high-speed recovery type rectifying element forming the voltage doubler rectifying circuit, and a normal mode low-pass filter including a filter choke and a filter capacitor provided for a commercial power source. And a converter transformer around which the series resonance winding is wound, wherein the series resonance circuit rectifies a switching output obtained in the series resonance winding through a capacitive coupling of the series resonance capacitor. The power factor correction converter circuit according to claim 13, wherein the power factor correction converter circuit is provided so as to be applied to the path. 19. The apparatus according to claim 13, further comprising a power factor control means for controlling the power factor to be substantially constant with respect to a change in input AC input voltage level. The power factor correction converter circuit described.