JPH08256189A - Carrier reproducing system - Google Patents

Abstract

PURPOSE: To provide the carrier reproducing system which reproduces a carrier in a very small hardware scale. CONSTITUTION: A distance calculation circuit 4 which obtains a signal proportional to the magnitude of the phase error from two outputs of a complex multiplier 3 and a polarity discrimination circuit 9 which discriminates the direction of the phase error are provided. The polarity of the output of the distance calculation circuit 4 is controlled in a code circuit 10 connected to the succeeding stage of the distance calculation circuit 4, thereby detecting the phase error with the simple circuit.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION The present invention relates to, for example, multilevel QAM.
The present invention relates to a carrier wave reproducing method for detecting a phase error of a carrier wave reproduced when demodulating a digital modulation signal or the like of a (quadrature amplitude modulation) method.

[0002]

2. Description of the Related Art When demodulating a digitally modulated signal, a carrier wave synchronized with the modulated wave must be reproduced. This type of carrier recovery circuit is described in, for example, the Technical Report of the Television Society of Japan, "Study of QPSK demodulator for digital satellite broadcasting".
(August 1992, TV studies technical report VOL.16, NO.52pp19-2
An example is disclosed in 4). FIG. 10 is a block diagram showing a carrier recovery circuit using a conventional phase error detection method. In the figure, 1 and 2 are input terminals, 3 is a complex multiplier, 23 is a calculator for calculating an Arc Tan, 5 is a loop filter, 6 is a numerically controlled oscillator (hereinafter referred to as NCO), 7 and 8. Is an output terminal.

The demodulation operation of the carrier recovery circuit will be briefly described. One of two orthogonal signals obtained by quadrature detection of a QPSK (four-phase phase shift keying) signal at the modulation frequency and frequency conversion near the baseband. Is the i signal and the other is the q signal. Then, the i signal is input to the input terminal 1 and the q signal is input to the input terminal 2. The complex multiplier 3 performs complex multiplication of the i signal and the q signal input to one side and two signals supplied from the NCO 6 input to the other side and having phases different from each other by 90 degrees, and QP
The I signal and the Q signal, which are demodulated outputs of the SK signal, are output from the output terminals 7 and 8, respectively.

On the other hand, the demodulated output of the complex multiplier 3 is also supplied to the arithmetic unit 23, where the arc tangent (Arctan) is calculated from the two signals and the signal proportional to the phase error is output to the loop filter 5. . In loop filter 5,
A frequency component unnecessary for control is removed from the output of the arithmetic unit 23 to form a control signal for the NCO 6. NCO6
Supplies the phase-controlled signal to the complex multiplier 3 again based on this control signal.

By constructing such a phase control loop, the NCO 6 reproduces a new carrier wave in synchronism with the suppressed carrier wave modulation signal, and a demodulated signal having no phase error can be output to the output terminals 7 and 8. .

By the way, in the conventional method, when detecting the phase error between the reproduced carrier wave and the modulated signal, the phase error is Arctan with respect to the two output signals of the complex multiplier 3, that is, the I signal and the Q signal. The value of Arctan (Q / I) is obtained from the signal of the complex multiplier 3 by utilizing the fact that it is proportional to (Q / I). Generally, the calculation of Arctan (Q / I) cannot be easily realized, but the following two methods are usually used.

The first method is a look-up table method (hereinafter referred to as LUM method) using a ROM or the like.
In this method, the value of Tan ⁻¹ for the input signal is stored in advance as a table, the I signal and the Q signal of the complex multiplier 3 are referred to in the table, and the reference value is output.

The second method is the method disclosed in the above document, which simplifies the operation by approximation by Taylor expansion. That is, when the phase error is θ, θ = Arctan (Q / I) is Taylor-expanded so that the error is minimized in the vicinity of (Q / I) = 1, and the approximation up to the first order is taken. Then, θ = π / 4 + ((Q / I) −1) / 2. Therefore, this equation is constructed by hardware to obtain θ.

[0009]

However, the above-mentioned LUT
In order to realize the method, it is necessary to provide a large-capacity storage device, which causes a problem that the circuit scale becomes large. For example, if an 8-bit Arctan (Q / I) output is to be obtained for 8-bit I and Q signal inputs, 512
A storage means with a capacity of K bits must be prepared.
Of course, in order to further improve the accuracy, it is necessary to increase the number of bits of the input / output signal.

When the Taylor approximation method is used, since the above approximation formula includes a division term of (Q / I), the means for calculating the reciprocal of the I signal is multiplied by Q and 1 / I. Means and are needed. When these arithmetic means are assembled by hardware, there is a problem that the circuit scale becomes large.

Further, in the latter method, since an approximation formula developed by Taylor is used so as to minimize the error of the signal having the phase difference of 45 degrees with respect to the orthogonal axis as described above, the other formula is used. For a multilevel QAM signal including a phase signal, the cutoff frequency of the loop filter after detecting the phase error must be lowered. As a result, there arises a problem that the response of the loop is reduced.

The present invention has been made to solve the above problems, and a first object thereof is to provide a carrier wave reproducing system capable of realizing carrier wave reproduction with an extremely small hardware scale.

A second object of the present invention is to provide a carrier recovery system with improved output characteristics when the phase error is near zero.

Further, a third object of the present invention is to provide a carrier recovery system which can be applied to the detection of a wider range of frequency errors.

[0015]

According to a first aspect of the present invention, there is provided a carrier wave reproduction method, wherein complex multiplication is performed between an oscillating means for outputting two signals having different phases and a quadrature modulated signal and the two signals from the oscillating means. To detect a phase error between the quadrature modulation signal and the two signals from the oscillating means from the output of the complex multiplication means to demodulate the quadrature modulation signal in which the carrier wave is suppressed. In the carrier recovery method, the distance between the axis of the oscillating means, which is deviated by 45 degrees from the orthogonal axis of the two signals, and the output of the complex multiplying means is obtained as the phase error output.

According to a second aspect of the present invention, there is provided a carrier recovery system which comprises oscillating means for outputting two signals having different phases, and complex multiplying means for performing a complex multiplication between the quadrature modulated signal and the two signals from the oscillating means. In the carrier recovery method, the phase difference between the quadrature modulation signal and the two signals from the oscillating means is detected from the output of the complex multiplication means, and the quadrature modulation signal in which the carrier is suppressed is demodulated. It is characterized in that a distance between a point obtained by projecting the output of the complex multiplication means on an axis deviated from the orthogonal axis of the two signals of the oscillation means by 45 degrees and the origin of the orthogonal axis is obtained as the phase error output. .

According to a third aspect of the present invention, the carrier recovery system adds a constant to the phase error output according to the number of modulation levels of the input quadrature modulation signal, and corrects the phase error of the demodulated signal in the vicinity of zero phase error. It is characterized by performing.

According to a fourth aspect of the present invention, in the carrier recovery method, the phase error direction of the demodulated signal is obtained based on the asymmetry of the signal arrangement with respect to the axis deviated from the orthogonal axis of the two signals by 45 degrees. Is characterized by.

A carrier recovery system according to a fifth aspect is characterized in that the phase error outputs are time-averaged and the frequency error of the demodulated signal is obtained based on the temporal change.

[0020]

In the carrier recovery system according to the first aspect, two signals supplied from the oscillating means and having different phases by 90 degrees and the input signal represented by the complex number are complex-multiplied by the complex multiplying means, and the complex multiplication is performed. The two outputs are used to determine the distance between the signal point and the axis having a phase difference of 45 degrees with the orthogonal axis of the signal of the oscillating means, and output as a phase error signal.

In the carrier wave reproducing method according to the second aspect, two signals supplied from the oscillating means are different in phase by 90 degrees, and
The input signal expressed as a complex number is subjected to complex multiplication by the complex multiplication means, and the output of the complex multiplication means is used to obtain a phase difference of 45 degrees from the orthogonal axis of the signal of the oscillation means using the two outputs obtained by the complex multiplication. The distance between the projected point on the axis and the origin of the orthogonal axis is calculated and output as a phase error signal.

In the carrier recovery system according to the third aspect, a constant according to the modulation system can be added to the phase error output to improve the output characteristic in the vicinity of zero phase error.

In the carrier recovery system according to the fourth aspect, it is judged that the demodulated signal including the phase error is arranged asymmetrically with respect to the axis having a phase difference of 45 degrees from the orthogonal axis, and the phase error is detected. The polarity of the output of the means can be changed to detect the phase error direction.

In the carrier recovery system according to the fifth aspect, the frequency error signal can be output from the output of the phase error detecting means.

[0025]

Embodiments of the present invention will be described below with reference to the accompanying drawings. Example 1

FIG. 1 is a block diagram showing a carrier recovery circuit according to a first embodiment of the present invention. In the figure, reference numerals 1 to 3 and 5 to 8 indicate the same components as the carrier recovery circuit (FIG. 10) described as the prior art. That is, as the digital multilevel QAM signal, the i signal and the q signal are input to the input terminals 1 and 2, respectively, and the complex multiplier 3 outputs the I signal and the Q signal which are demodulation outputs. Here, the i signal and the q signal are obtained by performing quadrature detection of the multilevel QAM signal at the modulation frequency and performing frequency conversion near the base band. In the complex multiplier 3, in addition to the i signal and the q signal, the NCO
Two signals having a phase difference of 90 degrees are input from 6 and are subjected to complex multiplication to obtain a demodulation output.

The demodulated I and Q signals are output to the output terminals 7 and 8 and at the same time supplied to the distance calculating circuit 4 and the polarity determining circuit 5, respectively. The distance calculation circuit 4 calculates the distance proportional to the phase error from the output signal of the complex multiplier 3, and the polarity determination circuit 5 calculates, for example, DS.
Like the well-known Costas loop used in the carrier recovery circuit for the B-SC modulated wave, the phase error direction is determined using the first symbol and the error symbol.

The output of the distance calculating circuit 4 is the polarity determining circuit 9
The encoding circuit 1 based on the error direction determined by
A code is added at 0 and the phase error signal is supplied to the loop filter 5.

The loop filter 5 removes a frequency component unnecessary for control from the phase error signal input thereto,
Get control signal. The NCO 6 is connected to the complex multiplier 3 again to supply the two signals whose phases are controlled based on the control signal from the loop filter 5.

By constructing such a phase-locked loop, the NCO 6 reproduces a new carrier wave synchronized with the modulated signal in which the carrier wave is suppressed, and as a result, a demodulated signal having no phase error is outputted from the output terminals 7 and 8. Can be output from.

FIG. 2 is a block diagram showing the distance calculating circuit 4 used in the carrier wave reproducing circuit of FIG. See also FIG.
FIG. 3 is a signal point arrangement diagram showing a signal point arrangement of a 16-value QAM signal in the distance calculation circuit 4 of FIG. First, NCO6
The case where the two signals supplied to the complex multiplier 3 from 1 have the phase error θ shown in FIG. 3 will be described.

In FIG. 3, if the phase error is zero,
It is assumed that the coordinate value of the point P in the first quadrant should originally be (A, B), but the coordinate value becomes (X, Y) because the phase error θ remains. Where the orthogonal axis X-Y
When the distance between the axis (the axis shown by the broken line in the figure) and 45 degrees counterclockwise from the point P is α, this distance α is α = | X−Y | / √2. expressed.

On the other hand, the coordinate value (X, Y) of the point P is represented by X = Acosθ + Bsinθ Y = -Asinθ + Bcosθ using the original coordinate values (A, B) and θ, respectively. Therefore, α is α = | γ · cos (θ− (π / 4 + φ)) | (where φ = Arctan (B / A)) using the distance γ of the point P from the origin O. Can be represented. This is because the distance α is a function of the phase error θ,
Moreover, it is shown that the distance can be derived by an arithmetic expression consisting of an extremely simple proportional function of | X−Y |.

Therefore, if the distance calculating circuit 4 is configured by the subtracting circuit 11 and the absolute value circuit 12 as shown in FIG. 2, each of the distance calculating circuits 4 is calculated based on the I signal and the Q signal which are the demodulated outputs from the complex multiplier 3. The distance α between the signal point and the axis deviated from the orthogonal axis by 45 degrees can be calculated, and as a result, a signal proportional to the phase error can be output from the distance calculation circuit 4.

FIG. 4 shows the output characteristic of the phase error signal. In the carrier recovery method described above, a known constant corresponding to the modulation level number of the multilevel QAM signal is output even if the phase error θ is zero. Then, due to the influence of this output, a discontinuity occurs in the output characteristic even if the phase error is zero. Such phase jitter that occurs near the error zero when detecting the phase error is inconvenient for handling the phase error.
This phase jitter will be discussed later in Example 3.

Embodiment 2 In Embodiment 2, the distance between the point projected on the axis having a phase difference of 45 degrees with the orthogonal axis of the signal of the oscillating means and the origin of this orthogonal axis is obtained and output as a phase error signal. Figure 5
3 shows a distance calculation circuit 4 for that purpose.

As in the case of the first embodiment described above,
The two signals supplied from the NCO 6 to the complex multiplier 3 are
When the phase error θ shown in FIG.
The distance between the point projected on the axis (the axis indicated by the broken line in the figure) shifted 45 degrees counterclockwise from Y and the origin O of the orthogonal axis is β.
Then, this distance β is represented by β = | X + Y | / √2.

On the other hand, the coordinate values (X, Y) of the point P are represented by X = Acosθ + Bsinθ Y = -Asinθ + Bcosθ, respectively, as in the first embodiment. Therefore, β is β = | γ · −sin (θ− (π / 4 + φ)) | (where φ = Arctan (B / A)) using the distance γ of the point P from the origin O. Can be expressed as This shows that the distance β is also a function of the phase error θ like the distance α, and that the distance can be derived from an extremely simple arithmetic expression of | X + Y |.

Therefore, even when the distance calculating circuit 4 is configured by the adding circuit 13 and the absolute value circuit 12 as shown in FIG. 5, the I signal which is the demodulated output from the complex multiplier 3,
Based on the Q signal, it is possible to calculate the distance β between a point obtained by projecting each signal point on the axis deviated from the orthogonal axis by 45 degrees and the origin O of the orthogonal axis, and as a result, a signal proportional to the phase error is obtained. It can be output from this distance calculation circuit 4.

Third Embodiment In the third embodiment, a constant adder circuit 14 for adding a constant is provided after the distance calculating circuit 4 in the above-described first and second embodiments, and the output near the zero phase error of the demodulated signal is provided. It is intended to improve the characteristics.

FIG. 6 is a block diagram showing the constant addition circuit 14 connected to the distance calculation circuit 4. In the figure, 1
Reference numeral 5 is a constant circuit, and 16 is an adder. Constant addition circuit 14
Then, the constant circuit 15 generates a constant according to the multi-valued number of the QAM signal, and the adder 15 adds the constant to the output of the distance calculation circuit 4 to output the distance α ₀ . That is, QAM
The modulated quadrature modulated signal is a four-valued QAM (QPSK), 6
Appropriate constants are added according to the number of modulation levels such as 4-level QAM and 256-level QAM, and output.

In this way, by adding a constant corresponding to the number of modulation levels of the QAM signal, the output of the distance calculation circuit 4 at the zero phase error can be canceled, and the phase error θ can be reduced as shown in FIG. 4 (b). If it is zero, its output α
_{0 is} also adjusted to zero. As a result, the entire phase error detection range can be handled linearly, and the phase jitter that has occurred near error zero at the time of phase error detection can be suppressed.

Fourth Embodiment In the carrier wave reproducing method of the first and second embodiments, the distance calculating circuit 4 calculates a distance proportional to the magnitude of the phase error and outputs a demodulated signal having no phase error. However, the distance calculation circuit alone cannot determine whether the reproduced carrier wave is ahead of or behind the input modulation signal. Therefore, it is necessary to determine the error direction of whether the reproduced carrier wave is ahead of or behind the input modulated wave by some method. In the first embodiment, as described above, the well-known Costas loop is used to determine the phase error direction. In the fourth embodiment, in the determination of the phase error direction, by utilizing the asymmetry of the data arrangement with respect to the axis deviated by 45 degrees from the orthogonal axis, the output of the distance calculation circuit is used, and the phase error of the phase error can be determined with a simpler configuration. A phase error detection circuit including a circuit for determining the direction will be described.

FIG. 7 is a block diagram showing a polarity judgment circuit 9 according to the fourth embodiment of the present invention. 8 is a signal point arrangement diagram showing the signal point arrangement of the 16-value QAM signal in the polarity determination circuit 9 of FIG. In FIG. 7, 17 is an integrating circuit and 18 is a judging circuit. In the polarity determination circuit 9, first, in the integration circuit 17, the output (I
The sign of -Q) is integrated for a certain period to obtain the expected value of the sign of (I-Q). This expected value increases or decreases in accordance with the difference in appearance probability between the signal point with I> Q and the signal point with I <Q, that is, the phase error direction. Therefore, the determination circuit 18
Then, the phase error direction is determined based on this expected value, and the polarity signal P / N is output.

The determination of the phase error direction will be further described with reference to the signal point arrangement diagram of FIG. When the four signal points P1 to P4 of the 16QAM signal are arranged as shown in this figure and the phase error is zero, these signal point arrangements are arranged on an axis (straight line I = 45) which is deviated from the orthogonal axis by 45 degrees. Be symmetrical about Q). Therefore, if the appearance probabilities of the respective signals are equal, when comparing the I coordinate value and the Q coordinate value of the appearing signal point, the probability that the signal point with I> Q and the signal point with I <Q are equal Should appear in.

However, if two orthogonal signals supplied from the NCO 6 have a phase error θ, the signal point P
The arrangement of 1 to P4 is asymmetric with respect to the straight line I = Q. Therefore, when the I coordinate value and the Q coordinate value of each signal point are compared, there is a difference in the appearance probability between the signal point with I> Q and the signal point with I <Q. For example, in FIG. 8, points P2 and P4 where I = Q when the phase error is zero.
Becomes I> Q when the phase error θ exists. Then, the signal point with I <Q is only P1, but I> Q
There are three signal points P2, P3, and P4. Therefore, assuming that the appearance probabilities of the signal points P1 to P4 are equal,
A signal point with I> Q appears three times more frequently than a signal point with I <Q. The difference in the appearance probabilities can be used to determine the phase error direction.

Fifth Embodiment In the fifth embodiment, the carrier recovery system of each of the embodiments described so far is applied to frequency error detection. FIG. 9 is a block diagram showing an example of the frequency error detection circuit.
Is an averaging circuit, 20 is a delay device, and 21 is a subtractor. Reference numeral 22 is a phase error detection circuit that outputs the phase error signal α (or β or α ₀ ) in the first to fourth embodiments.

The phase error signal obtained by the phase error detection circuit 22 is time averaged by the averaging circuit 19. The output of this averaging circuit 19 is delayed by a delay device 20,
Further, the subtracter 21 can obtain the time derivative of the average value by taking the difference between the two signals and output it as a signal proportional to the frequency error. By reproducing the carrier wave using such a frequency error signal, it is possible to reliably deal with a wide range of frequency errors that cannot be dealt with only by detecting the phase error.

[0049]

Since the present invention is constructed as described above, it has the following effects.

According to the carrier recovery system of the first aspect, the phase error signal can be detected with an extremely simple circuit configuration as compared with the conventional system.

According to the carrier recovery method of the second aspect, the phase error signal can be easily detected by the distance calculating circuit including the adding circuit and the absolute value circuit.

According to the carrier wave reproducing method described in claim 3, by only adding a constant adding circuit according to the modulating method,
The output characteristic near zero phase error can be improved.

According to the carrier recovery method of the fourth aspect, the phase error signal is detected by the polarity determination circuit utilizing the fact that the demodulated signal including the phase error has an asymmetric data arrangement with respect to the axis deviated from the orthogonal axis by 45 degrees. The polarity of
Moreover, the polarity determining circuit can be realized with a simple configuration including the integrating circuit and the determining circuit.

According to the carrier recovery method of the fifth aspect, it can be applied to the detection of frequency error in a wider range.

[Brief description of drawings]

FIG. 1 is a block diagram showing a carrier recovery circuit according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a distance calculation circuit of FIG.

FIG. 3 is a signal point arrangement diagram showing a signal point arrangement of 16QAM signals in the distance calculation circuit of FIG.

FIG. 4 is a diagram illustrating output characteristics of a phase error signal.

FIG. 5 is a block diagram showing a distance calculation circuit according to a second embodiment of the present invention.

FIG. 6 is a block diagram showing a constant adder circuit according to a third embodiment of the present invention.

FIG. 7 is a block diagram showing a polarity determination circuit according to a fourth embodiment of the present invention.

8 is a signal point arrangement diagram showing a signal point arrangement of a 16QAM signal in the polarity determination circuit of FIG.

FIG. 9 is a block diagram showing a frequency error detection circuit according to a fifth embodiment of the present invention.

FIG. 10 is a block diagram showing a carrier recovery circuit using a conventional phase error detection method.

[Explanation of symbols]

1, 2 input terminals, 3 complex multiplier, 4 distance calculation circuit, 5 loop filter, 6 numerically controlled oscillator, 7, 8
Output terminal, 9 polarity determination circuit, 10 code circuit.

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Taku Fujiwara, No. 1 Baba Institute, Nagaokakyo, Kyoto Prefecture In the Video System Development Laboratory, Mitsubishi Electric Corp. (72) Inventor Jun I, No. 1 Baba Institute, Nagaokakyo, Kyoto Prefecture Mitsubishi Electric Video System Development Laboratory Co., Ltd.

Claims (5)

[Claims] 1. An oscillating means for outputting two signals having different phases, and a complex multiplying means for performing a complex multiplication between a quadrature modulated signal and the two signals from the oscillating means, the output of the complex multiplying means. From the quadrature modulation signal and the two signals from the oscillating means are detected to demodulate the quadrature modulation signal in which the carrier is suppressed. A carrier recovery system characterized in that a distance between an axis deviated by 45 degrees from an orthogonal axis and an output of the complex multiplication means is obtained as the phase error output. 2. The output of the complex multiplying means comprises: an oscillating means for outputting two signals having different phases; and a complex multiplying means for performing a complex multiplication between the quadrature modulated signal and the two signals from the oscillating means. In the carrier recovery system, in which the phase error between the quadrature modulated signal and the two signals from the oscillating means is detected to demodulate the quadrature modulated signal in which the carrier is suppressed, the output of the complex multiplying means is A carrier wave reproducing system characterized in that a distance between a point projected on an axis deviated by 45 degrees from an orthogonal axis of two signals of an oscillator and an origin of the orthogonal axis is obtained as the phase error output. 3. The phase error output is corrected by adding a constant in accordance with the number of modulation levels of the input quadrature modulation signal to correct the phase error of the demodulated signal in the vicinity of zero phase error. Item 3. The carrier recovery system according to Item 1 or 2. 4. The phase error direction of the demodulated signal is obtained based on the asymmetry of the signal arrangement with respect to the axis deviated by 45 degrees from the orthogonal axis of the two signals of the oscillating means. The carrier wave reproduction method described in 2. 5. The carrier recovery system according to claim 1, wherein the phase error output is averaged over time, and the frequency error of the demodulated signal is obtained based on the temporal change.