JPH06269189A - Current controller for pwm inverter - Google Patents

Abstract

PURPOSE:To increase current control operating speed by employing a speed type current control operating section and a model voltage operating section and performing two axes/three-phase coordinate conversion by adding an output from a first coordinate converting section for determining a previous value. CONSTITUTION:Current control operating sections 3A, 4A are configured into speed type in place of conventional position type. Similarly, a model voltage operating section 5A is configured to determine the difference and add the difference to a speed type PI operating section. A switch circuit 12 is provided at the output stage in order to bring the difference of feed forward voltage forcibly to zero at the time of starting an induction motor 1. Output voltages V1d-oi, V1q-pi from the current control operating sections 3A, 4A are added to outputs from the model voltage operating section 5A and a first coordinate converting section 13, respectively. Addition results are passed through a second coordinate converting section 14 for converting two axes of a rotary coordinate into three-phase PWM voltage component on a fixed coordinate using a circle tracking method and converted into a three-phase voltage control signal for a PWM inverter 11.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a PWM inverter current control device having a digital control current control system.

[0002]

2. Description of the Related Art In many cases, a PWM voltage type inverter is adopted for variable speed driving of an induction motor, a vector control system is adopted for a control system, and a current control system is provided for high speed response.

Further, due to the high performance of CPUs and DSPs (digital signal processors), it has become possible to execute a current control system conventionally composed of analog circuits by digital operation.

Further, an IGBT or the like has been put into practical use for the main circuit switch of the inverter, which can increase the switching frequency as compared with the conventional bipolar transistor, and the current response is further improved by speeding up the current control system.

However, in recent years, not only high speed but also improvement of inverter output voltage and current accuracy and shortening of calculation time have been demanded by applying magnetic flux observer and parameter fluctuation compensation.

As a solution to these problems, the inventors of the present application have described, "A digital current control method for an induction motor by PWM synchronous current sampling, IEEJ Transactions D, 112.
Vol. 7, 1992, pp. 613-622 "has already been proposed.

This current control method has a simplified block configuration shown in FIG. Three-phase current detection signal in the fixed coordinate system of the induction motor 1 with respect to the biaxial components of the exciting current command I _d * of the induction motor 1 of the rotating coordinate system and the torque current command I _q * orthogonal thereto by vector non-interference calculation. The corresponding currents i _d and i _q are obtained by the coordinate conversion circuit 2 that performs coordinate conversion from i _u , _iv , and i _w into fixed three-phase-rotation two-axis currents I _b and I _q , and the deviation between these current commands and detected current Are position type current control calculation units 3 and 4, respectively.
Is calculated by proportional integral (PI).

On the other hand, the ideal voltage when the vector control condition is satisfied is obtained from the current commands I _d *, I _q * as a feedforward term in the model voltage calculation unit 5. The voltages v _1d-m and v _{1q-m that} are the result of this calculation are obtained by adding the outputs v _1d-pi and v _1q-pi from the voltage control calculation units 3 and 4, and A current control output that suppresses error current components due to disturbances and motor constant errors is obtained.

This current control output is converted into a voltage command whose saturation is suppressed by limiter circuits 6 and 7, respectively, and is converted from biaxial rotation coordinates to rotation polar coordinates by a coordinate conversion circuit 8. The magnitude R and the phase φ _i are coordinate converted. The circuit 9 converts the polar coordinates into fixed three-phase components U, V, and W-phase voltages by PWM calculation by the circular locus method, and the limiter circuit 1 respectively.
PWM voltage source inverter 11 through 0 _U , 10 _V , 10 _W
To generate the primary voltage of the induction motor 1.

[0010]

The conventional method has the following four characteristics.

(1) The current control calculation units 3 and 4 are constructed in position type.

(2) The output of the model voltage calculator 5 is added to the outputs of the current control calculators 3 and 4 as feedforward.

(3) A limiter is applied to the added value of the model voltage and current control calculation unit in the rotating coordinate system.

(4) PWM calculation is performed by the circle locus method.

Regarding the above characteristic points, there are the following problems.

First, regarding the item (2), when the magnetic flux and the exciting current command are suddenly changed from zero to a set value at the start of pre-excitation when the induction motor is rotating at a high speed, the magnetic flux of the electric motor is zero, which is the reverse. The electromotive voltage is also zero, whereas the feedforward voltage rapidly gives a voltage in the presence of magnetic flux.

As a result, an overcurrent is generated in the electric motor, which causes overcurrent stoppage or failure. If the exciting current command is gradually increased from zero with a cushion characteristic as a countermeasure against this, the feedforward voltage is also delayed by the cushion time, and the establishment of the magnetic flux is delayed.

Next, regarding the item (3), 2 on the rotational coordinate
When limiters are applied to the axis components by limiter circuits 6 and 7, as shown in FIG. 6, when each axis is saturated, the limit widths V _dL and V _qL are 2 ^1/2 times the two-axis composite value, and the desired value is obtained. No limit value can be obtained. Actually, there is an output limit for three-phase voltages of U, V, and W phases, which is different from the range shown in FIG.

As a countermeasure against this, there is a case where the biaxial component is converted into polar coordinates as shown by a broken line in FIG. 5 and then the amplitude value is limited and approximated by a circular saturation limiter. In this case, the number of calculations is increased by the polar coordinate conversion.

Next, with respect to the item (4), the circular locus method is generally calculated by using commands of voltage amplitude and phase. For this reason,
It is necessary to temporarily convert the two-axis components into polar coordinates, which is one of the causes of lowering the calculation speed.

An object of the present invention is to provide a current control device which solves the above problems.

[0022]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention relates to a speed with respect to a deviation between an excitation current command and a torque current command of a rotating coordinate system of a vector-controlled PWM inverter and respective detected values. A current control calculation unit (3A, 4A) for performing a proportional proportional integration calculation, and a speed type model voltage calculation unit 5A for obtaining an ideal voltage when a vector control condition is satisfied as a feedforward term from the excitation current command and the torque current command. A first coordinate conversion unit (13) for inversely converting a three-phase voltage command of the PWM inverter into biaxial components of an exciting current component and a torque current component of a rotating coordinate system, the coordinate transforming unit, and the model voltage computing unit. And a value obtained by adding the corresponding two-axis components of the current control calculation unit to a three-phase component of fixed coordinates,
A second coordinate conversion unit (14) having limiter means for giving a three-phase voltage command of the PWM inverter.

Further, according to the present invention, the model voltage calculation unit is a PW.
The present invention is characterized in that the induction motor serving as a load of the M inverter is provided with a switch means for forcing the output to zero for a time corresponding to the secondary time constant of the induction motor.

[0024]

In the current control calculation and the model voltage calculation, the speed type (differential type) is used instead of the position type to increase the current control calculation speed and enhance the current responsiveness. When the current control method of the 1-input 1-output system described below is configured by the discrete value system, the position type is as shown in FIG. 1A, and the velocity type is as shown in FIG. 1B. become. The output v _k of these current control calculation units is given by the following equation. Equation (1) is a position type, and equation (2) is a velocity type.

[0025]

[Equation 1]

I _k *; current command (time k) i _k ; current detection value (time k) Z ^-1 ; 1 sample delay K _P ; proportional gain T _i ; integral time constant ΔT; calculation time (1) , As is clear from the equation (2), the integral term and the limiter are separated in the position type, and the integration progresses even at the output limit, so that it is necessary to add an operation such as further stopping the integration in FIG. 1 when the limiter operates. There is. On the other hand, in the velocity type, since the integral value and the limiter value are equal, no additional operation of the integral value is necessary, the number of calculations is small, and the response is quick even when returning from saturation.

By the current control calculation for the velocity type, the feedforward term is obtained by the velocity type calculation in the biaxial coordinates of the rotating coordinate system before the boundary A.

Next, the subsequent stage of the boundary A is calculated by three-phase coordinates in a fixed coordinate system. For this reason, the first coordinate conversion unit for obtaining the previous value at the boundary A and the second coordinate conversion unit for the two-axis / three-phase conversion
The coordinate conversion unit of is provided. Then, the limiter performs the three-phase voltage component, and the value after the limiter is inversely converted into two axes to be the limit value for the current control calculation so that the voltage saturation limiter and the output voltage match.

Further, according to the present invention, the model voltage calculation unit is provided with the switch means, so that the differential component of the feedforward voltage is temporarily set to zero at the start of pre-excitation, and the differential component of the feedforward voltage is set after the activation. Add before integrator. At the time of this addition, the model voltage component equivalent is included in the integral term of the current control calculation unit by feedback, and feedforward is added to this.

As a result, the feedforward voltage is applied without the delay due to the cushion characteristic while preventing the overcurrent at the time of starting.

[0031]

FIG. 2 is a block diagram showing an embodiment of the present invention. In the figure, the current control calculation units 3A and 4A are speed type instead of the conventional position type. Similarly, the model voltage calculation unit 5A is configured to obtain a difference and perform an output to be added to the speed type PI calculation unit. A switch circuit 12 is provided at the output stage of the model voltage calculation unit 5A and the difference in the feedforward voltage when the induction motor 1 is started. Force the components to zero.

Output voltage v of the current control arithmetic units 3A and 4A
_1d-pi and v _1q-pi are added to the output of the model voltage calculation unit 5A and the output of the first coordinate conversion unit 13, respectively, and the addition result is a fixed coordinate from the two axes of the rotating coordinate using the circular locus method or the like. Second coordinate conversion unit 1 for performing coordinate conversion into the three-phase PWM voltage component
After that, it is converted into a three-phase voltage control signal for the PWM inverter 11 via the signal.

The coordinate conversion unit 13 includes a conversion circuit 13A for reversely converting the three-phase voltage control signal into the biaxial coordinates of the rotating coordinates and the conversion coefficient units 13B and 13C.

The coordinate conversion section 14 includes a biaxial / three-phase conversion circuit 14A and voltage and PWM command conversion coefficient sections 14B and 1B.
4C, limiter circuits 14D, 1 for each of the three-phase components
It is composed of 4E and 14F.

According to this embodiment, the current control arithmetic unit 3A,
4A and the model voltage calculation unit 5A are speed type and high speed calculation is obtained. Further, by converting these calculation results into three-phase fixed coordinates by the coordinate conversion unit 14 and performing zero-phase voltage correction and output voltage limiter calculation, the coordinate conversion unit 13 inversely converts them into two axes. A plurality of types of limiter calculations that have been executed can be improved by one type, and the saturation amount in the current control calculation can be matched with the actual output voltage to eliminate a response delay in returning from the saturation level.

The model voltage calculation is performed by the switch circuit 12
By adopting a speed type having, it is possible to prevent the output voltage from becoming excessive at startup. This will be explained below.

FIG. 3 shows an example of response waveforms at each part at the start of excitation when the model voltage is added from the beginning while the motor is rotating. Exciting current command i _d * at the same time when the inverter starts operating
There is set, the model voltage _v 1dm, v 1qm outputs a voltage in a steady state. At this time, although the secondary magnetic flux of the electric motor 1 is not established, the voltage when the secondary magnetic flux in the steady state exists is output.

As a result, since the model voltage is output in the model voltage v _1qm direction, the q-axis current i _q increases. Originally, since the q-axis current command i _1q * = 0, the current control calculation unit 4A that serves as a feedback term works in the direction of canceling the model voltage, and the d-axis current i _d does not increase depending on the model voltage. , The current control arithmetic unit 3A, which also serves as a feedback term, tries to flow an exciting current.

As described above, when the model voltage is applied at the time of start-up, in the period in which the model magnetic flux and the actual magnetic flux of the electric motor do not coincide with each other, the electric current of the motor has a poorer responsiveness than that at the time of normal feedback and an overcurrent is generated. May occur. Therefore, as shown by the broken line, the current command i _q * may be provided with a cushion to suppress sudden output of the feedforward voltage.

In this embodiment, the model voltage is turned off at the time of start-up to prevent the q-axis current from becoming excessive. The off period of this model voltage is set to a time corresponding to the secondary time constant of the electric motor, and the excitation of the electric motor is established. In addition, when the model voltage is turned on after the excitation is established, the current waveform is disturbed when the model voltage is applied stepwise. Therefore, in the present embodiment, the current control calculation is performed by adding the difference of the model voltage component before the integral term. Only the change in the model from the start of model feedforward is added to the integral term in the section.

This correction is as shown in FIG. Model voltage is generated in the form of the difference between the model voltage at the time of the previous sample, but the start-up of the difference _{v 1dm (k) -v 1dm (} k-1), v 1qm (k) -
v _{1qm (k-1)} is deleted by the switch circuit 12.
Immediately thereafter, the switch circuit 12 is switched to add the model voltage after the magnetic flux of the electric motor 1 is established. At this time, although the torque current command i _q * is suddenly changed, a fast and stable response of the actual torque current i _q can be obtained by applying the model voltage.

As described above, since the model voltage calculation is also handled in the speed form, the model voltage can be smoothly switched on and off, and the switch on / off control can be simplified.

As shown in FIG. 3, there is a method of approximating with a first-order lag (cushion) in order to establish the magnetic flux in the calculation of the model voltage, but in this case, the gate is cut off for a moment due to a momentary power failure or the like. If the engine is restarted immediately after the occurrence of (the residual magnetic flux exists in the motor), the model voltage will increase from zero even though the magnetic flux actually exists, and the current response will be disturbed.

That is, since the magnetic flux of the electric motor changes according to the secondary time constant, the output current can obtain a stable response by adopting the configuration of this embodiment in which the model voltage is not generated during this period. Then, in order to apply the model voltage midway, the velocity type of this embodiment is suitable.

[0045]

As described above, according to the present invention, the current control calculation unit and the model voltage calculation unit are set to the speed type, and the output of the first coordinate conversion unit for obtaining the previous value is added to add two axes / three axes. Since the phase coordinate conversion is performed, the following effects are obtained.

(1) Conventionally, a plurality of types of limiters executed on each coordinate may be one type of three-phase component, and the saturation amount in the current control calculation can be matched with the actual output voltage. The response delay of returning from the output voltage saturation is eliminated.

(2) By subtracting the model voltage calculation and adding it to the integrator in the speed form, the feedforward of the model voltage can be deleted and applied easily, and no disturbance is given to the output current.

(3) The coordinate conversion can be easily performed, and the current control calculation and the like can be speed-typed to simplify the calculation processing of each part and obtain a high-speed response by high-speed calculation.

[Brief description of drawings]

FIG. 1 is a block diagram for explaining the principle of the present invention.

FIG. 2 is a block diagram showing an embodiment of the present invention.

FIG. 3 is a waveform chart of each part at the time of conventional startup.

FIG. 4 is a waveform diagram of each part at the time of startup of the embodiment.

FIG. 5 is a conventional block diagram.

FIG. 6 is a conventional limiter characteristic diagram.

[Explanation of symbols]

 3A, 4A ... Current control arithmetic unit 5A ... Model voltage arithmetic unit 12 ... Switch circuit 13 ... Coordinate conversion unit 14 ... Coordinate conversion unit.

Claims (2)

[Claims] 1. A current control calculation unit (3A, 4A) for performing speed-type proportional integral calculation with respect to a deviation between an excitation current command and a torque current command in a rotating coordinate system of a vector-controlled PWM inverter and respective detected values. ), A speed type model voltage calculation unit (5A) for obtaining an ideal voltage when a vector control condition is satisfied as a feedforward term from the excitation current command and the torque current command, and a three-phase voltage command of the PWM inverter for a rotating coordinate system. A first coordinate transformation unit (13) for inversely transforming the biaxial components of the exciting current component and the torque current component, and the corresponding biaxial components of the coordinate transformation unit, the model voltage computing unit and the current control computing unit, respectively. A second coordinate conversion unit (1) which converts the added value into a three-phase component having a fixed coordinate and has a limiter means to generate a three-phase voltage command of the PWM inverter.
4) and a current control device for a PWM inverter. 2. The model voltage calculation unit comprises a switch means for forcing the output to zero for a time corresponding to the secondary time constant of the induction motor, which is a load of the PWM inverter, from the start-up time. Current control device for PWM inverter.