JP3173022B2 - Control device for brushless DC motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for a brushless DC motor having a permanent magnet field.

[0002]

2. Description of the Related Art DC brushless motors that obtain a field magnetic flux by means of permanent magnets have not been able to pass a direct-axis current in the past. In other words, a control method that actively uses the direct-axis current has been proposed. Further, a DC brushless motor cannot directly weaken the field, but a control method has been proposed in which a speed control range is expanded by using a d-axis armature reaction to obtain an action equivalent to the weakened field. For example, the Institute of Electrical Engineers of Japan, Materials for the Study of Semiconductor Power Conversion, SPC-90-1
1, page 1 to page 8, "Segment Structure PM Motor Wide Range Speed Control".

Here, a control device for a DC brushless motor often has a current control system as a minor loop control system of a speed control system or as an independent control system. This current control system has, for example, a configuration shown in FIG. To be.

The DC power from the DC power supply 1 is converted into a voltage output of a PWM waveform controlled by the inverter main circuit 2 and supplied as an armature current of the DC brushless motor 3. The rotor position of the motor 3 is detected by the absolute encoder 4 as a phase signal θ. The current command I _1ref is set to a multiplier of the multipliers 5 ₁ and 5 ₂ , and the multiplier 5 _1
1, 5 to ₂ multiplicand 120 mutually from the sine wave generator 6
It is converted into a sine wave signal that is phase shifted. This sine wave phase is controlled according to the phase signal θ of the encoder 4.

[0005] Multiplier 5 _1, 5 ₂ of the output 3 to the motor 3
The u-phase and w-phase sine wave current commands I _u , I _{w of the} phase input are taken out, and these current commands I _u , I _w are current control amplifiers using the motor currents I _u ′, I _w ′ as feedback signals. 7 _1, 7 ₂ are proportional-integral calculation, the voltage command V _u of u-phase and w-phase, are taken out as V _w.

The voltage commands V _u and V _w are added by an adder 8 to generate a v-phase voltage command V _{v at} the output of the adder 9. These voltage commands V _u , V _v , V _w are used as comparison inputs of comparators 9 ₁ , 9 ₂ , 9 ₃ as a PWM generation circuit, and a triangular wave signal from the carrier generator 10 is given as a reference for comparison. Sine-wave approximated PWM waveforms are taken out from the outputs of 9 _{1 to} 9 ₃ , and these PWM waveforms are converted into gate signals of respective phases of the inverter main circuit 2, amplified by the gate circuit 11, and are switched by the phase switch elements of the inverter main circuit 2. Drive signal.

With such a configuration, the current control system controls the torque of the motor 3 by feedback-controlling the motor current.

[0008]

In a control device for a DC brushless motor, a constant torque region and a constant output region are set from the upper limit of an armature current and a terminal voltage as control conditions of the motor. The q-axis current is controlled.

Therefore, the conventional control device is based on the assumption that desired armature currents and terminal voltages are obtained. However, the above condition may change depending on the system configuration of the inverter.

For example, when the DC power supply 1 is configured to be obtained from an AC power supply by a rectifier, a battery (stand-by power supply) for uninterruptible power supply is provided, or a configuration having only a battery power supply is used. The DC voltage changes due to the voltage fluctuation or the voltage drop due to the discharge degree of the battery, and greatly changes particularly in the case of a battery power supply. When the internal resistance of the motor 3 is large, the resistance drop becomes equal to the DC voltage drop.

[0011] Such a change in DC voltage, especially a drop,
The voltage required to supply the current value commanded to the motor to the motor cannot be secured, and the outputs of the amplifiers 7 ₁ and 7 ₂ are saturated in the current control system that performs feedback control of the current.
In the PWM control of the sine wave approximation, the saturation output becomes close to a trapezoidal wave and the three-phase current becomes unbalanced, so that a normal torque current cannot be supplied.

The above problem is caused by having a current control system, and can be solved by performing a voltage control type control.
However, in voltage control, compared to current control due to the increase in current harmonics in the low-speed region due to the effect of securing a dead band to prevent short-circuiting between the upper and lower arms of each phase switch element of the inverter main circuit and the effect of motor parameter errors. Control performance deteriorates.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a control device capable of performing stable control operation without saturating a current control system even with DC voltage fluctuation.

[0014]

According to the present invention, in order to solve the above-mentioned problems, a current command I _1ref of a current supplied from an inverter to a brushless DC motor is provided with a rotor position of the motor.
Sine by the multiplier 5 _1, 5 ₂ multiplying a sine wave to match the phase
Wave current commands I _u , I _w are obtained, and the current commands I _u , I _w and
Calculate the deviations from the detected values I _u ′ and I _w ′ by proportional integration.
Inverter voltage command by current control amplifiers 7 ₁ and 7 ₂
V _u, to obtain a V _w, in the voltage command V _u, the control device for a brushless <br/> DC motor to a sine wave approximation PWM controlling an output voltage of the inverter <br/> over data based on V _w, the For inverter
From the DC input voltage V _dc , the motor rotational angular velocity ω, and the motor parameters, the current I _1max is _calculated as _follows : I _1max = {((V _dc / 2) ² − (nωλ) ² ) ^1/2 } / (nω
L) where n is the number of pole pairs of the motor λ is the number of flux linkages of the armature winding L is the arithmetic circuit 13 obtained from the armature inductance and the current I _1max and the current
A limiter that limits the command I _1ref and uses it as an input to the multiplier
And a circuit 12 .

Further, according to the present invention, the inverter output voltage V ₁ and the phase φ are calculated by the following equation V ₁ = (V _d ² + V _q ² ) ^1/2 φ = from the motor rotational angular velocity ω, the current command I _1ref and the motor parameters. tan ^-1 (V _d / V _q ) where V _d = nωL · I _1ref V _q = −RI _1ref −nωλ n: the number of motor pole pairs λ: the number of flux linkages of the armature winding L: the armature inductance R : a first arithmetic circuit 15 to 17 for obtaining from the armature resistance, the rotor position detection signal θ and the inverter DC voltage of the motor V _dc and the phase from φ inverter phase voltage command V _u, V _v, and V _w V _u = (V _dc / 2) sin (θ + φ) V _v = (V _dc / 2) sin (θ + φ−2π / 3) V _w = (V _dc / 2) sin (θ + φ + 2π / 3) 2 arithmetic circuits 18 and 19 and the voltage V ₁
And the DC voltage V _dc , and when V ₁ ≦ (V _dc / 2), the output of the current control amplifier is compared with the inverter voltage command.
_{And, V 1> (V dc /} 2) the voltage V _u of the second arithmetic circuit, V _v, switching the V _w and the voltage command of the inverter when the
Control circuits 20 and 21 are provided.

[0016]

In the former control device, the maximum voltage that the inverter can output in accordance with the DC voltage is determined, the maximum current I _1max that can be supplied to the motor is determined by this voltage, and the current command is limited by this current. Control to eliminate system saturation. This will be described in detail below.

The voltage equation of a DC brushless motor is d-
On the q coordinate, it is given by the following equation (1).

[0018]

(Equation 1)

Where R: armature resistance L: armature inductance n: number of pole pairs ω: rotational angular velocity λ: number of flux linkages of the armature winding by the field P: differential operator In the above equation (1), become the state (P = 0) at _{V d = R · i d -nωLi} q ... (2-1) V q = R · i q + nωLi d + nωλ ... (2-2). Further, in a region where the output voltage of the inverter becomes insufficient and saturation of the current control amplifier becomes a problem, a region where the motor rotational speed is high becomes a region. In a steady state where the rotational speed is relatively high, R≪nωL. 2-1), (2-2)
Expression is the _{V d = -nωLi q ... (3-1} ) V q = nωλ + nωLi d ... (3-2). The motor terminal voltage V ₁ and current I _{1 at} this time are

[0020]

(Equation 2)

## EQU1 ##

[0022] Here, since the DC brushless motor for controlling the motor current I ₁ in the same phase as the motor speed electromotive force Enuomegaramuda _{(q-axis), i d = 0, I} 1 = i q , and the above (3
-1), (3-2) and (4-1), (4-2)

[0023]

(Equation 3)

## EQU1 ## By giving the motor terminal voltage V ₁ and the rotational angular velocity ω from the equation (5-2), the motor current I ₁ is obtained. In other words, when the maximum voltage V _1max that can be output from the inverter is substituted into the equation (5-2) as V ₁ , the maximum motor current I that can flow according to each rotational angular velocity ω is obtained.
_1max is required.

On the other hand, the relationship between the inverter input DC voltage V _dc and the maximum output voltage V _1max is given by the following equation (6) when the current control amplifier is not saturated in the sine wave approximation PWM control, where V _1max = V _{1max dc} / 2 ... (6) Substituting this into equation (5-2)

[0026]

(Equation 4)

## EQU1 ##

Accordingly, by detecting the DC voltage V _dc and the rotational angular velocity ω from the equation (7), the maximum current I _1max that can normally control the current without saturating the current control amplifier can be obtained. _{By limiting} the current command I _1ref at _1max , the saturation of the current control amplifier is eliminated even when the DC voltage _Vdc fluctuates, and normal current control is always obtained.

Next, another control device of the present invention calculates the absolute value V ₁ and the phase φ of the inverter output voltage, and at the same time, the maximum output voltage V of the inverter determined by the DC voltage.
_1max seeking (= V _dc / 2), when the inverter output voltages V ₁ exceeds V _1max to control eliminates the saturation of the current control system by switching the voltage control from the current control system. This will be described in detail below.

The motor terminal voltage V ₁ is switched to the voltage control method stops the current feedback control in the case of exceeding the voltage V _1max to saturate the current control amplifier.

The voltage control method is based on (2-1), (2-
I _d = 0 by controlling the motor current I ₁ to the speed electromotive force nωλ the same phase in 2) the steady state type, I ₁ =
i _q = V _d by substituting _{-nωL I 1 ... (8-1) V} q = RI 1 + obtain Enuomegaramuda ... a (8-2), the motor terminal voltage V ₁ and the phase phi (4-1),
Obtained from (4-3), the three-phase motor terminal voltage V _u ,
_{V v, V w V u =} V 1 using the rotor position detection signal θ sin (θ + φ) ... (9-1) V v = V 1 sin (θ + φ-2π / 3) ... (9-2) V _w = V ₁ sin (θ + φ + 2π / 3) (9-3) Therefore, the inverter output voltages are (8-1) and (8
V _di = nωL I _1ref ... (10-1) in which the polarity of equation (2) is inverted V _qi = −RI _1ref −nωλ (10-2) is substituted into equations (4-1) and (4-3). And (9-
1), (9-2), can flow (9-3) equation is substituted into the inverter output voltage V _ui, V _vi, by providing the V _wi, a current equal to the motor current command I _1ref.

Therefore, the voltage V ₁ obtained by substituting the voltages V _di and V _qi of the equations (10-1) and (10-2) into the equation ( _4-1 ) is lower than the maximum voltage V _{1max of the} equation (6). sometimes performs current feedback control the saturation does not occur in the current control amplifier, by switching the voltage control from the current feedback control when the voltages V ₁ exceeds V _1max, current excellent in also control performance to variations in voltage V _dc Control is performed as much as possible, and when saturation of the current control amplifier cannot be avoided, voltage control is switched to obtain a stable operation state.

[0033]

FIG. 1 shows an embodiment of the present invention, and is a circuit diagram of a current control system according to the first embodiment. The current command I _1ref is output to the limiter circuit 12
It is inputted as a multiplier of the multiplier 5 _1, 5 ₂ via, the limitation of the current ± I _1max made by the limiter circuit 12. The arithmetic circuit 13 performs an arithmetic operation according to the equation (7) from the detected value of the DC voltage _Vdc of the inverter main circuit 2 (see FIG. 3) and outputs the limiter value of the limiter circuit 12. The phase angle calculation circuit 14 calculates the rotational angular velocity ω of the motor 3 by integrating the phase signal θ.

With such a configuration, the current command I _1ref is a limiter value ± I _1max
In limited, this limitation is controlled in accordance with the fluctuation of the DC voltage V _dc, it is possible to perform current control that maximizes the current command while eliminating the saturation of the current control amplifier 7 _1, 7 ₂ by the DC voltage drop. As a result, loss of control capability due to saturation of the current control amplifier and unbalance phenomenon of three-phase current can be prevented.

FIG. 2 shows another embodiment of the present invention, and is a control device circuit diagram corresponding to claim 2. In the figure, the arithmetic circuit 15 is the (10-1), (10-2) the inverter output voltage command V _qi by performing a calculation in accordance with equation, V _di
Ask for. The voltage calculation circuit 16 obtains a signal of the primary voltage V ₁ according to the equation (4-1) from the voltage commands V _qi and V _di , and the phase angle calculation circuit 17 calculates the phase angle φ according to the equation (4-3). Ask for. The divider 18 _divides the detection signal of the DC voltage _Vdc by half.
To determine the maximum output voltage V _1max of the inverter. 3
The phase voltage calculation circuit 19 calculates the values (9-1) and (9-) based on the voltage maximum value V _1max , the rotor position signal θ and the phase angle φ.
2), Inverter voltage command V _u according to (9-3),
Find V _v and V _w . The comparison circuit 20 compares the primary voltage V ₁ with the maximum value V _1max, and when the primary voltage V ₁ exceeds the maximum value, that is, when the current command I _1ref causes the saturation of the current control amplifiers 7 ₁ and 7 _2. When, the comparison output is obtained. Switching circuit 21 is normally at the time of the input of the current control amplifier 7 _1, 7 ₂ and the voltage command V _u which becomes the output from the adder 8, V _v, the V _w comparator 9 _to 93 _3, the comparison of the comparator 20 When an output occurs, the output V _u of the three-phase voltage operation circuit 19,
V _v, the V _w to switch to the input of the comparator 9 ₁ to 9 _3.

Therefore, in the present embodiment, when the current command I _1ref causes saturation of the current control system, the current control is switched from the current control to the voltage control, and this switching is controlled in accordance with the fluctuation of the DC voltage _Vdc . examples and while eliminating the saturation of the current control amplifier 7 _1, 7 ₂ by the same direct current voltage drop can perform current control that maximizes the current command.

Each arithmetic circuit in the embodiment is realized by an analog arithmetic operation, a digital arithmetic operation, or a combination thereof, or may be a software processing using a microcomputer.

[0038]

As described above, according to the present invention, the saturation of the current control system is determined by taking the current command I _1ref
When the saturation of the current control system is detected, the current command is limited or the current control is switched from voltage control to voltage control. Thus, there is an effect that stable control can be obtained by eliminating the saturation of the current, and current control can be performed with excellent control performance as much as possible for DC voltage fluctuation.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing one embodiment of the present invention.

FIG. 2 is a circuit diagram showing another embodiment of the present invention.

FIG. 3 is a configuration diagram of a conventional apparatus.

[Explanation of symbols]

2: Inverter main circuit, 3: Brushless DC motor, 5
₁ , 5 ₂ ... multiplier, 6 ... sine wave generator, 7 ₁ , 7 ₂ ... current control amplifier, 9 ₁ , 9 ₃ ... comparator, 10 ... carrier wave generator, 12 ... limiter circuit, 13 ... arithmetic circuit, 14 ... Phase angle calculation circuit, 15 ... Calculation circuit, 16 ... Voltage calculation circuit, 1
7: phase angle calculation circuit, 18: divider, 19: three-phase voltage calculation circuit, 20: comparison circuit, 21: switching circuit.

Claims (2)

(57) [Claims] A current command I _1ref of a current supplied from an inverter to a brushless DC motor includes a rotor position of the motor.
Sine by the multiplier 5 _1, 5 ₂ multiplying a sine wave to match the phase
Wave current commands I _u , I _w are obtained, and the current commands I _u , I _w and
Calculate the deviations from the detected values I _u ′ and I _w ′ by proportional integration.
Inverter voltage command by current control amplifiers 7 ₁ and 7 ₂
V _u, to obtain a V _w, in the voltage command V _u, the control device for a brushless <br/> DC motor to a sine wave approximation PWM controlling an output voltage of the inverter <br/> over data based on V _w, the From the DC input voltage _Vdc to the inverter , the motor rotational angular velocity ω, and the motor parameters, the current I _1max is _calculated by the following equation: I _1max = {((V _dc / 2) ² − (nωλ) ² ) ^1/2 } / (nω
L) where n is the number of pole pairs of the motor λ is the number of flux linkages of the armature winding L is the arithmetic circuit 13 obtained from the armature inductance and the current I _1max and the current
A limiter that limits the command I _1ref and uses it as an input to the multiplier
A control device for a brushless DC motor, comprising a circuit (12) . 2. A motor command I _1ref of a current supplied from an inverter to a brushless DC motor includes a rotor position of the motor.
Sine by the multiplier 5 _1, 5 ₂ multiplying a sine wave to match the phase
Wave current commands I _u , I _w are obtained, and the current commands I _u , I _w and
Calculate the deviations from the detected values I _u ′ and I _w ′ by proportional integration.
Inverter voltage command by current control amplifiers 7 ₁ and 7 ₂
V _u, to obtain a V _w, in the voltage command V _u, the control device for a brushless <br/> DC motor to a sine wave approximation PWM controlling an output voltage of the inverter <br/> over data based on V _w, the motor From the rotational angular velocity ω, the current command I _1ref and the motor parameters, the inverter output voltage V ₁ and the phase φ are calculated as _follows : V ₁ = (V _d ² + V _q ² ) ^1/2 φ = tan ⁻¹ (V _d / V _q ) Where V _d = nωL · I _1ref V _q = −RI _1ref −nωλ n: the number of motor pole pairs λ: the number of magnetic flux linkages of the armature winding L: the armature inductance R: the first calculation obtained from Circuits 15 to 17 , the rotor position detection signal θ of the motor, and the inverter DC voltage V
_{From dc} and the phase φ, each phase voltage command V _{u of} the inverter,
V _v, following equation _{V w V u = (V dc} / 2) sin (θ + φ) V v = (V dc / 2) sin (θ + φ-2π / 3) V w = (V dc / 2) sin (θ + φ + 2π / a second arithmetic circuits 18 and 19 obtained from 3), comparing the voltages V ₁ and the DC voltage _{V dc, V 1 ≦ (V} dc /
In the case of 2), the output of the current control amplifier is
When V ₁ > (V _dc / 2), the voltages V _u , V _v , and V _w of the second arithmetic circuit are set as voltage commands for the inverter.
And a switching control circuit (20 , 21) .