JPH0564551B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a power conversion device comprising a DC voltage source supplied with power from an AC power source and its load device.

[Technical background of the invention]

Load devices using a DC voltage source as a power source include a pulse width modulation control (PWM) inverter + induction motor, or a DC chopper + DC motor. There are not many problems when using a battery as this DC voltage source, but when obtaining DC voltage from a commercial power supply via a DC power converter, reactive power and harmonics generated on the commercial power supply side have recently become a problem. It's getting old.

FIG. 1 shows a configuration diagram of a conventional power conversion device using a PWM inverter and an induction motor as load devices. In the diagram, BUS is the three-phase AC power line, AF is the active filter device, TR is the power transformer,
CNV represents an AC/DC power converter, _C0 represents a DC smoothing capacitor, INV represents a PWM inverter, and IM represents an induction motor.

The AC/DC power converter CONV is composed of a rectifier bridge circuit SS ₁ , a thyristor bridge circuit SS ₂ , and DC reactors L ₁ and L ₂ , and supplies power to a smoothing capacitor C ₀ as a DC voltage source, and
It functions to regenerate the energy accumulated in C ₀ into AC power supply.

The PWM inverter INV converts the DC voltage V ₀ of the capacitor C ₀ into variable voltage variable frequency AC power, and is an induction motor IM (hereinafter referred to simply as the motor).
It serves to control the current and frequency supplied to the

It is known that the torque generated by the electric motor IM can be controlled by adjusting the slip frequency _sl . That is, when it is desired to accelerate the electric motor IM, the slip frequency _sl is controlled to be a positive value, and when it is desired to apply regenerative braking to decelerate, the slip frequency _sl is controlled to be a negative value.

When the electric motor IM is in power running mode, from the AC power supply BUS, transformer TR → rectifier SS ₁ → reactor L ₁ → smoothing capacitor C ₀ → PWM inverter
Active power is supplied to the motor IM via INV.

Conversely, if regenerative braking is applied to the electric motor IM,
Rotational energy is converted to active power, PWM inverter INV → smoothing capacitor C ₀ → reactor
L ₂ → Thyristor bridge circuit SS ₂ → Transformer TR
It is regenerated to the AC power BUS through the .

[Problems with background technology]

The thyristor bridge circuit SS ₂ operates only during regeneration and keeps the firing phase angle α at a nearly constant value within the range where power commutation (natural commutation) is possible. Generally, it is controlled at a constant value of α = approximately 150°. Ru. At this time, from the output voltage V _d1 of rectifier SS ₁ , the above thyristor bridge circuit
So that the output voltage V _d2 of SS ₂ is larger,
Select the secondary voltage of the power transformer TR. That is,
When the input voltage of the rectifier SS ₁ is V _S1 and the input voltage of the thyristor bridge circuit SS ₂ is V _S2 , the design is made to satisfy the following equation to prevent power supply short circuits.

V _d1 <V _d2 ∴kV _S1 , cos0°<-kV _S2・cosα ∴V _S1 <V _S2・cosα (atα=150°) In this conventional power converter, only rectifier SS ₁ operates during power running, so The fundamental wave power factor seen from the power supply side is always maintained at 1, but during regeneration, the thyristor bridge circuit SS ₂ is naturally rectified, so
The fundamental wave power factor is 0.866 because it operates at α = 150°.
decreases to a certain degree. A further problem is that the amount of reactive power Q taken from the power supply varies greatly during regeneration. That is, the reactive power Q is proportional to the magnitude of the current I _s supplied (or regenerated) from the power source and the sine value sin α of the firing phase angle α of the thyristor bridge circuit SS ₂ , so the phase angle α Even if controlled to be constant, if the regenerative current I _s changes from 0 to the maximum value, naturally Q = k _Q・I _s・sin α will also change from 0 to ・k _Q
(max) fluctuates as 0.5.

This is because fluctuations in the reactive power Q cause fluctuations in the power supply voltage and have various adverse effects on other electrical equipment.

Furthermore, the current supplied to the rectifier SS ₁ and the thyristor bridge circuit SS ₂ is a square wave current of 120° conduction (in the case of three-phase power supply) or 180° conduction (in the case of single-phase power supply), and the input current The harmonic components are large and can cause communication problems.

Conventionally, an active filter AF has been installed at the receiving end of a power converter in order to compensate for fluctuations in the reactive power Q and harmonic current, but its capacity has become comparable to that of a power converter, and the device It had the disadvantage of being large and expensive.

[Purpose of the invention]

The present invention has been made in view of the above, and it is possible to reduce harmonic components of the current supplied from an AC power source without providing a device such as the active filter AF at the power receiving end, and to reduce the fundamental wave power at the power receiving end. It is an object of the present invention to provide a power conversion device that can control the ratio to 1 or an arbitrary value.

[Summary of the invention]

In order to achieve this object, the present invention uses a first PWM inverter to control the active current supplied from the AC power supply or the combined current of the active current and the reactive current so that the DC voltage of the smoothing capacitor is almost constant. It is characterized by

[Embodiments of the invention]

FIG. 2 is a configuration diagram showing an embodiment of the power conversion device of the present invention. In the figure, TR is a power transformer,
L _s is the AC reactor, INV1 is the first PWM inverter, INV2 is the second PWM inverter, C ₀ is the DC smoothing capacitor, IM is the induction motor, PG is the rotary pulse generator, CONT1 is the control of the first PWM inverter The circuit CONT2 each represents a second PWM inverter control circuit.

The first PWM inverter INV1 is a thyristor
It consists of a full bridge with S ₁₁ to S ₁₄ and diodes D ₁₁ to _{D 14} , and the DC voltage V ₀ of the smoothing capacitor C ₀
The current I _s supplied from the power supply is controlled so that the current I s remains constant. Therefore, elements such as gate turn-off (GTO) thyristors are used as the thyristors S ₁₁ to _{S 14} . In some cases, a forced commutation circuit is also provided.

The second PWM inverter INV2 is a thyristor
It is configured as a full bridge with S ₂₁ to S ₂₆ and diodes D ₂₁ to _{D 26} and controls the three-phase current supplied to the motor IM. Similarly, elements such as GTO thyristors are used for the thyristors _S21 to _S26 , or a forced commutation circuit is added.

FIG. 3 is a block diagram showing an embodiment of the control circuit of the first PWM inverter. In the figure, C ₁ , C ₂ ,
C ₃ is a comparator, G ₁ (S) and G ₂ (S) are control compensation circuits, K _n is an operational amplifier, ML is a multiplier, TRG ₁ is a triangular wave generator, and GC is a gate circuit.

FIG. 4 is a block diagram showing an embodiment of the control circuit of the second PWM inverter. In the figure, F/V is a frequency-voltage converter, A is an adder, C _N , C _U1 ,
C _U2 , C _V1 , C _V2 , C _W1 , C _W2 are comparators, VRN is speed setting device, SFC is slip frequency control circuit, G _N (S),
C _U (S), G _V (S), G _W (S) are control compensation circuits, PTG is a three-phase pattern generator, M _U , ML _V , ML _W are multipliers,
TRG ₂ is a triangular wave generator, and GC _U , GC _V , and GC _W are gate circuits.

Hereinafter, the operation of the present invention will be explained with reference to the above, FIG. 2, FIG. 3, and FIG. 4.

First, a method for controlling the current supplied to the motor IM will be described.

The rotational speed N of the electric motor IM is determined by the rotational pulse generator PG.
and an analog quantity is provided by an F/V converter. The speed command value N ^* from the speed setter VRN and the above actual speed N are compared by a comparator C _N. The deviation ε _N = N ^* −N is controlled by the compensation circuit
G _N (S) and its output I _Ln is used as the load current peak value command.

Furthermore, a slip frequency control circuit SFC provides a slip frequency _sl of the motor IM. Add slip frequency _sl and motor rotation frequency _n using adder A to find _e = _n + _sl . PTG generates a three-phase unit sine wave of frequency _e . The three-phase pattern outputs _U , _Pv , and _PW are as shown in the following equation.

P _U = sin (2πet) P _V = sin (2πet−2π/3) P _W = sin (2πet + 2π/3) Multiply the peak value command I _Ln by the unit sine wave P _U by the multiplier M _U. Together, the command value of the current I _U of the U-phase armature winding of the motor IM is I _U ^* . comparator
C _U1 compares the U-phase current detector I _U and its command value I _U ^* , and inputs the deviation ε _U = I _U ^* −L _U to the next control compensation circuit G _U (S) . G _U (S) consists simply of a proportional amplifier, and outputs a voltage ε _U ′ proportional to the input ε _U . In another example, an integral element or a differential element may be added to optimize the current control system.

TRG ₂ generates an AC triangular wave V _a with a constant peak value, and serves as a carrier wave for a so-called PWM inverter. Compare the triangular wave V _a and the output signal ε _U ′ of the above G _U (S), and when ε _U ′V _a , the output signal is “1”, and when ε _U ′<V _a, the output signal is “0”. Comparator C _U2 and sends it to the next gate circuit GC _U. GC _U is thyristor S ₂₁ of inverter INV2
and _S24 , and the above output signal is “1”
When , S ₂₁ is turned on and S ₂₄ is turned off. Conversely, when the output signal is "0", _S21 is turned off and _S24 is turned on. Therefore, when the period of "1" is larger than the period of "0", the U-phase current I _U is increased in the positive direction, and conversely, when the period of "0" is larger than the period of "1", U
Increase the phase current I _U in the negative direction. That is, a voltage proportional to the output voltage ε _U ' of the control compensation circuit G _U (S) is applied to the U-phase armature winding of the motor IM.

For example, when the command value I _U ^* of the U-phase current becomes larger than the detected value I _U , ε _U = I _U ^* − _{I U} becomes a positive value,
ε _U ′, that is, the U-phase output voltage V _U is increased. Therefore, I _U increases and is finally controlled so that I _U =I _U ^* . Conversely, if I _U ^* < I _U , ε _U becomes a negative value, and ε _U ′, that is, by making V _U a negative value, I _U
decrease. After all, it eventually settles down to I _U = I _U ^* . Here, if the command value I _U ^* is changed in a sinusoidal manner, the actual current I _U is also controlled in a sinusoidal manner following it.

Similarly, the V-phase and W-phase armature currents I _V and I _W are also controlled according to the respective command values I _V ^* and I _W ^* .

As mentioned above, the peak values I _Ln of the armature current command values I _U ^* , I _V ^* , I _W ^* are given from the speed control circuit. When the speed command value N ^* is larger than the actual speed N, the deviation ε _N =N ^* -N becomes a positive value, increases the wave height value I _Ln , and increases the torque generated by the electric motor IM. Conversely, when N ^* <N, the deviation ε _N becomes a negative value, decreases the above-mentioned wave height value I _Ln , and IM
Reduces the generated torque. Therefore, finally
It settles down to N ^* = N. Here, the control compensation circuit G _N
(S) uses proportional + integral elements and is controlled so that the steady-state deviation ε _N becomes zero. Differential elements may also be used to further enhance control response.

The slip frequency _sl of the electric motor IM is, during power running,
It is set to a positive value, and to a negative value during regenerative braking, but the slip frequency sl is set to a constant value.The method is to control the slip frequency _sl = constant.The method is to change it according to the current peak value command I _{Ln.Furthermore} , the exciting current and secondary current of the motor IM There are methods such as changing the slip frequency _sl to control the primary current vector so that sl maintains an orthogonal relationship.
Here, a constant value is calculated from the slip frequency control circuit SFC.
An example of giving _sl was shown.

When the slip frequency _sl > 0, the motor IM enters the power running mode (motor mode), and power is supplied to the motor IM from the DC smoothing capacitor C ₀ serving as a constant voltage source. Conversely, when the slip frequency _sl < 0, the motor IM enters the regeneration mode (generator mode), power is regenerated from the motor IM to the DC smoothing capacitor _C0 , and the motor IM applies regenerative braking.

Next, a method of controlling the current I _s supplied from the power supply by the first PWM inverter INV1 will be explained.

In FIG. 3, V ₀ ^* is a DC voltage command value for smoothing capacitor C ₀ and is compared with voltage detection value V ₀ by comparator C ₁ . Deviation ε ₁ =V ₀ ^* −V ₀
is input to the next control compensation circuit G ₁ (S), where it is amplified or integrated. The output of the control compensation circuit G ₁ (S) is the peak value I _sn of the power supply current I _s and is therefore input to the multiplier ML.

On the other hand, the power supply voltage V _s =V _n ·sinωst is detected, and a unit sine wave synchronized with the power supply is generated via the operational amplifier Km. In other words, the peak value of the power supply voltage at Km
V is multiplied by the reciprocal of _n . By multiplying this unit sine wave by the peak value I _sn , a current command value I _s ^* as shown in the following equation is obtained.

I _s ^* = I _sn · sinωst However, ωs is the power supply angular frequency. Comparator C ₂ compares the current command value I _s ^* and the detected power supply current value I _s , and the deviation ε ₂ = I _s ^* −I _s get. This is input to the next control compensation circuit G ₂ (S) and amplified.

TRG ₁ is the carrier wave of the PWN inverter, i.e.
It generates a unit triangular wave V _b at about 500 Hz, and is compared with the output signal ε ₂ ' of the control compensation circuit G ₂ (S) by a comparator C ₃ .

The comparator _C3 generates an output signal "1" when ε ₂ 'V _b , and an output signal "0" when ε ₂ '<V _b , which are input to the gate circuit GC.

The thyristor in the main circuit has the above output signal “1”
When , S ₁₂ and S ₁₃ are on and S ₁₁ and S ₁₄ are off. Conversely, when the output signal is "0", _S11 and _S14 are turned on and _S12 and _S13 are turned off.

The current source I _s is determined by the voltage V _L applied to the AC reactor Ls. Also, the reactor voltage V _L is the current voltage V _s and the first PWN inverter
It is determined by the voltage V _c generated by INV.1.
Figure 5 shows the voltage and current vector diagram on the power supply side, (a) when power is being supplied from the power supply to the motor IM, and (b) when power is being regenerated from the motor IM to the power supply. represents the vector relationship of

In order for the power supply voltages Vs→ and Is→ to be in phase, see Figure 5a.
The voltage VL applied to the AC reactor is as follows:
Vs→−Vc→ must be advanced in phase by 90° from the power supply voltage Vs→.

6a to 6c show voltage and current waveforms corresponding to the vector diagram of FIG. 5a. The output V _c ' of the control compensation circuit G ₂ (S) in FIG. 3 is compared with the triangular wave V _b , and the waveform signal shown at the bottom of FIG. 6 is input to the gate circuit GC. As a result, the first PWM
Thyristors S ₁₁ to _{S 14} of inverter INV.1 are controlled to fire as described above, and the first PWM inverter
The fundamental wave component of the AC side voltage V _c of INV.1 is the above V _c ′
The voltage is proportional to . Therefore, AC reactor
The voltage V _L applied to L _s is the difference voltage between the power supply voltage V _s and the AC voltage V _c of the inverter, and as a result,
The power supply current I _s becomes a sine wave current having the same phase as the voltage V _s .

To make I _s and V _s have opposite phases as shown in Figure 5b,
This is achieved by controlling V _c in a manner similar to that described above.

In other words, the power supply current I _s can be set in phase or in phase with the power supply voltage V _s according to its command value I _s ^* , and furthermore,
The phase and magnitude can be freely controlled.

By controlling the power supply voltage V _s to be in the same phase or in opposite phase to the power supply voltage V _s , the power factor on the power supply side can always be kept at 1 regardless of whether power is being transferred or received. However, it is not certain what value the magnitude of the active power P _s =V _s ·I _s should be controlled to.

One possible method is to detect the active power P _i at the AC input terminal of the motor IM and control the above P _s accordingly. However, there is a drawback that P _s and P _i do not necessarily match due to losses in the inverter, etc. Another possible method is to similarly detect the active power P _DC using a DC line, but the same problem still remains.

In order to solve this problem, the device of the present invention detects the voltage V ₀ of the DC smoothing capacitor C ₀ and controls the power supply side active power P _s so that this voltage is constant.

When the detected amount V ₀ is smaller than the voltage command V ₀ ^* , the deviation ε ₁ =V ₀ ^* −V ₀ becomes a positive value, and the peak value I _sn of the power supply current becomes a positive value via G ₁ (S). Increase by value.
Therefore, the power supply voltage V _s and the current I _s are in phase, and the active power P _s is supplied from the power supply to the smoothing capacitor C ₀ . Therefore, the voltage V ₀ of the smoothing capacitor C ₀ increases,
It is controlled so that V ₀ ^* =V ₀ . When the load on the motor IM increases, V ₀ decreases, so that even greater active power P _s is supplied.

Furthermore, when the load on the electric motor IM is lightened,
When regenerative braking is applied and power is regenerated to the smoothing capacitor, V ₀ increases and V ₀ ^* < V ₀ . Then, ε ₁ =V ₀ ^* −V ₀ becomes a negative value, and G ₁ (S)
Through , the peak value command I _sn of the power supply voltage decreases,
Furthermore, it becomes a negative value. When I _sn becomes a negative value,
The phase of the current command I _s ^* is opposite to the power supply voltage V _s . As a result of controlling the current I _s according to the command I _s ^* , the active power P _s becomes a negative value, that is, it is regenerated to the power supply, and the energy of the smoothing capacitor C ₀ decreases accordingly. and the voltage V ₀ decreases. Eventually, it settles down to V ₀ ^* = V ₀ .

That is, there is no need to detect any active power, and active power including loss can be automatically exchanged.

Figure 7 shows the first PWM inverter in Figure 2.
This shows another embodiment of the control circuit CONT.1 of INV.1, which controls the reactive power Q at the power receiving end to an arbitrary value.

In the figure, the ROM includes _{a phase} shifter _, MLp,
MLq is a multiplier, AD is an adder, G ₃ (S) is a control compensation circuit, C ₄ is a comparator, and the other symbols are the same as those explained with the symbols in FIG.

The difference from FIG. 3 is that the power supply current command value I _s ^* includes both the active current I _p and the reactive current I _Q. As explained in Fig. 3, the effective current I _p is
The value of the smoothing capacitor V ₀ is given to be equal to the command value V ₀ ^* . The reactive power Q at the receiving end is controlled as follows.

Detect reactive power Q at the receiving end and set its command value Q ^*
Compare with. The deviation ε ₄ =Q ^* −Q is input to the next control compensation circuit G ₃ (S) to obtain the peak value command I _Qn of the reactive current I _Q. G ₃ (S) usually uses an integral element and is controlled so that the steady-state component of the deviation ε ₄ becomes zero.

The phase shifter POM outputs a unit sine wave q = C ₀₀ ωst that is 90° ahead of the power supply voltage V _s , and the multiplier ML _Q
I _Q = I _Qn・C ₀₀ ωst is created by

Effective current command I _p = I _pn sinωst by adder AD
and the reactive current command I _Q =I _Qn C ₀₀ ωst are added to obtain the power supply current command value I _s ^* .

If the reactive power command value Q ^* is set to 0, operation with a power factor of 1 is performed, as in the control shown in FIG.

If a load that takes delayed reactive power is connected to the same power supply, if a reactive power command value Q ^* that cancels out the delayed reactive power is given, the first PWM inverter
INV.1 is controlled to take the leading reactive power of Q=Q ^* in addition to the active power _Ps , so that the power factor including other loads can always be 1.

FIG. 8 shows another embodiment of the power conversion device of the present invention. As a load device for a DC voltage source,
In the figure, which shows the case where a DC chopper device and a DC motor are connected, CHO is a transistor chopper device,
D ₁ and D ₂ are diodes, SW is a regenerative brake switch, DCL is a DC reactor, DCM is a DC motor body, TG is a speed generator, and CONT3 is a control circuit for the chopper device. The circuit shown to the left of the smoothing capacitor C ₀ has the same configuration as that shown in FIG. The control on the power supply side is the same as described above, so it will be omitted, and next, the speed control of the DC motor DCM will be briefly explained.

When powering the DC motor DCM, close the switch SW and let the field current If _flow in the direction of the arrow.
The rotational speed N of the DC motor DCM is detected by the speed generator TG and compared with the speed setting value N ^* . N ^*
>N, the on period of the transistor chopper CHO is increased, the armature current _Ia is increased, and the torque generated by the DC motor DCM is increased. Conversely, N ^* < N
If , increase the off period of Chiyotupa CHO,
Decrease the armature current I _a to reduce the generated torque. Eventually, it settles down to N ^* = N.

The operation of the transistor chopper CHO is well known, and can be briefly explained as follows.

When the transistor chopper CHO is on, the current I _a is increased through the path of smoothing capacitor C ₀ → chopper CHO → DCL → DCM → switch SW. CHO
is turned off, the current I _a becomes DCM→SW→D ₁ →
It flows through the DCL path and gradually attenuates. Therefore,
By adjusting the ratio of the on period to the off period, the magnitude of the current I _a can be controlled.

When applying regenerative braking to the DC motor DCM to rapidly decelerate it, the following will occur.

First, open the switch SW. Next, the field current
Reverse I _f (in the opposite direction to the arrow in the figure). Then, the speed electromotive force of the DC motor DCM is reversed, and when the chopper CHO is turned on, DCM → D ₂ → CHO → DCL
Short circuit current I _a flows through the path of DC reactor DCL
energy is stored in. Next Chiyotsupa CHO
When turned off, the energy of DCL becomes DCM → D ₂ →
It is transferred to the smoothing capacitor C ₀ along the path C ₀ →D ₁ →DCL. When the voltage of smoothing capacitor C ₀ increases,
As explained above, this becomes active power and is regenerated to the AC power source.

The strength of the regenerative brake is proportional to the magnitude of the armature current _Ia , so by adjusting the on and off periods of the transistor chopper CHO,
Torque control can be performed by controlling Ia.

In this way, even if the load device is a chopper device + DC motor, the smoothing capacitor C ₀ which is the DC voltage source
The active current I _p supplied from the AC power source can be controlled so that the voltage V ₀ of is constant, and at the same time, the reactive power can also be controlled to take an arbitrary value.

〔Effect of the invention〕

As explained above, the power conversion device of the present invention has the following effects.

(1) By controlling the active current supplied from the power supply so that the voltage V ₀ of the smoothing capacitor remains constant, it is possible to supply the active power required by the load, including losses, and to avoid complicated An active power detection circuit and the like can be omitted, and an inexpensive device can be provided.

(2) At the same time as controlling the active power, the reactive power at the receiving end can also be controlled to an arbitrary value. In particular, operation with a power factor of 1 is also possible.

(3) Since the input current (current supplied from the power supply) is controlled sinusoidally, harmonic components can be reduced.

(4) Furthermore, it is no longer necessary to provide the conventionally required active filter device at the power receiving end, making it possible to reduce the size and weight of the entire device and provide an economical system.

[Brief explanation of the drawing]

FIG. 1 is a block diagram of a conventional power conversion device, FIG. 2 is a block diagram showing an embodiment of the power conversion device of the present invention, and FIG. 3 is a block diagram of the first PWM inverter of the device shown in FIG. FIG. 4 is a block diagram showing an embodiment of the control circuit of the second PWM inverter of the device shown in FIG. 2, and FIGS. Figures 6a to 6c are voltage and current waveform diagrams corresponding to the vector diagram in Figure 5a, and Figure 7 is the voltage and current vector diagram of the first PWM inverter of the device in Figure 2. FIG. 8 is a block diagram showing another embodiment of the control circuit. FIG. 8 is a block diagram showing another embodiment of the apparatus of the present invention. TR...Power transformer, _Ls ...AC reactor, INV1...1st PWM inverter, _C0 ...
Smoothing capacitor, INV2...Second PWN inverter, IM...Induction machine, PG...Rotating pulse generator, CONT1...Control circuit of INV1, CONT2
...Control circuit of INV2, CHO...Transistor chopper, SW...Regeneration switch, _D1 , _D2 ...
Diode, DCL...DC reactor, DCM...
...DC motor, TG...Speed generator.

Claims (1)

[Claims] 1. A single-phase AC power source, a full-bridge pulse width modulation control inverter connected to the single-phase AC power source via an AC reactor, and a pulse width modulation control inverter connected to the DC side of the pulse width modulation control inverter. The pulse width modulation control circuit includes a smoothing capacitor, a load device using the smoothing capacitor as a voltage source, and a pulse width modulation control circuit that controls the pulse width modulation inverter. voltage control means for detecting and outputting the peak value of the power supply current by controlling it to match the voltage command value, and synchronizing the peak value of the power supply current output from the voltage control means with the voltage of the single-phase AC power supply. a multiplier for multiplying by a unit sine wave, a current control means for controlling the effective current supplied from the single-phase AC power supply by using the output signal of the multiplier as a current command value, a triangular wave generator for generating a triangular wave, and the current control unit. A power conversion device comprising a gate circuit that compares an output signal of the means with a triangular wave output from the triangular wave generator to generate a gate signal for the pulse width modulation control inverter. 2. A single-phase AC power supply, a full-bridge pulse width modulation control inverter connected to the single-phase AC power supply via an AC reactor, a smoothing capacitor connected to the DC side of the pulse width modulation control inverter, and a smoothing capacitor connected to the DC side of the pulse width modulation control inverter. It includes a load device using a capacitor as a voltage source, and a pulse width modulation control circuit that controls the pulse width modulation inverter, and the pulse width modulation control circuit detects the DC voltage applied to the smoothing capacitor and generates a voltage command. A voltage control means that outputs the peak value of the active current as the command value of the power supply current by controlling it so that it matches the command value of the reactive power, detects the reactive power at the receiving end, and controls the peak value of the active current so that it matches the command value of the reactive power, and outputs the peak value of the active current as the command value of the power supply current. Reactive power control means outputs a peak value of as a command value of the power supply current, and generates a unit sine wave having the same phase as the voltage of the single-phase AC power supply and a unit sine wave having a phase lead of 90° from the voltage of the single-phase AC power supply. a first multiplier that multiplies the peak value of the active current output from the voltage control means by a unit sine wave having the same phase as the power supply voltage generated from the phase shifter; a second multiplier that multiplies the peak value of the output reactive current by a unit sine wave whose phase is 90° ahead of the power supply voltage generated by the phase shifter; output signals of these first and second multipliers; a current control means for controlling the power supply current supplied from the single-phase AC power supply by adding the value to a current command value; a triangular wave generator for generating a triangular wave; and a triangular wave for outputting the output signal of the current control means from the triangular wave generator. A power conversion device comprising a gate circuit that compares and generates a gate signal for the pulse width modulation control inverter based on the comparison result.