JPH0628517B2 - Power converter - Google Patents

Description

TECHNICAL FIELD OF THE INVENTION The present invention relates to a power converter including a DC voltage source that receives power from an AC power source and a load device for the DC voltage source.

[Technical background of the invention]

As a load device using a DC voltage source as a power supply, there are a pulse width modulation control (PWM) inverter + induction motor, or a DC chipper device + DC motor. When using a battery as the DC voltage source, there is no problem. However, when a DC voltage is obtained from a commercial power supply through an AC / DC power converter (converter), reactive power and harmonics generated on the commercial power supply side have become a problem in recent years. I'm running.

As a method for solving this problem, Japanese Patent Application No. 57-171886 has been proposed. In this system, a voltage type inverter with pulse width modulation control is used in the AC / DC power converter (converter), and an input current waveform supplied from an AC power supply is controlled to be a sine wave in phase with the power supply voltage. Therefore, the power factor seen from the power source side is always 1, and the harmonic component can be extremely reduced.

[Problems of background technology]

However, the voltage-type inverter controlled by pulse width modulation requires an element having a self-extinguishing ability such as a high-power transistor or a gate turn-off thyristor as its constituent element, which makes it difficult to increase the capacity. .

In particular, at present, it is very difficult to raise the DC voltage because the withstand voltage of the above element is low. Generally, elements are connected in series to increase the DC voltage rating by enlarging the capacitor of the snubber circuit in order to balance the voltage. According to this method, the loss of the snubber circuit becomes large, and moreover, it is necessary to pay attention to the timing of firing or extinguishing the element, which makes the design very troublesome.

[Object of the Invention]

The present invention has been made in view of the above, and it is possible to reduce the harmonic components of the current supplied from the AC power supply and to control the fundamental wave power factor at the power receiving end to 1 or an arbitrary value. It is an object of the present invention to provide a power conversion device that can easily increase its capacity and can increase its DC voltage rating.

[Outline of Invention]

In order to achieve this object, the present invention connects a plurality of pulse width modulation control converters in series on the DC side, individually controls the DC voltage of the smoothing capacitors connected to each converter, and loads the total voltage as a load. It is configured to be applied to the device.

Example of Invention

FIG. 1 is a configuration diagram showing an embodiment of a power conversion device of the present invention. In the figure, SUP is a single-phase AC power supply, TR is a power supply transformer, L _s1 , L _s2 AC reactor, CONV-1 and CONV-2 are pulse width modulation control (PWM) converters, C _d1 and C _d2 are DC smoothing capacitors, LOAD is a load device, and CT _s1 , CT _s2 , and GT _L are current detectors.

The PWM converter CONV-1 converts single-phase AC power into DC power, and includes four elements (for example, gate turn-off thyristors) S _{11 to} S ₁₄ having self-extinguishing ability and four wheeling diodes D ₁₁ It is composed of to D _14. Similarly, the PWM converter CONV-2 also has four self-extinguishing elements S
And it is configured ₂₁ to S ₂₄ and in four wheeling diode D ₂₁ to D _24.

The load device LOAD is, for example, a combination of an inverter that converts a DC voltage into a three-phase AC voltage having a variable voltage and a variable frequency and a three-phase squirrel-cage induction motor, and consumes a load current I _L on the DC side.

FIG. 2 shows a control block diagram of the device of FIG. In the figure, GV ₁ and GV ₂ are voltage control compensation circuits, GI ₁ and GI ₂ are current control compensation circuits, ML ₁ and ML ₂ are multipliers, K _m , K _m1 and K _m2 ,
K _L , K _X , and K _Y are operational amplifiers, C _{1 to} C ₈ are comparators, and A _{1 to} A ₄
Is an adder, TGR is a carrier wave generator, GC ₁₁ , GC ₁₂ , GC ₂₁ , GC
₂₂ is a gate control circuit.

The operation of the device of the present invention will be described below with reference to FIGS. 1 and 2.

First, the pulse width modulation control operation of the PWM converter CONV-1 will be described.

The carrier wave generator TBG generates triangular waves X and Y having a frequency of about 1 kHz. The phases of the triangular waves X and Y are 90 ° out of phase, and the converter CONV-1 is controlled by X and the converter CONV-2 is controlled by Y.

FIG. 3 shows a pulse width modulation operation waveform of the PWM converter CONV-1, where X is the output signal from the carrier generator TRG, is its inverted value, e _i1 is the control input signal, and g ₁₁ and g ₁₂ are The gate control signal, V _c1 , represents the AC side output voltage waveform of CONV-1.

That is, the triangular wave X is compared with the control input signal e _i1 to _generate the gate control signal g ₁₁ for the elements S ₁₁ and S ₁₃ , and the triangular wave is compared with the control input signal e _i1 for the gate control of the elements S ₁₂ and S ₁₄ . Producing signal g ₁₂ .

The elements S ₁₁ and S ₁₃ are alternately turned on and off, and both are not conducted at the same time. Similarly, the elements S ₁₂ and S ₁₄ are alternately turned on and off, and the elements are controlled so as not to be conductive at the same time.

In the case of e _i1 X, g ₁₁ becomes “1”, and an on signal is sent to the element S ₁₁ and an off signal is sent to S ₁₂ . Conversely, if e _i1 <X, then g ₁₁
Becomes "0", and an off signal of the element S ₁₁ and an on signal to S ₁₃ are sent.

Further, in the case of e _i1 , g ₁₂ becomes “1”, and an on signal is sent to S ₁₄ and an off signal is sent to S ₁₂ . Conversely, when e _i1 <, g ₁₂ becomes “0” and an off signal is sent to S ₁₄ and an on signal is sent to S ₁₂ .

As a result, when e _i1 > 0, the on periods of S ₁₁ and S ₁₄ overlap, and the AC-side output voltage V _c1 of the converter CONV-1 becomes V
The value is either _d1 (voltage of the DC smoothing capacitor C _d1 ) or 0 ^v . When e _i1 <0, the on-periods of S ₁₂ and S ₁₃ overlap each other, and V _c1 becomes either −V _d1 or 0 ^v . As a result, the output voltage V _c1 is controlled at twice the frequency of the carrier wave, and its average value (shown by the broken line) becomes a value proportional to the control input signal _i1 .

The same applies to the pulse width modulation control operation of the PWM converter CONV-2. However, carrier wave Y is out of phase with X by 90 °. This effect will be explained later.

Next, a method of controlling the power supply I _s1 from the power supply by the PWM converter CONV-1 will be described.

In FIG. 2, ▲ V ^* _d1 ▼ is a DC smoothing capacitor _Cd1.
Which is the DC voltage command value of ( ₁₎ and is compared with the voltage detection value V _d1 by the comparator C ₁ . Deviation _{^{ε 1 = ▲ V * d1 ▼}} -V d1 is input to the next voltage control compensation circuit GV _1, is amplified or integrated. The output signal of the control compensation circuit GV ₁ is the adder A ₁
Via, the _peak value I _m1 of the power supply current I _s1 is given. In addition,
In the adder A ₁ , a value proportional to the load current I _L ΔI _m = K _L ·
I _L is added as a correction amount.

On the other hand, the power source voltage V _s = V _m · sinω _t is detected, and a unit sine wave sin _{ω t} synchronized with the power source is generated via the operational amplifier K _m . That is, it is the reciprocal of V _m of the peak value of the power supply voltage at K _m . By multiplying this unit sine wave sin _{ω t} and the current peak value I _m1 , the current command value ▲
I ^* _s1 ▼ is obtained.

^{_{▲ I * s1 ▼ = I m1}} · sinω t ...... (1) where, ω is the power supply angular frequency multiplier ML ₁ is intended to carry out the multiplication. Yotsute to the comparator C _3, compares the current command value ▲ I ^* _s1 ▼ and supply current detection value I _s1, obtain deviation ^{_{ε 3 = ▲ I * s1 ▼}} -I s1. This is input to the next current control compensation circuit GI ₁ and inverted and amplified. If the amplification factor is multiplied by −K ₁ , the signal −K ₁ · ε ₃ is input to the adder A ₃ . Here, the secondary side terminal voltage V _s1 of the power transformer TR is detected and input to the adder A ₃ via the operational amplifier K _m1 .

Therefore, the PWM control input signal e _i1 is given by the following equation.

e _i1 = -K ₁ · ε ₃ + K _m1 · V _s1 (2) The AC side output voltage V _c1 of the converter CONV-1 has a value proportional to the input signal e _i1 as described above. . If k _c is its proportional constant, V _c1 can be expressed by the following equation.

V _c1 = k _c · e _i1 = k _c (−K ₁ · ε ₃ + K _m1 · V _s1 ) ... (3) Here, when K _m = 1 / k _c is set, V _c1 = −k _c K ₁ ε ₃ + V _s1 (4), and V _c1 is always output in the form of being added by the secondary voltage V _s1 of the power transformer. This is the power supply current I ₃
This is for removing the influence of the power supply voltage V _{s1 in} the control of.

That is, the AC reactor L _s1 of the apparatus shown in FIG. 1 has a power supply voltage V _s1 and an AC side output voltage V _{c1 of the} converter CONV-1.
A differential voltage V _L1 = V _s1 −V _c1 is applied, and as a result, the power supply current I _s1 flows as in the following equation.

By substituting V _c1 in equation (4) into equation (5) Becomes

▲ I ^* _s1 ▼> If there summer and I _s1, deviation epsilon ₃ becomes a positive value, increasing the supply current I _s1. Conversely, ▲ I ^* _s1 ▼ <I _s1
In such a case, the deviation ε ₃ has a negative value, and I _s1 decreases based on the equation (6). That is, finally I _s1 = ▲
It is controlled to be I ^* _s1 .

The DC smoothing capacitor C _d1 is controlled as follows.

When ▲ V ^* _d1 ▼> _Vd1 , the deviation ε ₁ = ▲ V ^* _d1 ▼ −
V _d1 becomes a positive value, and the _peak value I _m1 of the power supply current command value ▲ I ^* _s1 ▼ is increased. Since the power supply current I _s1 is controlled so as to match the command value ▲ I ^* _s1 ▼ as described above, the increase of the peak value I _m1 causes the active power P _s1 shown by the following equation to be _output from the power supply. Supplied.

This active power P _s1 is stored in the smoothing capacitor C _d1 via the converter CONV-1.

Therefore, the DC voltage V _d1 rises and is controlled so that V _d1 = V ^* _d1 .

On the contrary, when ▲ V ^* _d1 ▼ < _Vd1 , the deviation ε ₁ has a negative value, and the current peak value I _m1 is decreased to a further negative value. As a result, the power P _s1 supplied from the power source becomes a negative value and the energy of the smoothing capacitor C _d1 is regenerated to the power source. Therefore, the DC voltage V _d1 decreases, and again V _d1 = ▲ V ^*
_d1 ▼ and calm down.

Supply current I _s2 from the power supply by the PWM converter CONV-2
The control method of is also the same.

That is, the power supply current I _s2 is controlled to match the command value ▲ I ^* _s2 ▼, and the DC voltage V of the smoothing capacitor C _d2 is controlled.
_d2 is controlled so as to match the command value ▲ V ^* _d2 ▼.

The load device LOAD is applied with the sum of the DC voltages V _d1 and V _d2 .

When the load current I _L increases, the energy of the smoothing capacitors C _d1 and C _d2 is temporarily supplied to the load, and the DC voltages V _d1 and V _d2 thereof decrease. As a result, the supply currents I _s1 and I _s2 from the power source increase as described above, and the direct current voltage is controlled again to V _d1 = ▲ V ^* _d1 ▼ and V _d2 = ▲ V ^* _d2 ▼, respectively. Conversely, the load current I _L decreases or
When a negative value (power regeneration) is reached, the DC voltage V _d1 ,
Since V _d2 becomes larger than each command value, the supply currents I _s1 and I _s2 from the power source decrease as described above, or become negative values (power regeneration), and V _d1 = ▲ V ^* _d1 ▼, V _d2 =
Hold ▲ V ^* _d2 ▼.

However, when the load current I _L changes suddenly, the smoothing capacitor C
The DC voltages V _d1 and V _d2 of _d1 and C _d2 can only move slowly. What corrects this is the output ΔL _m of the operational amplifier K _L in FIG. DC voltage command value ▲ V ^* _d1 ▼ =
▲ V ^* _d2 ▼.

Since the load current I _L flows, the converter CONV−
The DC side power P _d1 of ₁ consumes only P _d1 = V _d1 · I _L (9). Therefore, if the AC power P _s1 commensurate with this is supplied from the power supply, the energy transfer to and from the smoothing capacitor C _d1 becomes a cancel cell, and the DC voltage V _d1 does not change.

(7) ignoring variation Coo2omega _t in active power P _s1 of formula Yotsute be combined with (9) Becomes Therefore, as the correction amount ΔI _m1 Should be given.

By selecting V _d1 = V _d2 and V _s1 = V _s2 , K _L1 = K _L2
= K _L = (2V _d1 ) / V _s1 .

Therefore, even if the load current I _L suddenly changes, it follows that and
I _m is changed immediately can increase or decrease the supply current I _s1, I _s2 from the power supply, as a result, the smoothing capacitor C
It is possible to reduce the fluctuations of the DC voltages V _d1 and V _d2 of _d1 and C _d2 .

When the DC voltages V _d1 and V _d2 of the two converters are controlled to be the same, the currents I _s1 and I _s2 supplied from the power supply to the converter have substantially the same value. Under these conditions, the carrier currents X and Y used for the pulse width modulation control of the two converters are shifted in phase by 90 ° as shown in FIG. I _s1
And the pulsation of I _s2 work to cancel each other, and the sum current I _s
The pulsation of = I _s1 + I _s2 is extremely small. That is, the input currents I _s1 and I _s2 of each converter are controlled at a frequency twice as high as the carrier frequency, and by further shifting the phases of X and Y by 90 °, the sum current I _s = I _s1 + I _s2 becomes the carrier frequency. The waveform is similar to that controlled by a frequency four times higher than As a result, the current I _s supplied from the power supply has extremely small harmonic components.

In the above, two converters have been explained, but when three converters are used, the carrier wave has a phase of 6
The same effect as described above can be obtained by using signals shifted by 0 °. Even when four or more converters are used, the same effect as above can be obtained by using a carrier having an appropriate phase difference.

In addition, in the embodiment of FIG. 1, the single-phase AC power supply has been described, but it goes without saying that the same can be applied to a three-phase power supply.

In the embodiment, by adding K _m1 · V _s1 to the adder A ₃ , correction is performed so that the power supply current I _s1 does not fluctuate due to fluctuations in the power supply voltage V _s1 , but when K _m1 · V _s1 is not added. As is clear from the above equation (5), if the power supply voltage V _s1 changes, I _s1 also changes, and if I _s1 ^* > I _s1 as a result, the deviation ε3 becomes a positive value, and conversely. When I _s1 ^* <I _s1 , the deviation ε3 becomes a negative value, and in the end it is controlled to be Is1 = I _s1 ^* as in the above-mentioned operation.

Further, in the above-mentioned embodiment, the correction amount ΔIm is added to suppress the fluctuations of the DC voltages V _d1 and V _d2 to a small change in the load current. As described above, the operation is still performed so as to hold V _d1 = V _d1 ^* V _d2 = V _d2 ^* .

〔The invention's effect〕

As described above, according to the device of the present invention, the following effects can be expected.

(1) Since the current supplied from the power supply is controlled to be a sine wave current in phase with the power supply voltage, the fundamental wave power factor at the receiving end is always maintained at 1 and the harmonic components of the input current are extremely small. Become.

(2) In order to increase the DC voltage applied to the load, the DC voltage control for each converter is performed, and by connecting them in series, the converter components (high-power transistors, gate turn-off thyristors, etc.) Even a low withstand voltage can be used, and system cost can be reduced by mass production.

(4) In PWM control of a plurality of converters, by displacing the phase of the carrier signal by an appropriate value, the pulsation of the current supplied from the power supply can be reduced by the above-mentioned number.

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of the power conversion device of the present invention, FIG. 2 is a control block diagram of the device of FIG. 1, and FIG. 3 is a time for explaining the operation of the device of FIG. It is a chart. SUP: Single-phase AC power supply, TR: Power supply transformer L _s1 , L _s2 ...... AC reactor CONV-1, CONV-2 ...... PWM converter C _d1 , C _d2 ...... DC smoothing capacitor LOAD ...... Load device

Claims (1)

[Claims] 1. A power transformer having a primary winding connected to an AC power source and having a plurality of secondary windings, and a plurality of pulse widths connected to the plurality of secondary windings via AC reactors, respectively. A modulation control converter, a DC side of the converter are connected in series, a load device that receives power from a DC power supply obtained by the series connection, a DC smoothing capacitor connected to the DC side of each converter, A voltage control compensating circuit for applying a deviation signal between each DC voltage setting value of the DC smoothing capacitor and each DC voltage detection value of each DC smoothing capacitor and outputting a signal according to the peak value of the AC current of each converter. , The output of this voltage control compensation circuit, the peak value is proportional to the output of the voltage control compensation circuit from the unit sine wave signal synchronized with the AC power supply voltage,
Means for obtaining each AC current reference of the pulse width modulation control converter whose phase is in phase with the AC power supply voltage, deviation signal between each AC current reference and each AC current detection value of each pulse width modulation control converter Each current control compensating circuit that generates a pulse width modulation control input signal that is applied and serves as a command value of the AC output voltage of each converter, and based on a carrier having a predetermined phase difference from the pulse width modulation control input signal An electric power converter characterized by controlling each pulse width modulation control converter.