JPH0442918B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to an AC motor drive device that receives power from a DC power source to drive an AC motor.

(Prior Art) Generally, AC motors include induction motors, synchronous motors, reluctance motors, and the like. As a conventional device for driving these AC motors by receiving power from a DC power supply, there is, for example, a voltage type self-excited inverter.

This voltage type self-excited inverter uses transistors and
It consists of a self-extinguishing element such as a GTO (gate turn-off thyristor), and can supply variable voltage and variable frequency AC power to an AC motor.

In this case, some motors use pulse width modulation (PWM) control to make the current supplied to the motor have a waveform approximating a sine wave in order to reduce control retorque pulsations.

(Problems to be Solved by the Invention) Conventional AC motor drive devices using such self-excited inverters require self-extinguishing elements such as transistors and GTOs, which increases the cost of the device. It was difficult to protect the device from such damage, and there were problems in terms of reliability.

Further, the switching frequency of a large capacity self-extinguishing element such as a GTO is limited to about 500 to 1000 Hz, and the modulation frequency of pulse width modulation control is also limited to the above frequency range. Therefore, when the rotational speed of the AC motor is increased, torque pulsations occur due to current pulsations.

Thus, conventional AC motor drive devices using self-excited inverters are suitable for small to medium capacity devices, but have the problem of being difficult to apply to large capacity and high speed AC motors.

The present invention was made to solve the above-mentioned problems, and enables a large-capacity AC motor to shift from low to high speed without torque pulsation, and also significantly improves reliability. The object of the present invention is to provide an AC motor drive device that can drive an AC motor.

[Structure of the invention]

(Means for Solving the Problems) The present invention provides a double converter in which the DC sides of two converters connected in antiparallel are connected to a DC power source via a reactor, and an AC side terminal of the double converter. a circulating current type cycloconverter whose input side is connected to the phase advance capacitor and whose output side is connected to the AC motor, such that the peak value of the voltage of the phase advance capacitor is approximately constant. The present invention is characterized by comprising means for controlling the double converter and means for controlling the cycloconverter so that the rotational speed of the AC motor matches a command value.

(Function) In the present invention, the control means controls the double converter so that the peak value of the voltage applied to the phase advance capacitor is approximately constant, and also so that the rotational speed of the AC motor matches the command value. A circulating current type cycloconverter is controlled.

At this time, the phase advance capacitor becomes a leading reactive power source for the double converter and the circulating current type cycloconverter, and its frequency (several hundred Hz) advances the delayed reactive power generated by the double converter and the cycloconverter. It is determined so that the leading reactive power of the phase capacitor is equal to the leading reactive power. In other words, by providing a phase control reference signal for the double converter and the cycloconverter using an external sine wave oscillator, the cycloconverter is circulated so that the frequency and phase of the voltage of the phase advance capacitor match the frequency and phase of this oscillator. Current flows.

The phase advance capacitor voltage established in this way allows the double converter and the cycloconverter to perform power conversion with only natural commutation, and it is possible to provide a high-speed, large-capacity AC motor drive device.

(Embodiment) FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention.

In the figure, the DC side of the double converter CONV is connected to the DC power supply V _d via a DC reactor L _d . This double converter CONV has a positive group converter SSP and a negative group converter SSN connected in antiparallel, and a phase advance capacitor CAP (the capacitors between the terminals are shown as one) is connected between the AC side terminals. has been done.

Further, the input side of a cycloconverter CC is connected to the phase advance capacitor CAP, and the AC motor M is connected to the output side of this cycloconverter CC. Here, the cycloconverter CC is mainly composed of converters SS ₁ to SS ₃ and DC reactors L ₁ to _{L 3} , and the converters SS ₁ to _{SS 3} are mutually insulated by a high frequency transformer TR.

Also, to control the double converter CONV, there is a current detector that detects the current flowing here.
CT _d , voltage detector PT CAP that detects the voltage applied to the phase advance capacitor _CAP , rectifier circuit D that rectifies the output of this voltage detector PT _CAP , voltage peak value of the phase advance capacitor CAP output from this rectifier circuit D V _CAP
The voltage control circuit AVR outputs a DC current command value I _d ^* based on the DC current command value I _d ^* , and the converter current control signal is output based on the deviation between this DC current command value I d * and the current detection value I _d of the current detector CT _d . Make current control circuit ACR ₁ ,
and external oscillator OSC output and current control circuit
Dual converter CONV based on ACR ₁ output and
Equipped with a phase control circuit PHC ₁ to control.

Furthermore, in order to control the circulating current type cycloconverter CC, a rotational pulse generator PG detects the rotation speed of the AC motor M, and current detectors CT _U , CT _V , CT _W detect each phase current of the AC motor M. , a speed control circuit SPC that obtains current command values I _U ^* , I _V ^* , I _W ^* based on the rotation speed detection value ω _r of the rotation pulse generator PG, and these current command values I _U ^* , I _V ^* , A current control circuit ACR ₂ that compares I _W ^* and detected values I _U , I _V , I _W of current detectors CT _U , CT _V , CT _W for each phase, integrates or proportionally amplifies and outputs the deviation, A phase control circuit PHC ₂ is provided to control converters SS ₁ to _{SS 3} based on the output of the current control circuit ACR 2 and the output of the external oscillator OSC.

After explaining the general operation of this embodiment configured as described above, the detailed operation will be explained.

First, the double converter CONV converts power between the DC power supply V _d and the phase advancing capacitor CAP, and when the DC current I _d flows in the direction of the arrow shown in the figure, the positive group converter SSP operates, direct current
Negative group converter when I _d is flowing in the opposite direction to the arrow, i.e. regenerating power
The SSN is controlled to operate, and the DC current I _d is further controlled so that the voltage applied to the phase advance capacitor CAP is approximately constant.

Next, the cycloconverter CC is of a delta-connected circulating current type and is controlled to supply three-phase alternating current power of variable voltage and variable frequency to the alternating current motor M using the phase advance capacitor CAP as a three-phase power source.

In addition, the phase control of the double converter CONV and the cycloconverter CC uses the signals of three-phase reference voltages e _a , e _b , e _c from the external oscillator OSC.
The frequencies and phases of the voltages V _a , V _b , and V _c of the phase advance capacitor CAP are controlled to the frequencies and phases of the reference voltages _ea , _eb , and _ec .

A detailed explanation of these operations will be given below.

First, the voltages of the phase advance capacitors CAP are V _a , V _b , V _c
The startup operation for establishing this will be explained.

Figure 2 shows the equivalent circuit of the positive group converter SSP and phase advance capacitor CAP at startup.
is composed of thyristors _S1 to _S6 . Furthermore, the phase advance capacitor CAP consists of C _ab , C _bc , and C _ca .

Now, if an ignition pulse is applied to thyristors S ₄ and S ₂ , the DC current I _d will be the power supply V _d ⁺ → reactor L _d
→Thyristor S ₄ →Capacitor C _ab →Thyristor S ₂
→ Path of power supply V _d ^- and power supply V _d ⁺ → Reactor L _d →
Thyristor S ₄ → Capacitor C _ca → Capacitor C _bc →
Flows through the path of thyristor S ₂ → power supply V _d ^- . As a result, capacitor C _ab is charged with DC voltage V _d ,
A voltage of −(V _d /2) is applied to the capacitors C _ac ′ and C _ca .

Figure 3 shows the thyristor S ₁ ~ of the positive group converter SSP.
The ignition mode of S ₆ and the waveforms of the voltage V ab applied to the phase advance capacitor C _ab and the phase voltage _{V a that} change correspondingly are shown.

After the mode of FIG. 2, thyristor _S3 is given a firing pulse. Then, a reverse bias voltage is applied to thyristor S ₂ by the voltage charged in capacitor C _bc , and S ₂ is turned off. That is, at startup, the phase advance capacitor CAP plays the role of a commutating capacitor. When the thyristors S ₄ and S ₃ are turned on, the voltages applied to the phase advance capacitors C _ab ′, C _bc ′, and C _ca also change.

The voltage V _ab applied to the phase advance capacitor C _ab changes stepwise depending on the firing mode of the thyristors S ₁ to _{S 6} as shown in FIG. But actually the voltage
Since V _ab is charged via the reactor L _d , it gradually rises as shown by the broken line. If that time is 2δ, the fundamental wave component of V _ab is delayed by δ. Further, the phase of the phase voltage V _a lags the line voltage V _ab by (π/6) radians.

Firing mode of thyristors S ₁ ~ S ₆ and above phase voltage
As can be seen by comparing V _a , the phase control angle α _p at startup is α _p =π−δ (radian) (1). Since δ is not very large, approximately it is operated at α _p ≒180°.

The output voltage V _p of the converter SSP at this time is as follows, assuming that the direction of the arrow in FIG. 2 is a positive value, V _p =k·V _cap ·cos α _p (2). However, k is a proportional positive number, and V _cap is the phase voltage peak value of the capacitor.

The inverted value _-Vp of this output voltage _Vp is balanced with the DC voltage _Vd . However, if this continues, the phase advance capacitor CAP will not be charged to a voltage higher than the DC voltage V _d .

Therefore, the ignition phase angle α _p is slightly shifted in the direction of 90°. Then, the inverted value -V _p of the output voltage V _p decreases, V _d > -V _p , and the DC current I _d flows in the direction of the arrow. As a result, active power P _d = V _d · I _d is supplied from the DC power supply V _d to the phase advance capacitor CAP, and energy P _d · h = (1/2) C _cap , V ² _cap is stored in the phase advance capacitor. . As a result, the voltage V _cap increases and settles down to V _d ≈−V _p . At this time, the current I _d is zero.

If it is desired to further increase V _cap , this can be achieved by further shifting the ignition phase angle α _p in the direction of 90° and decreasing the inverted value -V _p of the output voltage. α _p
= 90°, −V _p = 0 ^V , and theoretically the power supply voltage
Even if V _d is a very small value, the capacitor voltage V _cap can be charged to a large value. However, in reality, since there is circuit loss, it is essential to supply power for that amount.

In this way, the voltage of the phase advance capacitor CAP
V _cap can be charged to any value.

Next, the voltages V _a , V _b , V _c of the phase advance capacitor CAP established in this way are applied to the three-phase reference voltages ea , V a , V c applied to the phase control circuits PHC ₁ and PHC _{2 in FIG} .
Explain that the frequencies and phases of e _b and e _c match.

Figure 4 shows the phase control reference voltages e _a , e _b , e _c output from the external oscillator OSC in Figure 1 and the converters SS 1 to SS ₁ making up the circulating current type cycloconverter CC.
This shows the firing pulse mode of SS ₃ .

The phase control reference voltages e _a , e _b , and e _c are expressed as follows.

e _a = E _n・sin (ω _c・t) …(3) e _b = E _n・sin (ω _c・t−2π/3) …(4) e _c = E _n・sin (ω _c・t+2π/3)...(5) Here, E _n is the unit voltage peak value, ω=2πf _c is the angular frequency of the high frequency, and is selected to be approximately f _c =500 Hz, for example.

When the phase voltages V _a , V _b , V _c of the phase advancing capacitors CAP match the frequency and phase of the reference voltages e _a , e _b , e _c , each output voltage of converters SS ₁ to SS ₃ is as follows.
It will look like this:

V ₁ =k・V _cap・cos α ₁ ……(6) V ₂ =k・V _cap・cos α ₂ ……(7) V ₃ =k・V _cap・cos α ₃ ……(8) At startup are the output voltages of each converter V ₁ , V ₂ ,
Since V ₃ is set to zero, the control phase angles α ₁ , α ₂ ,
α ₃ is α ₁ =α ₂ =α ₃ =90°. of course
V ₁ +V ₂ +V ₃ = 0, and the cycloconverter CC
There is no increase or decrease in the circulating current.

From this state, if the frequency of the capacitor voltage becomes low, V _a ′, V _b ′,
Consider the case where V _c ′.

In this case, the firing phase angle of converter SS ₁ is from α ₁ to α ₁ ′, and the firing phase angle of SS ₂ is from α ₂ to α ₂ ′,
Furthermore, the firing phase angle of SS ₃ changes from α ₃ to α ₃ ′.
Therefore, V ₁ +V ₂ +V ₃ >0, which increases the circulating current of the cycloconverter CC. This circulating current becomes delayed reactive power on the input side of the cycloconverter CC as seen from the phase advancing capacitor CAP side.

FIG. 5 shows an equivalent circuit for one phase on the input side of the cycloconverter CC, in which the cycloconverter CC and the double converter CONV are replaced by a variable inductance L _CC that takes a lagging current. The resonant frequency _cap of this circuit is _cap = 1/(2π√· _cap ) (9).

An increase in the circulating current of the cycloconverter CC is equivalent to a decrease in the equivalent inductance L _CC , and the above frequency _cap increases and V _a ′, V _b ′,
The frequency _cap of V _c ′ is the frequency _C of the reference voltages e _a , e _b , e _c
approach.

Similarly, when _cap > _c , the circulating current decreases, L _CC increases, and it settles down to _cap = _c .

When the phase of the voltage of the phase advance capacitor CAP lags behind the phase of the reference voltage, the circulating current increases in the same way as when _cap < _c , and the phase advance capacitor CAP lags behind the phase of the reference voltage.
Advance the voltage phase of CAP. Conversely, when the voltage phase of the phase advance capacitor CAP leads the reference voltage, the circulating current decreases in the same way as when _cap > _c , and the voltage phase of the phase advance capacitor CAP is delayed.
In this way, the voltage V _a of the phase advance capacitor CAP,
The magnitude of the circulating current of the cycloconverter CC is automatically adjusted so that the frequency and phase of V _b and V _c match the reference voltages e _a , e _b , and e _c . Therefore, these capacitor voltages V _a , V _b , and V _c are expressed as follows.

V _a = V _cap・sin (ω _c・t) …(10) V _b = V _cap・sin (ω _c・t−2π/3) …(11) V _c = V cap・sin (ω _c・t+2π/3) ...(12) However, V _cap is the phase voltage peak value.

The same operation occurs when the rotational speed of the AC motor M increases and each converter SS ₁ to SS ₃ is generating an output voltage. That is, in a steady state (when the voltage phase of the phase advance capacitor CAP matches the reference voltage), V ₁ +V ₂ +V ₃ =0 is satisfied, and the circulating current of the cycloconverter CC does not increase or decrease. If it deviates from this state, the circulating current of CC is automatically adjusted, and the frequency and phase of the phase advance capacitor CAP are always operated to match that of the reference voltage.

As described above, the voltage of the phase advance capacitor CAP is
V _a , V _b , and V _c are established. Next, the control operations of the double converter CONV and the cycloconverter CC after these voltages are established will be explained.

FIG. 6 shows the control section of the double converter CONV in detail, and when it corresponds to the control circuit of FIG. 1, it becomes as follows.

First, the voltage control circuit AVR shown in FIG. 1 consists of the voltage setter VR shown in FIG. 6, the comparator _C1 , and the voltage control compensation circuit _Gc (s).

Further, the next current control circuit ACR ₁ includes the comparator C ₂ shown in FIG. 6, the current control compensation circuit G _I (s), the adder AD, the operational amplifier K _V , the inverter INV, the hysteresis circuit HS,
Consists of mono multi-circuit MM and logic circuit LC.

Furthermore, the phase control circuit PHC ₁ in Fig. 1 is the same as the phase control circuit PHP, PHN and switch circuit in Fig. 6.
Consists of AS ₁ and AS ₂ .

Here, the voltage of the phase advance capacitor CAP is detected by the voltage detector PT _cap , and the obtained three sets of voltages are rectified by the rectifier D. As a result, the voltage peak value V _cap of the phase advance capacitor CAP is input to the comparator C ₁ .

Further, a voltage command value V ^* _cap is outputted from the voltage setting device VR, and is compared with the detected value _Vcap by the comparator _C1 . The deviation between them ε _c =V ^* _cap −V _cap is input to the next voltage control compensation circuit G _c (s), where it is integrated or proportionally amplified. The output signal I _d ^* of this voltage control compensation circuit G _c (s) is sent to the comparator C ₂ as a DC current command value.
is input.

On the other hand, the DC current I _d is detected by the DC current transformer CT _d , and the deviation ε _I =I _d ^* −I _d from the above-mentioned current command value I _d ^* is determined by the comparator C ₂ . The deviation ε _I is integrated or proportionally amplified by the following current control compensation circuit G _I (s), and then sent to the positive group converter via the adder AD.
Input to SSP's phase control circuit PHP. Also,
After the adder AD, the signal is input to the phase control circuit PHN of the negative group converter SSN via the inverter INV.

The adder AD subtracts the DC voltage detection value V _d multiplied by a constant by the operational amplifier K _V from the output signal of the current control compensation circuit G _I (s) and outputs the result.

Therefore, the phase control input voltage _V〓p · _V〓N of the positive group and negative group converters is expressed as follows.

V〓 _p = G _I (s)・ε _I −K _V・V _d … (13) V〓 _N = −G _I (s)・ε _I +K _V・V _d … (14) On the other hand, voltage control compensation circuit The output signal I _d ^* of G _c (s) is changed to “1” or “0” via the hysteresis circuit HS.
is converted into a digital signal sig1. This signal sig
1, one generates the pulse signal sig2 via the monomulti-circuit MM, and the other is input as is to the logic circuit LC.

FIG. 7 is a time chart for explaining the operation, and shows each signal sig1 to sig4 when the DC current command I _d ^* changes from negative to positive or from positive to negative.

First, when the DC current command value I _d ^* changes from negative to positive, the output signal sig1 of the hysteresis circuit HS becomes "1" when it reaches I _d ^* ΔI ^* .
When the DC current command value I _d ^* changes from positive to negative,
When I _d ^* −ΔI ^* , the hysteresis circuit is activated.
The output signal of HS becomes "0".

The next mono multi circuit MM is a hysteresis circuit
A pulse signal sig2 with a pulse width ΔT is generated at the rise and fall of the HS output signal sig1.

The output signals sig3 and sig4 of the logic circuit LC are generated by the above-mentioned signals sig1 and sig2. i.e. sig
3 is created by ANDing sig1 and 2 (the inverted value of sig2), and sig4 is created by 1 (the inverted value of sig2).
It is obtained by taking the AND of 2 (inverted value of sig2) and 2 (inverted value of sig2).

The output signals sig3 and sig4 of the logic circuit LC thus obtained open and close the switch circuits _AS1 and _AS2, respectively. In other words, when sig3 is “1”, switch AS ₁ is turned on and the positive group converter SSP
Give a gate signal to. Also, when sig4 is “1”, switch _AS2 is turned on and negative group converter SSN
Give a gate signal to.

Therefore, when the DC current command value I _d ^* changes from negative to positive, the negative group converter SSN stops operating when I _d ^* ΔI ^* , and then the positive group converter SSP starts operating after a period of ΔT. Start. This time ΔT is to prevent the positive group and negative group converters from operating at the same time, and the actual flowing DC current I _d
It is determined by taking into consideration the time when becomes completely zero.

Conversely, when I _d ^* changes from positive to negative, I _d ^* −
When ΔI ^* is reached, the positive group converter SSP stops operating, and after a period of ΔT, the negative group converter SSN starts operating. At this time, since I _d ^* <0, the mode is in which power is regenerated to the DC power supply V _d .

The actual DC current I _d is determined according to its command value I _d ^* _.
≒ I _d ^* However, at the time of switching when I d ^* changes from positive to negative or from negative to positive, I _d * as shown by the solid line with respect to the command value I _d ^* shown by the broken line become. That is, there is a rest period in which the current becomes zero when switching from positive to negative.

The pause period ΔT is several tens of milliseconds (m sec)
It is sufficient if there is one, and there is no practical problem even if power running and regeneration are performed frequently.

When switch circuit AS ₁ is on and AS ₂ is off, direct current I _d is controlled by positive group converter SSP as follows.

When I _d ^* > I _d , the deviation ε _I = I _d ^* −I _d becomes a positive value, and the phase control input _{v〓 p} of the positive group converter SSP is increase. The output voltage V _p of the positive group converter SSP increases in proportion to this phase control input v〓 _p , and as a result, the DC current
I _d increases and is controlled so that I _d ≈I _d ^* .

Conversely, when I _d ^* <I _d , the deviation ε _I becomes a negative value and decreases the phase control input v〓 _p . Therefore, the output voltage V _p of the positive group converter SSP decreases,
Decrease the DC current I _d . Again, it is controlled so that I _d ≒ I _d ^* .

At this time, the compensation signal input to the adder AD -
k _v · V _d is a voltage opposite to the DC power supply voltage V _d that is generated from the positive group converter SSP, and when the signal from the current control compensation circuit G _I (s) is zero, V _p
= -V _d , and is in equilibrium with the DC voltage V _d . Therefore, there is no increase or decrease in DC current I _d in this state.

Next, the control operation of the DC current I _d when the switch circuit AS ₂ is on and AS ₁ is off will be explained. In this case, the negative group converter SSN operates,
Electric power will be regenerated into a DC power source.

For I _d ^* (e.g., I _d ^* = −90A, I _d = −
100A), the deviation ε _I becomes a positive value, and the phase control input v〓 _N = −G _I (s)・ε _I +K _v・V of the negative group converter SSN is applied via the current control compensation circuit G _I (s). Decrease _d .
Therefore, V _d >V _N , the DC current I _d increases in the direction of the arrow, and settles down to I _d ≈I _d ^* .

Conversely, if I _d ^* < I _d (for example, I _d ^* = −
100A, I _d = −90A), the deviation εI is a negative value, and G _I
(s) to increase the phase control input v〓 _N. Therefore, V _d <V _N , and the DC current I _d is decreased,
Again, it is controlled so that I _d ≒ I _d ^* .

Next, the control operation of the voltage peak value V _cap of the high frequency phase advancing capacitor CAP will be explained.

When V ^* _cap > V _cap , the deviation ε _c = V ^* _cap − V _cap
becomes a positive value, and increases the DC current command value I _d ^* via the voltage control compensation circuit G _c (s). As described above, the actual DC current I _d is controlled in accordance with the DC current command value I _d ^* so that I _d ≈I _d ^* . As a result,
Electric power P _d = V _d・I _d is supplied from the DC power supply V _d , and the energy is accumulated in the phase advance capacitor CAP as P _d・t=1/2C _cap V ² _cap (15), and the voltage
Increase V _cap . Eventually, it settles down to V _cap ≒ V ^* _cap .

When V ^* _cap < V _cap , the deviation ε _c becomes a negative value and decreases the DC current command value I _d ^* via G _c (s),
Furthermore, make it a negative value. When I _d ≒ I _d ^* < 0, the energy of the phase advance capacitor CAP is regenerated to the DC power supply V _d , and the voltage V _cap is decreased, and control is performed so that V _cap ≈ V ^* _cap .

In this way, the voltage peak value V _cap of the phase advance capacitor CAP is controlled to match the command value V ^* _cap .

Next, the operation of controlling the currents I _U , _IV , and I _W supplied to the AC motor M by the cycloconverter CC will be explained.

Note that here, a case will be described in which an induction motor is used as the electric motor M.

Figure 8 shows the circulating current type cycloconverter CC
FIG. 2 is a block diagram showing a detailed configuration of a control section of the controller.
Corresponding to the control circuit shown in FIG. 1, the result is as follows.

First, the speed control circuit SPC in Fig. 1 includes the comparator C ₃ in Fig. 8, the speed control compensation circuit G _N (s), the excitation current setting device EX, the calculation circuits CAL1 to CAL3, and the three-phase sine wave pattern generator PTG. and multipliers ML _U to ML _W.

In addition, the current control circuit ACR ₂ is the comparator shown in Fig. 8.
C ₄ to _{C 6} , current control compensation circuits G _U (s), G _V (s), G _W (s), and adders A ₁ to _{A 3} .

Further, the phase control circuit PHC ₂ of FIG. 1 is composed of phase control circuits PHC ₂₁ to PHC ₂₃ of FIG. 8.

First, the speed control operation of the induction motor M will be explained.

An induction motor in which the secondary current I and the excitation current _Ie are vectorally orthogonal so that they can be controlled independently is known as a vector control induction machine. Here, an example of speed control using this method is taken.

There are many literatures on vector control techniques, so a detailed explanation will be omitted and only a summary will be given.

First, a pulse train proportional to the rotational speed ω _r is extracted from the rotational pulse generator PG directly connected to the rotor of the electric motor.

Comparator C ₃ compares this rotational speed N (=ω _r ) with its command value N ^* , and inputs the deviation ε _N =N ^* −N to the speed control compensation circuit G _N (s). G _N (s) consists of a proportional element, an integral element, etc., and provides a torque current command I〓 ^* as an output.

Further, the detected rotational speed value ω _r is input to the excitation current setting device EX, and provides an excitation current command I _e ^* .

The torque current command I ^* and the excitation current command _Ie ^* are input to the calculation circuits CAL1 to CAL2 to perform the following calculations.

That is, in calculation circuit CAL1, ω ^* _sl = R ^* / _r / L ^* / _r・I ^* / 〓 / I ^* / _e ...... (16) R ^* _r : Secondary resistance L ^* _r : Secondary inductance For calculation Therefore, find the slip angular frequency ω ^* _sl .

In addition, the calculation circuit CAL2 calculates the phase of the primary current command value I _L ^* with respect to the excitation current I ^* _e by calculating θ ^* _r = tan ^-1 I ^* / 〓 / I ^* / _e (17). Find the angle θ _r ^* .

Furthermore, in calculation color CAL3, the peak value I _Ln of the primary current command value I _L ^* is determined by the calculation of I _Ln = √ ^*2 _e + ^*2 〓 ...(18)
seek.

Figure 9 shows the current vector diagram of this induction motor. The exciting current I _e ^* and the secondary current (torque current) I ^* are in an orthogonal relationship, and the generated torque T _e of this motor is calculated by the following formula: It can be expressed as

T _e = K _e・I〓 ^*・I _e ^* ……(19) Normally, a constant value is given as the excitation current command I _e ^* , and the generated torque T _e of the motor is the secondary current command (torque current command) I 〓 Controlled by changing ^* .

However, when operating at a rotational speed higher than the rated speed, field weakening control is performed and the excitation current setting
The excitation current command I _e ^* may be changed depending on the rotational speed ω _r by EX.

The slip angular frequency ω ^* _sl obtained in this way,
Phase angle θ _r ^* and rotational angular frequency (rotation speed detection value) ω _r
is input to the sine wave pattern generator PTG to obtain the next three-phase unit sine waves φ _U , φ _V , φ _W .

φ _U = sin {(ω _r + ω ^* _sl )・t + θ _r ^* } …(20) φ _V = sin {(ω _r + ω ^* _sl )・t + θ _r ^* −2π/3}
…(12) φ _W = sin {(ω _r +ω ^* _sl )・t + θ _r ^* +2π/3}
...(22) These unit sine waves φ _U , φ _V , φ _W determine the frequency and phase of the primary current I _L supplied to the induction motor M.

The multipliers ML _U to ML _W generate a three-phase unit sine wave φ _U ,
The command values I _U *, I _{V *} , I _W ^* of the three-phase current (primary current) supplied to the induction motor M are obtained by multiplying _{φ V} ^, φ _W and ^the peak value command I _Ln .

I ^* _U = I _Ln・sin {(ω _r +ω ^* _sl )・t+θ _r ^* ) ...(23) I ^* _V = I _Ln・sin {(ω _r +ω ^* _sl )・t+θ _r ^* −2π/3
)
...(24) I ^* _W = I _Ln・sin {(ω _r +ω ^* _sl )・t + θ _r ^* −2π/3
}
...(24) Vector control of an induction motor is based on the excitation current I _e and 2
The feature is that the next current I〓 can be controlled independently.
Therefore, by changing the magnitude of the secondary current I while keeping the excitation current I _e of the motor constant, the generated torque can be controlled and a speed control response equivalent to that of a DC machine can be achieved. It becomes possible.

Next, the primary current command value given as above
The operation of controlling the actual currents I _U , I _V ^{, and} _{I W according} to I _U ^* , I _{V *} , and I W * will be explained.

The armature currents I _U , I _V , I _W are detected by the current transformers CT _U , CT _V , CT _W shown in FIG.

These motor primary current detection values I _U , I _V , I _W are input to comparators C ₄ to C ₆ respectively, and the command values I _U ^* , I _V ^* , I _W ^* are obtained.
Compare each.

The control operation will be explained using the U-phase current as an example.

The actual current I _U and the command value I _U ^* are compared by the comparator _C4 , and the deviation ε _U = I _U ^* −I _U is calculated by the current control compensation circuit G _U
Enter (s). G _U (s) integrates or proportionally amplifies the output and sends it to the phase control circuit via adder _A1 .
Input to PHC ₂₁ . Further, the inverted value of the output of G _U (s) is input to the phase control circuit PHC ₂₃ via the adder A ₃ .

The output voltages V ₁ to V ₃ of each converter SS ₁ to _{SS 3} are proportional to the input voltages v〓 ₁ to v〓 ₃ of the phase control circuits PHC ₂₁ to _{PHC 23} .

Therefore, when I _U ^* > I _U , the deviation ε _U becomes a positive value, and the input voltage v〓 ₁ of the phase control circuit PHC ₂₁ is increased via the control compensation circuit G _U (s), and the converter The output voltage V ₁ of SS ₁ is increased in the direction of the arrow in FIG. At the same time, the input voltage v〓 ₃ of the phase control circuit PHC ₂₃ is decreased, and the output voltage of the converter SS ₃ is
V ₃ is generated in the opposite direction to the arrow in Figure 1. As a result, the output current I ₁ of converter SS ₁ increases,
The output current I ₃ of converter SS ₃ decreases.

As a result, the U-phase current I _U =I ₁ −I ₃ of the motor increases and is controlled so that I _U ≈I _U ^* .

Conversely, when I _U ^* < I _U , the deviation ε _U becomes a negative value, and the output voltage V ₁ decreases and V ₃ increases. Therefore, I _U =I ₁ −I ₃ is controlled to decrease, so that I _U ≈I _U ^* . If the command value I _U ^* is changed in a sinusoidal manner, the actual current also becomes I _U ≈I _U ^* , and a sinusoidal current is supplied to the induction motor M.

The V-phase and W-phase currents I _V and I _W are similarly controlled.

Therefore, the rotational speed N of the induction motor M is controlled as follows.

When N ^* >N, the deviation ε _N becomes a positive value and increases the torque current (secondary current) command I〓 ^* via the control compensation circuit G _N (s).

As a result, the peak value I Ln and phase angle θ _r ^* of the primary current command I _L ^* (I _U ^* , I _V ^* , I _W ^* ) of the induction motor shown in FIG. ₉ are increased, and the actual current I _U , I _V , and I _W are also controlled accordingly.

In this way, the actual secondary current I of the induction motor M increases, increasing the generated torque T _e and accelerating it. As a result, N increases and is controlled so that N≈N ^* .

Conversely, when N ^* < N, the deviation ε _N becomes a negative value, decreasing the torque current command I〓 ^* , and increasing the primary current command I _L ^* (I _U ^* , I _V ^* , I _W ^* ) decreases the peak value I _Ln and phase angle θ _r ^* . Therefore, the generated torque T _e decreases, and the rotational speed N decreases, so that control is performed so that N≈N ^* .

In the above example, a delta-connected circulating current cycloconverter was used as the circulating current cycloconverter, but it is also possible to use a circulating current cycloconverter with normal forward and reverse converters for three phases. It goes without saying that there is.

Further, in the above embodiment, the case where an induction motor is used as the AC motor has been described, but the present invention can be similarly applied to a case where a synchronous motor, a reluctance motor, a hysteresis motor, etc. are driven.

Further, in the above embodiment, a so-called non-circulating current type double converter has been described as the double converter, but it goes without saying that a circulating current type double converter can be implemented in the same manner. In that case, there is no need to temporarily reduce the output current (DC current I _d ) to zero when switching the operation of the forward/reverse converter, allowing smooth power running and regenerative operation.

(Effects of the Invention) As is clear from the above explanation, according to the present invention, the following effects can be obtained.

Power is supplied from a DC voltage source to establish a high-frequency reactive power source for the phase advance capacitor, and the voltage of this high-frequency reactive power source is used to naturally commutate the double converter and the circulating current type cycloconverter. Therefore, compared to conventional self-excited inverters, it is possible to drive an AC motor at variable speed without using forced commutation circuits or self-extinguishing elements (transistors, GTOs, etc.). This increases the reliability of the device and allows for larger capacity.

Further, the circulating current type cycloconverter that drives the AC motor can obtain an output frequency equivalent to the input frequency (frequency of the voltage applied to the phase advance capacitor), and can rotate the AC motor at an extremely high speed. For example, if the frequency _cap of the voltage applied to the phase advance capacitor is 500Hz, the armature winding of the AC motor should be supplied with sinusoidal currents I _U , I _V , and I _W at frequencies of about 0 to 500 Hz. I can do it.
In other words, in the case of a two-pole electric motor, the rotation speed is
It can reach 30,000rpm, making ultra-high-speed operation possible.
As a result, the blower motor, which conventionally had to be sped up using a gear, does not need a gear, and the operating efficiency is improved, and the blower motor can be made smaller and lighter.

In addition, when the rotation speed is set to around 3000 rpm, the number of poles of the motor can be reduced to 20, which not only reduces torque ripple at low speeds, but also improves speed control accuracy by an order of magnitude compared to conventional methods. Take.

Furthermore, the current I _U・I _V supplied to the motor,
_IW is controlled to a sine wave, resulting in a device with extremely small torque ripple. At the same time, electromagnetic noise is eliminated, making it possible to provide a quiet rotating machine.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention, FIG. 2 is an equivalent circuit diagram for explaining the operation of the main parts of the embodiment, and FIGS. 3 and 4 are diagrams of the embodiment. A time chart for explaining the operation of the main parts, FIG. 5 is an equivalent circuit diagram for explaining the operation of the main parts of the same embodiment, and FIG. 6 is a time chart for explaining the operation of the main parts of the same embodiment.
A block diagram showing details of the control section of the heavy converter,
FIG. 7 is a time chart for explaining the operation of the control section, and FIG. 8 is a block diagram showing details of the control section of the cycloconverter constituting the above embodiment.
FIG. 9 is a vector diagram for explaining the operation of the control section. V _d …DC power supply, L _d …DC reactor, CONV
...double converter, CAP...phase advance capacitor, CC
...Circulating cycloconverter, M...AC motor,
SSP, SSN...Positive group, negative group converter, SS ₁ to _{SS 3}
…separately excited converter, TR…high frequency transformer, L ₁ ~
_L3 ...DC reactor, CT _d , CT _U , CT _V , CT _W ...
Current detector, PT _cap ...voltage detector, D...rectifier circuit, AVR...voltage control circuit, SPC...speed control circuit,
ACR ₁ , ACR ₂ ...Current control circuit, PHC ₁ , PHC ₂ ...
Phase control circuit, OSC...external oscillator.

Claims (1)

[Claims] 1 A double converter in which the DC sides of two converters connected in antiparallel are connected to a DC power supply via a reactor, a phase advance capacitor connected between the AC side terminals of this double converter, and a phase advance capacitor whose input side is connected to the a circulating current cycloconverter connected to a phase capacitor and whose output side is connected to an alternating current motor;
means for controlling the double converter so that the peak value of the voltage of the phase advance capacitor is substantially constant;
An AC motor drive device comprising: means for controlling the cycloconverter so that the rotational speed of the AC motor matches a command value.