JPH01110092A - Variable speed controller for induction motor - Google Patents

Abstract

PURPOSE:To control the speed of an induction motor with high performance without directly detecting a speed and a motor voltage by so adjusting as to set a deviation between a command value of a motor voltage component and the output from a magnetizing current adjustor to zero. CONSTITUTION: E calculator 30 in an adjusting loop 300 obtains the calculated value of an M-axis component voltage of a voltage vector from a torque current, command values i'T, i'M of magnetizing currents and a motor constant, and outputs a deviation E from the output V''M of a magnetizing current adjustor 6. Since the E is generated due to the difference of a presumed speed value and an actual speed value and the noncoincidence in magnetic flux axes, this deviation E is input to an adjustor 25 in the loop 300, an adjusting loop in which the E always becomes zero is formed, thereby calculating its speed.

Description

[Detailed Description of the Invention] [Industrial Application Field] The present invention does not have a speed detector that directly detects speed.
This invention relates to a high-performance speed control device for an induction motor.

[Conventional technology]

As an example of controlling the speed of an induction motor without a speed detector, there is a well-known drive device that controls an induction motor (carrying motor) supplied with power from a transistor inverter, current source inverter, etc. to a constant V/f. . In this method, the output frequency of the inverter is given in an open loop, the output voltage of the inverter is controlled in a closed loop, and the voltage (V) command and the frequency (f) command are made proportional. However, this method has poor speed control responsiveness and poor control accuracy, so it is not often used as a high-performance speed control device.

A control device that applies vector control is a high-performance induction motor variable speed device, but this control device cannot succeed unless it is equipped with a speed detector. Therefore, the applicant has applied for a control device that estimates the amount of speed by calculation without detecting it with a detector, and controls the estimated value as the amount of speed (Japanese Patent Application No. 1983-
No. 212625: Hereinafter also referred to as the applied device. ).

FIG. 4 is a block diagram showing the device for which the application was filed. child~
The vector control principle is applied to the control of an induction motor (induction machine), which considers the motor's current, voltage, etc. as vector quantities, and converts these, which are alternating current quantities when observed from the stator windings. This method observes the amount from above the rotating magnetic field of the electric motor, converts it into a DC amount, separates this into a component parallel to the magnetic field, and a component perpendicular to the magnetic field, and attempts to control each component independently. It has been announced and is publicly known. The device shown in the figure is also a known speed control device with a speed detector (for example, Fuji Times Vo1.57.N110°1984,
The basic part of the control is exactly the same as in the section ``Application of GTO thyristors to inverters'' in 609-615@GTO), but since the actual value of speed cannot be obtained directly with this method, the speed must be determined by calculation. The major feature is that it is made as follows.The following describes Fig. 4.

In the figure, the primary current of an induction motor (IM) 2 is converted into a two-phase quantity, iβ, by a three-phase to two-phase converter 12. In addition, this amount is determined by the rotation coordinate (M-T
Coordinate system) Coordinates are transformed to quantity -+ IT. At this time, M
The axis, that is, the magnetic flux axis, is a current model type magnetic flux calculator (hereinafter referred to as
It is also simply called the current model. ) 10) is determined by the calculated phase φ of the magnetic flux. The current model is calculated according to the following equation.

φknee fω, dt-f(ω2+ω3勺dt...
...(1) As can be seen from equation (1), this current model λ equation is not valid without the rotor angular velocity, so the rotor angular velocity estimate ω calculated by the adjustment loop 200 is
2 is used to construct the current model. The speed calculation by this adjustment loop 200 will be described later. Coordinate transformation is performed from this φ according to the following equation.

In this way, if the -order current 11 is separated into lB*lt, then I
It is well known that M is a component that creates magnetic flux (magnetizing current), and IT is a component that creates torque (torque current).

This magnetizing current command IM' is given as the output of the magnetizing current command calculator 4. In the case of constant magnetic flux control, the arithmetic unit 4 gives a constant -, and when performing speed-dependent field weakening in a high-speed region, the arithmetic unit 4 gives <IM, which decreases as the speed increases.

The magnetizing current command Il is transmitted by the vector rotator 11.
- compared with the IM converted from the next current at the summing point 14;
This deviation is amplified by a PI (proportional integral) regulator 6) and becomes the M-axis component vM of the motor primary voltage vector command v1.

On the other hand, the command value N given by the speed setting device 100 is
The summing point 13 is compared with the calculated speed estimate N-ω2 (by the adjustment loop 200), and this deviation is amplified by the PI regulator 5 to become the torque current command iT. This IT
is iT created by vector rotator 11 and addition point 1
5, and this deviation is amplified by the PI controller 7 and becomes the T-axis component VT of the primary voltage vector command v1. This vM.

vT is input to the coordinate conversion circuit 8, and is converted into a stator coordinate quantity as shown in the following equation according to the phase φ□ of the magnetic flux calculated by the current model 10.

......(4) The primary voltage command V (1, Vβ converted into stator coordinate quantity is converted into an inverter pulse by the pulse generation circuit 9, and P
The power is applied to the WM inverter 1 and is then supplied to the induction motor 2.

Next, the speed estimation method will be explained. As shown in FIG. 4, the motor's primary voltage (1) and primary current i are input to the induced voltage calculation circuit 22 to obtain a fixed coordinate quantity Eα of the secondary induced voltage vector E2 of the induction machine.

Calculate Eβ. The calculation method is as follows. Note that vector quantities are shown with arrows, but are omitted unless particularly necessary.

The induced voltage calculation circuit 22 calculates the induced voltage according to the following equation.

Further, the relationship between the secondary induced voltage E2 and the secondary interlinkage flux F2 is expressed by the following equation (6).

As can be seen from equation (6), the phase of the secondary induced voltage leads the secondary interlinkage flux F2 by π/2. If the phase of the secondary interlinkage flux F2 from the fixed coordinate α axis is φ7, then the fixed coordinate jaEct of the secondary induced voltage E2.

Eβ is expressed by the following equation (7).

FIG. 5 shows a vector diagram of induced voltage showing the relationship of equation (7).

Next, the coordinates of EaIEβ obtained by the above calculation are converted to the magnetic flux axis calculated by the stain flow model 10 by the vector rotator (VD) 24. Here, if the phase of the magnetic flux V□ from the α-axis calculated using the current model is φ□, the relationship between φ7 and φ1 can be found.

(7) 2K By substituting equation (8), we get the following (9), (1
0) The formula is found.

EM--ECO8φ! S-φV + E sixφ! (
9)φV--Esia(φ7-φ1)...
...(9) E 7- E siaφIUnφ, +EC
QSφ, □□□φ7- E cos (φ7-φ, )
'・------(10) In equation (9), the magnetic flux calculated by the current model using the induced voltage vector E2, the detected secondary interlinkage magnetic flux vector 1F20 phase φ7, and the estimated speed value ω2 Vector phase φ! EM occurs when these do not match. FIG. 6 shows a vector diagram of the induced voltage when φ□ and φ7 do not match. φ1 shown in the figure is obtained by integrating ω2 and ω using an integrator 102. The adjuster 25 operates to achieve EM-0 and matches the phase of the magnetic flux. When the phases of the magnetic flux match, the angular velocity ω of the magnetic flux inside the motor and inside the control device.

match as shown in equation (11).

△ωyoω+♂−ω2+ωS −・・・−(11)1
2s Here, when the phase of the ceramic flux matches, the actual value of the angular velocity ω3 matches the actual value and the command value of the torque current and magnetizing current, so as is clear from equation (2), ,
angular velocity command value ωr. As a result, the output ω2 of the adjustment loop 200 is equal to the straightness of the rotor angular velocity, and the correct velocity is calculated. Note that the polarity circuit 23 determines the phase rotation of the magnetic flux from the ETK, and determines the polarity of the EM in accordance with the phase rotation.

[Problem that the invention seeks to solve]

As shown in Fig. 4, the above device requires a detection transformer (20) for detecting the output voltage and a circuit (24) for converting the coordinates of AC voltage into DC voltage, so the circuit configuration is complicated. There are problems in that it is complicated and leads to an increase in the cost of the control device.

Furthermore, when a detection transformer is used, the increase rate dv/di of the output voltage is high because it is a PWM inverter, and therefore current flows to the secondary side due to the influence of stray capacitance between the primary and secondary sides of the transformer. There is also the problem that detection noise is generated due to current and impedance on the secondary side, making it impossible to obtain accurate detection values.

Therefore, an object of the present invention is to provide a control device capable of high-performance speed control without directly detecting speed and motor voltage.

[Rounding method to solve problems]

When the command value of the motor voltage component parallel to the motor magnetic flux is calculated from each command value of the magnetizing current and torque current, the angular velocity of the magnetic flux, and the motor constant, the command value of the motor voltage component and the output from the magnetizing current regulator are obtained. A computing unit that computes the deviation from the computing unit, and a regulator that adjusts the output of the computing unit to zero are provided.

[Effect]

In a control device that does not have a speed detector, the voltage component vM parallel to the magnetic flux axis of the voltage vector v1 on the magnetic flux axis φ1 in the control device calculated by the current model method ((5), (4)) By focusing on the fact that the two components on the magnetic flux axis inside the motor do not match if there is a deviation between the two magnetic flux axes, and estimating the rotor angular velocity so that both components always match), the speed and enables high-precision speed control without directly detecting motor voltage.

〔Example〕

FIG. 1 is a block diagram showing a speed control device for an induction motor without a speed detector, showing an embodiment of the present invention. According to this example, the basic control part is exactly the same as in the case of the speed control device without a speed detector shown in FIG. ing.

Below, FIG. 1 will be explained.

The M-axis component voltage vM of the voltage vector can be calculated from the torque current and magnetizing current commands [iTpmu] and motor constants R1 and Lσ according to the following equation (12).

V -R, IB(-L, (Ml, 5T------
(12) However, in Tohe, when performing the above calculation, the output of the current regulator 6 is directly set as the M-axis component voltage v, IHI. Here, the relationship between vM and VM will be considered.

That is, if the magnetic flux axis (CM-T axis) in the control device determined by the current model method and the magnetic flux axis in the electric motor match, the torque current inside the electric motor.

Each actual value of the magnetizing current and each detected value of the torque current and magnetizing current obtained by coordinate transformation of the primary current vector flowing through the motor inside the control device are equal to each other. Therefore, if this detected value and command value are input to the current regulator, the output of the current regulator will be the voltage for iX on the M-T axis of the voltage vector that makes the current command value and detected value much equal. will give.

FIG. 2A is a vector diagram showing the relationship between voltage and current when both magnetic flux axes match, and FIG. 2B is a vector diagram showing the relationship between voltage and current when both magnetic flux axes do not match.

That is, when both magnetic flux axes coincide, the calculated value (1) and the regulator output (2) coincide with each other as shown in FIG. 2A. On the other hand, if the two magnetic flux axes do not match, the actual values 'T'F'M' of the torque current and magnetizing current in the motor and the detected values in the control device M1. - are different numbers, numbers 6 and 7 in Figure 1.
The current regulator shown by is the voltage vM' required to output this voltage 1.

As a result, VM obtained by calculating equation (12) and vM do not match. Therefore, if the deviation between vM'' and vM is set to ΔE, the following relationship is obtained. .

ΔE−vM″41−vM” 1■;−Ri +L, ω, i, ・(13) ΔE with M is caused by the fact that the estimated speed value and the actual speed value are different), and the magnetic flux axes do not match. Therefore, in this embodiment, the deviation ΔE is inputted to the regulator 25 in the adjustment loop 1300, and the speed is calculated by forming an adjustment loop in which ΔE is always zero.

FIG. 3 shows a specific example of the arithmetic circuit 30 for calculating the deviation ΔE as described above, which performs the calculation of equation (13). That is, multipliers 31, 52, 55 and addition points 34 are provided,
The multiplier 31 calculates the second page on the right side of equation (13),
Further, the third term on the right side of the equation is calculated by multipliers 52 and 55, and ΔE is calculated by applying these and the regulator output vM to the addition point 34 with the polarity shown.

Note that this output is given to the regulator 25 as in FIG. 4, and the estimated value ω2 of the rotor angular velocity is obtained from the output.

〔Effect of the invention〕

According to the present invention, not only is it possible to perform high-performance speed control of an induction machine without providing a speed detector, but also a voltage detector is not required, which simplifies the circuit configuration and reduces the cost of the control device. . Furthermore, since the influence of detection noise and the like when using a voltage detector is eliminated, accurate detection values can be obtained. Furthermore, there is the advantage that high-performance speed control can be achieved simply by adding a relatively simple adjustment loop to a conventional control device equipped with a speed detector.

[Brief explanation of the drawing]

Fig. 1 is a configuration diagram showing an embodiment of the present invention, Fig. 2A is a vector diagram showing the relationship between voltage and current when the magnetic flux axis in the control device and the magnetic flux axis in the motor coincide, and Fig. 2B is A vector diagram showing the relationship between voltage and current when these two magnetic flux axes do not match, Fig. 3 is a block diagram showing a specific example of a ΔE (voltage deviation) calculation circuit, and Fig. 4 is a configuration diagram showing the applied device. , Fig. 5 is a vector diagram showing the secondary induced voltage, and Fig. 6 is a vector diagram showing the secondary induced voltage when a deviation occurs between the calculated magnetic flux phase and the actual magnetic flux phase. . Description of symbols 1... PWM inverter, 2... Induction motor, 4... Magnetizing current command calculator, 5...
... Speed regulator (SR), 6... Magnetizing current regulator (ACR), 7... Torque current regulator (A
CR), 8... Coordinate conversion circuit, 9...
Pulse generation circuit, 10... Current magnetic flux calculator (current model), 11.24... Vector rotator (VD), 12.21...3 Phase-to-two phase converter, 13, 14, 15, 34... Addition point, 2
0... Voltage transformer, 22... Induced voltage calculation circuit, 23... Polarity circuit, 25...
・Adjuster, 30... Song ΔE calculation circuit, 31, 32.33
・・・・・・Multiplier, 1゜O・・・・Speed setter,
101...Subeshi angular velocity calculator, 102...
...Integrator, 200.300...Adjustment loop. W2A figure 112BB! 1 $31 30 v-115 diagram 6 diagram

Claims (1)

[Claims] The primary current of an induction motor, which is supplied via a power converter that can control the magnitude, frequency and phase of the output voltage, is divided into a component parallel to the magnetic flux of the motor (magnetizing current) and a component parallel to the magnetic flux of the motor (magnetizing current). In a variable speed control device for an induction motor that controls at least the motor torque by separating orthogonal components (torque current) and adjusting each independently, each command value of the magnetizing current and the torque current is provided. , calculate a command value of a motor voltage component parallel to the motor magnetic flux from the angular velocity of the magnetic flux and a motor constant, and calculate a deviation between the command value of the motor voltage component and the output from the regulator that adjusts the magnetizing current. 1. A variable speed control device for an induction motor, comprising: a computing unit that adjusts the output of the computing unit; and a regulator that adjusts the output of the computing unit to zero, and uses the output of the regulator as an estimated value of the motor angular velocity.